DE102019214155A1 - POWER CONVERTER - Google Patents

Abstract

A hybrid power converter and a method with low power losses over an extended conversion range are presented. The converter maintains low conversion losses associated with reduced inductor ripples, not just for a single conversion ratio, but over a wide range of conversion ratios. The power converter has a ground terminal, an input terminal for receiving an input voltage, and an output terminal for providing an output voltage with a target conversion ratio. The power converter has an inductor; a first flying capacitor selectively coupled to the inductor; a second flying capacitor selectively coupled to the inductor; a network of switches; and a driver configured to operate the converter in a first mode associated with a first range of conversion ratios.

Description

Technical area

The present disclosure relates to a power converter and a method of operating the same. In particular, the present disclosure relates to a hybrid power converter with low power losses over an extended conversion range.

background

In recent years, portable computing devices such as smartphones, tablets and notebooks have increased their computing power, screen resolution and display frame rate. These advances have been made possible by submicron silicon technology approaching 10nm and below, enabling the formation of ultra-narrow gate structures. Ultra-narrow gate structures have an increased leakage current for each transistor.

Given that central processing units (CPUs) and graphics processing units (GPUs) consist of several hundred million transistors, the leakage current of a modern microprocessor is significant. In order to reduce battery consumption, the embedded computing cores are typically disconnected from the power supply as often as possible. As a result, the required computing power is provided within short operating “bursts”. Thus, the performance profile of a modern mobile computing device is dominated by relatively long periods of standby currents in the mA range, which are interrupted by pulses of high peak currents (in the range of 20 A and higher). The challenge for a power management unit is to provide low currents with high conversion efficiency to optimize battery life, combined with the provision of high currents without saturation effects and with a stable output voltage.

Smartphones and tablet computers are typically operated with a Li-Ion battery pack with a nominal output voltage of 3.6V. The CPUs and GPUs manufactured in silicon technology with gate lengths of 10 nm and less require a supply voltage of around 0.9V. The corresponding voltage step-down converter must optimize its efficiency by a typical V _out / V _in conversion ratio of 0.9V / 3.6V = 0.25V.

Conventional two-stage and three-stage buck converters are limited by significant conversion losses spanning a wide range of conversion ratios. The battery voltage of a typical Li-ion battery cell can drop from 4.2V to 2.5V in the course of its use. There is, therefore, a need for a converter that can maintain low conversion losses associated with reduced inductor ripples, not just for a single conversion ratio, but over a wide range of conversion ratios.

Summary

According to a first aspect of the disclosure, there is provided a power converter having a ground terminal, an input terminal for receiving an input voltage, and an output terminal for providing an output voltage with a target conversion ratio, the power converter having an inductor; a first flying capacitor selectively coupled to the inductor; a second flying capacitor selectively coupled to the inductor; a network of switches; and a driver configured to operate the converter in a first mode associated with a first range of conversion ratios; wherein in the first mode the driver is configured to drive the network of switches with a first sequence of states during a drive period, the first sequence of states having a first state and a second state, wherein in the first state one of the input terminal and the The ground terminal is coupled to the output terminal via a first path including the second flying capacitor and bypassing the inductor, and wherein the ground terminal is coupled to the output terminal via a second path including the first flying capacitor and the inductor; wherein in the second state the ground connection is coupled to the output connection via a third path comprising the first flying capacitor, the second flying capacitor and the inductor.

For example, the first range of conversion ratios may be Vout / Vin 1/3.

Optionally, the driver is configured to operate the converter in a second mode that is associated with a second range of conversion ratios; wherein in the second mode the driver is configured to control the network of switches with a second sequence of states, wherein the second sequence of states has the first state, the second state and a third state, wherein in the third state the input terminal has a Path that has the first flying capacitor with the The output terminal is coupled and wherein the ground terminal is coupled to the output terminal via a path comprising the second flying capacitor and the inductor.

Optionally, in the third state, the input terminal is coupled to the output terminal via a path that has the first flying capacitor and the inductor.

For example, the second range of conversion ratios can be 1/3 <Vout / Vin <1/2. The states in the second sequence of states can be provided in a certain order, for example: first state / second state / third state / first state.

Optionally, the driver is configured to operate the converter in a third mode that is associated with a third range of conversion ratios; wherein in the third mode the driver is configured to control the network of switches with a third sequence of states, the third sequence of states having the first state and the third state.

For example, the third range of conversion ratios Vout / Vin 1/2.

Optionally, the first sequence has a demagnetization state in which the ground connection is coupled to the output connection via a demagnetization path having the inductor. In addition, in the demagnetization state, the input connection can be coupled to the output connection via the first path.

The states in the first sequence can be provided in a certain order, for example: first state / demagnetization state / second state / demagnetization state.

Optionally, the third sequence has a magnetization state in which the input connection is coupled to the output connection via a magnetization path having the inductor. In addition, in the magnetization state, the input connection can be coupled to the output connection via the first path.

The states in the third sequence can be provided in a certain order, for example: first state / magnetization state / third state / magnetization state.

Optionally, the driver is designed to change a first duration of the first state, a second duration of the second state and a third duration of the third state based on the target conversion ratio.

Optionally, the driver is designed to change a duration of the magnetization state based on the target conversion ratio.

Optionally, the driver is configured to change a duration of the demagnetization state based on the target conversion ratio.

Optionally, the network of switches includes a first input switch coupled to the input port; a second input switch for coupling the first flying capacitor to the input terminal via the first input switch; a first ground switch for coupling the first flying capacitor to ground; and a second ground switch for coupling the second flying capacitor to ground; wherein the inductor has a first terminal and a second terminal, the second terminal being coupled to the output terminal.

Optionally, each of the first flying capacitor and the second flying capacitor has a first terminal selectively coupled to the input terminal and a second terminal selectively coupled to ground; wherein the network of switches includes a first capacitor switch coupled to the first terminal of the first flying capacitor; a second capacitor switch coupled to the second terminal of the first flying capacitor; and a fourth capacitor switch coupled to the second terminal of the second flying capacitor.

Optionally, the first connection of the inductor is coupled to the first flying capacitor via the second capacitor switch and to the second flying capacitor via the fourth capacitor switch; and wherein the first capacitor switch is coupled to the output terminal.

Optionally, the network of switches has a third capacitor switch that is coupled to the first terminal of the second flying capacitor.

Optionally, the network of switches has a third input switch in order to couple the second flying capacitor to the input terminal via the first input switch.

Optionally, the first connection of the inductor is coupled to the first flying capacitor via the first capacitor switch and the second capacitor switch; where the first port the inductor is coupled to the second flying capacitor via the third capacitor switch; and wherein the fourth capacitor switch is coupled to the output terminal.

Optionally, the first flying capacitor is coupled to the second connection of the inductor via an output switch.

According to a second aspect of the disclosure, there is provided a method for converting an input voltage provided at an input terminal into an output voltage provided at an output terminal, the method comprising: providing an inductor; Providing a first flying capacitor selectively coupled to the inductor; Providing a second flying capacitor selectively coupled to the inductor; Providing a network of switches; Operating the converter in a first mode associated with a first range of conversion ratios by driving the network of switches with a first sequence of states during a driving period, the first sequence of states having a first state and a second state, wherein in the first state one of the input terminal and the ground terminal is coupled to the output terminal via a first path which has the second flying capacitor and which bypasses the inductor, and wherein the remaining terminal of the input terminal and the ground terminal via a second path which the the first flying capacitor and the inductor coupled to the output terminal; wherein in the second state the ground connection is coupled to the output connection via a third path comprising the first flying capacitor, the second flying capacitor and the inductor.

Optionally, the first sequence has a demagnetization state in which the ground connection is coupled to the output connection via a demagnetization path having the inductor.

Optionally, the method includes operating the converter in a second mode associated with a second range of conversion ratios by actuating the network of switches with a second sequence of states, the second sequence of states being the first state, the second state and a third state, wherein in the third state the input terminal is coupled to the output terminal via a path including the first flying capacitor, and wherein the ground terminal is coupled to the output terminal via a path including the second flying capacitor and the inductor is coupled.

Optionally, in the third state, the input terminal is coupled to the output terminal via a path that has the first flying capacitor and the inductor.

Optionally, the method includes operating the converter in a third mode associated with a third range of conversion ratios by driving the network of switches with a third sequence of states, the third sequence of states being the first state and the third state having.

Optionally, the third sequence has a magnetization state in which the input connection is coupled to the output connection via a magnetization path having the inductor.

The options described in relation to the first aspect of the disclosure are also common to the second aspect of the disclosure.

description

• 1A ein Diagramm eines zweistufigen Abwärts-Wandlers ist;
• 1B ein Diagramm eines dreistufigen Abwärts-Wandlers ist;
• 1C eine Simulation der Induktorstromwelligkeit der Wandler der 1A und 1B als eine Funktion eines Umwandlungsverhältnisses ist;
• 2 ein Diagramm eines DC-DC-Wandlers gemäß der Offenbarung ist;
• 3A ein Diagramm des DC-DC-Wandlers von 2 ist, der in einem ersten Zustand betrieben wird;
• 3B ein Diagramm des DC-DC-Wandlers von 2 ist, der in einem zweiten Zustand betrieben wird;
• 3C ein Diagramm des DC-DC-Wandlers von 2 ist, der in einem Zwischenmagnetisierungszustand betrieben wird;
• 3D ein Diagramm des DC-DC-Wandlers von 2 ist, der in einem Zwischenentmagnetisierungszustand betrieben wird;
• 3E ein Diagramm des DC-DC-Wandlers von 2 ist, der in einem anderen Zustand betrieben wird;
• 4 eine beispielhafte Ansteuersequenz zum Betreiben des DC-DC-Wandlers von 2 ist;
• 5 ein Diagramm eines anderen DC-DC-Wandlers gemäß der Offenbarung ist;
• 6 ein Diagramm des DC-DC-Wandlers von 5 ist, der in einem Zwischenzustand betrieben wird;
• 7 ein Diagramm eines weiteren DC-DC-Wandlers gemäß der Offenbarung ist;
• 8 eine Simulation der Induktorstromwelligkeit der Wandler der 1A, 1B und 2 als eine Funktion des Umwandlungsverhältnisses ist;
• 9 eine Simulation des Induktorkernleistungsverlustes der Wandler der 1A, 1B und 2 als eine Funktion des Umwandlungsverhältnisses ist;

The disclosure is described in more detail below by way of example and with reference to the accompanying drawings, in which:

• 1A Figure 3 is a diagram of a two stage buck converter;
• 1B Figure 3 is a diagram of a three stage buck converter;
• 1C a simulation of the inductor current ripple of the converter 1A and 1B as a function of a conversion ratio;
• 2 Figure 3 is a diagram of a DC-DC converter in accordance with the disclosure;
• 3A a diagram of the DC-DC converter of 2 is operated in a first state;
• 3B a diagram of the DC-DC converter of 2 is operated in a second state;
• 3C a diagram of the DC-DC converter of 2 is operated in an intermediate magnetization state;
• 3D a diagram of the DC-DC converter of 2 which operates in an intermediate demagnetization state;
• 3E a diagram of the DC-DC converter of 2 is operated in a different state;
• 4th an exemplary control sequence for operating the DC-DC converter from 2 is;
• 5 Figure 12 is a diagram of another DC-DC converter in accordance with the disclosure;
• 6 a diagram of the DC-DC converter of 5 is operated in an intermediate state;
• 7th Figure 12 is a diagram of another DC-DC converter in accordance with the disclosure;
• 8th a simulation of the inductor current ripple of the converter 1A , 1B and 2 is as a function of the conversion ratio;
• 9 a simulation of the inductor core power loss of the converters of the 1A , 1B and 2 is as a function of the conversion ratio;

10 Figure 3 is a flow diagram of a method for converting power with a target conversion ratio.

The 1A and 1B show the topologies of conventional two-stage and three-stage buck converters. Such converters can be used to provide an output voltage over a range of conversion ratios. Conversion losses are proportional to the frequency of the inductor current ripple and to the square of the current ripple amplitude. Regulated DC-DC switching converters can minimize conversion losses by reducing switching frequency and inductor current ripple. 1C is a simulation of the inductor current ripple as a function of the conversion ratio required for the converters of the 1A and 1B is obtained.

The normalized inductor current ripple 110 and 120 is shown for the two-stage buck converter and the three-stage buck converter. At a conversion ratio of V _out / V _in = 0.25, the two-stage step-down converter shows 75% of its peak inductor current ripple. This requires either a high switching frequency, which reduces the converter efficiency, or a large inductance, hence a large inductor. For a given inductor form factor, this would lead to increased direct current resistance (DCR) and increased conduction loss, which would ultimately reduce the efficiency of the converter. Hybrid converter topologies, such as the three-stage step-down converter, typically reduce the inductor ripple at V _out / V _in = 0.25 by a factor of 3. Compared to the two-stage step-down converter, this corresponds to a three times lower switching frequency or three times lower inductance. At a conversion ratio V _out / V _in = 0.25, however, the inductor current ripple remains significant and has its highest amplitude for the topology of the three-stage buck converter.

2 Figure 13 is a diagram of a DC-DC converter 200 according to the revelation. The DC-DC converter 200 includes two capacitors C1 and C2 and an inductor L connected between an input node 202 and an exit node 204 via a network of switches, which consists of nine switches S1, S2, S3, S4, S5, S6, S7, S8 and S9, is coupled. An input capacitor Cin is between the input node 202 and ground and an output capacitor Cout is between the output node 204 and mass provided. The capacitors Cin and Cout are connected to a fixed ground voltage and can be referred to as reservoir capacitors. The capacitors C1 and C2 have connections that are provided with varying voltages and can be referred to as flying capacitors.

The first flying capacitor C1 is connected to ground via switch S4 and to the input node via switches S1 and S9 202 coupled. Similarly, the second flying capacitor C2 is connected to ground via switch S8 and to the input node via switches S5 and S9 202 coupled. The first flying capacitor C1 has a first terminal that connects to the node 206 is coupled, and a second connector that connects to the node 208 is coupled. The second flying capacitor C2 has a first terminal that connects to the node 210 is coupled, and a second connector that connects to the node 212 is coupled. The second flying capacitor C2 is also connected to the output node 204 coupled via switch S7. The inductor L has a first connection at a node 214 and a second connector connected to the output node 204 is coupled. The first connector on the node 214 is via the switch S2 to the node 206 , via switch S6 to the node 210 and via switch S3 to the node 208 coupled. A driver 220 is provided to receive a variety of control signals Ct1-Ct9 to generate to operate the switches S1-S9 respectively.

The topology of the converter 200 is called asymmetrical topology, as the voltage across C1 can differ from the voltage across C2. The voltage across C2 is V _in - V _out , while the voltage across C1 can assume different values depending on the selected conversion ratio. Continuous input current can be achieved if C1 is charged to around V _out .

oder $\frac{V_{out}}{V_{in}}\le \mathrm{0,33.}$

Um die Induktorstromwelligkeiten in diesem Umwandlungsbereich zu begrenzen, kann die Spannung über dem fliegenden Kondensator C1 auf V_C1 ~ (V_in - V_out)/2 geregelt werden. Für eine Minimumspannung über den Schaltern S2 und S3 kann der fliegende Kondensator C1 alternativ, z.B. auf V_C1 ~ V_out, geregelt sein. In diesem Fall ist der Induktorkernverlust geringfügig erhöht, zum Beispiel auf das Doppelte der Stromwelligkeitsamplitude bei der halben Frequenz. Der zweite Modus entspricht einem Umwandlungsverhältnisbereich von $2\le \frac{V_{in}}{V_{Out}}\le 3$

oder $\mathrm{0,33}\le \frac{V_{out}}{V_{in}}\le \mathrm{0,5.}$

In diesem zweiten Modus wird die Spannung über dem fliegenden Kondensator C1 V_C1 ~ V_out. Der dritte Modus entspricht einem Umwandlungsverhältnisbereich von $\frac{V_{in}}{V_{out}}\le 2$

oder $\frac{V_{out}}{V_{in}}\ge \mathrm{0,5.}$

In dem dritten Modus wird die Spannung über dem fliegenden Kondensator C1 V_C1 - Vout.The DC-DC converter can operate in three modes, referred to as first, second and third modes, which correspond to three different ranges of conversion ratios. The first mode corresponds to a conversion ratio range of $\frac{V_{in}}{V_{Out}}\ge 3$

or $\frac{V_{Out}}{V_{in}}\le \mathrm{0.33.}$

In order to limit the inductor current ripples in this conversion range, the voltage across the flying capacitor C1 can be regulated to V _C1 ~ (V _in - V _out ) / 2. For a minimum voltage across switches S2 and S3, the flying capacitor C1 can alternatively be regulated, for example to V _C1 ~ V _out . In this case the inductor core loss is increased slightly, for example to twice the current ripple amplitude at half the frequency. The second mode corresponds to a conversion ratio range of $2\le \frac{V_{in}}{V_{Out}}\le 3$

or $0.33\le \frac{V_{Out}}{V_{in}}\le \mathrm{0.5.}$

In this second mode, the voltage across the flying capacitor C1 becomes V _C1 ~ V _out . The third mode corresponds to a conversion ratio range of $\frac{V_{in}}{V_{Out}}\le 2$

or $\frac{V_{Out}}{V_{in}}\ge \mathrm{0.5.}$

In the third mode, the voltage across the flying capacitor C1 becomes V _C1 - Vout.

The driver 220 operates the converter 200 using a sequence of two or three states which are selected from a large number of states depending on the selected operating mode. The states can be selected from five states identified as states A, B, C, D and E.

3A shows the DC-DC converter of 2 which operates in a first state, referred to as state A. In state A, switches S2, S4, S5, S7 and S9 are closed, while the remaining switches S1, S3, S6 and S8 are open. The entry node 202 is via a first path comprising S9, S5, C2 and S7 with the output node 204 coupled, thereby bypassing the inductor L. The ground is connected to the output node via a second path comprising S4, C1, S2 and L 204 coupled.

3B shows the DC-DC converter of 2 which operates in a second state, referred to as state B. In state B, switches S1, S3, S6, S8 and S9 are closed, while the remaining switches S2, S4, S5 and S7 are open. The entry node 202 is via a third path that includes S9, S1, C1, S3 and L to the output node 204 coupled. The ground is via a fourth path that includes S8, C2, S6, and L to the output node 204 coupled.

3C shows the DC-DC converter of 2 operating in a third state referred to as state C or magnetization state. In state C, switches S1, S2, S5, S6 and S9 are closed, while the remaining switches S3, S4, S8 are open. The switch S7 may or may not be closed. The input is coupled to the output via a fifth double path, which has S9, S1, S2, L or S9, S5, S6, L. When S7 is closed, it is the input node 202 also via the first path (S9, S5, C2, S7) with the output node 204 coupled.

3D shows the DC-DC converter of 2 which operates in a fourth state referred to as state D or degaussing state. In state D, switches S3, S4, S5 and S9 are closed, while the remaining switches S1, S2, S6 and S8 are open. The switch S7 may or may not be closed. The ground is connected to the output node via a sixth path comprising S4, S3 and L 204 coupled. When S7 is closed, it is the input node 202 also via the first path (S9, S5, C2, S7) with the output node 204 coupled.

3E shows the DC-DC converter of 2 which operates in a fifth state referred to as state E. In state E, switches S1, S3, S5 and S8 are closed, while the remaining switches S2, S4, S6, S7 and S9 are open. The ground is connected to the output node via a seventh path that includes S8, C2, S5, S1, C1, S3 and L 204 coupled.

assoziiert ist, enthält eine erste Modussequenz die Zustände A, E und optional D. Für ein Sollumwandlungsverhältnis von 1/3 würde die erste Modussequenz nur die Zustände A und E enthalten. Für Umwandlungsverhältnisse von weniger als 1/3 würde die erste Modussequenz jedoch auch den Zustand D enthalten. Diese Zustände können in einer bestimmten Reihenfolge vorgesehen werden, zum Beispiel A/D/E/D.The driver can select a specific sequence of states for each operating mode. In the first mode, the one with a conversion ratio $\frac{V_{Out}}{V_{in}}\le 0.33$

is associated, a first mode sequence contains the states A, E and optionally D. For a target conversion ratio of 1/3, the first mode sequence would only contain the states A and E. For conversion ratios less than 1/3, however, the first mode sequence would also contain state D. These states can be provided in a specific order, for example A / D / E / D.

assoziiert ist, umfasst eine zweite Modussequenz die Zustände A, E und B. Diese Zustände können in einer bestimmten Reihenfolge vorgesehen werden, zum Beispiel A/E/B/A.In the second mode of operation, the one with the conversion ratio in the range of $0.33<\frac{V_{Out}}{V_{in}}<0.5$

is associated, a second mode sequence comprises the states A, E and B. These states can be provided in a certain order, for example A / E / B / A.

assoziiert ist, umfasst eine dritte Modussequenz die Zustände A, B und optional C. Diese Zustände können in einer bestimmten Reihenfolge vorgesehen werden, zum Beispiel A/C/B/C. 4 zeigt eine beispielhafte Ansteuersequenz zum Betreiben des DC-DC-Wandlers von 2 in dem dritten Modus. Der Treiber steuert den DC-DC-Wandler mit dem Zustand A (Wellenform 410) zwischen den Zeitpunkten t0 und t1 für eine Dauer TA, mit dem Zustand C (Wellenform 430) zwischen den Zeitpunkten t1 und t2 für eine Dauer TC, mit dem Zustand B (Wellenform 420) zwischen den Zeitpunkten t2 und t3 für eine Dauer TB, und dann mit dem Zustand C zwischen den Zeitpunkten t3 und t4 an. Diese Sequenz wird dann über die Zeit wiederholt, um die erforderliche Ausgangsleistung zu liefern. Es ist offensichtlich, dass zu den Zeitpunkten t1, t2, t3 und t4 eine Totzeit eingeführt werden kann. Die Werte für die Dauer TA, TB und TC, auch als Zustandsarbeitszyklus bezeichnet, können abhängig von dem Sollumwandlungsverhältnis variieren. Die Zustandsdauern müssen auch die Volt-Sekunden-Balance über den Induktor erfüllen.In the third operating mode, the one with the conversion ratio $\frac{V_{Out}}{V_{in}}\ge 0.5$

is associated, a third mode sequence includes the States A, B and optionally C. These states can be provided in a certain order, for example A / C / B / C. 4th FIG. 10 shows an exemplary control sequence for operating the DC-DC converter from FIG 2 in the third mode. The driver controls the DC-DC converter with state A (waveform 410 ) between times t0 and t1 for a duration TA, with state C (waveform 430 ) between times t1 and t2 for a duration TC, with state B (waveform 420 ) between times t2 and t3 for a duration TB, and then with state C between times t3 and t4. This sequence is then repeated over time to provide the required output power. It is obvious that a dead time can be introduced at times t1, t2, t3 and t4. The values for the duration TA, TB and TC, also referred to as the state duty cycle, can vary depending on the target conversion ratio. The state durations must also meet the volt-second balance across the inductor.

ist TC = 0 und TB = 2 TA. Die Topologie des DC-DC-Wandlers von 2 kann modifiziert werden, um eine Leistung in einem bestimmten Umwandlungsbereich zu verbessern.The voltage across the flying capacitor C1 is V _C1 ~ Vout. For continuous switching and to meet the volt x second balance across the inductor L, the duration of the switching state TB must be longer than TA. For a balanced average current through both switching phases (C1 and C2) the continuous ratio TB / TA is 2: 1. For a conversion ratio $\frac{V_{Out}}{V_{in}}=0.5$

is TC = 0 and TB = 2 TA. The topology of the DC-DC converter from 2 can be modified to improve performance in a particular conversion range.

betrieben. Der Wandler 500 ähnelt dem unter Bezugnahme auf 2 beschriebenen Wandler 200, in dem bestimmte Teile der Schaltung hinzugefügt oder modifiziert wurden. Die gleichen Bezugszeichen wurden verwendet, um entsprechende Komponenten darzustellen, und ihre Beschreibung wird der Kürze halber nicht wiederholt. 5 shows a diagram of a DC-DC converter 500 designed to minimize inductor core and conduction losses when using an output voltage $\frac{V_{Out}}{V_{in}}>\frac{1}{2}$

operated. The converter 500 is similar to the one referring to 2 described converter 200 in which certain parts of the circuit have been added or modified. The same reference numbers have been used to represent corresponding components and their description will not be repeated for brevity.

In this exemplary embodiment, a further switch S10 is provided. The switch S10 has a first connection that connects to the first flying capacitor at the node 208 is coupled, and a second port connected to the output node 204 is coupled.

assoziiert ist, kann der DC-DC-Wandler 500 mit einer Sequenz von Modi betrieben werden, die die Zustände A, F und optional C enthält. Diese Zustände können in einer bestimmten Reihenfolge vorgesehen werden, zum Beispiel A/C/F/C. Anders ausgedrückt, der Zustand F ersetzt den Zustand B, der oben unter Bezugnahme auf 3B beschrieben wurde. In the third operating mode, the one with the conversion ratio $\frac{V_{Out}}{V_{in}}\ge 0.5$

can be associated with the DC-DC converter 500 operated with a sequence of modes that includes states A, F and optionally C. These states can be provided in a specific order, for example A / C / F / C. In other words, state F replaces state B mentioned above with reference to FIG 3B has been described.

6 FIG. 13 is a diagram of the DC-DC converter of FIG 5 operated in a main state F. In state F, switches S1, S6, S8, S9 and S10 are closed, while the remaining switches S2, S3, S4, S5 and S7 are open. The entry node 202 is via a path comprising S9, S1, C1, and S10 to the output node 204 coupled. The ground is via a path that includes S8, C2, S6 and L to the output node 204 coupled. In states A and C, switch S10 remains open. The voltage across the converter's flying capacitor C1 500 becomes V _C1 ~ V _in - V _out and the duration TF of switching state F can be the same as the duration TA of switching state A.

Der Wandler 700 umfasst zwei fliegende Kondensatoren C1 und C2, einen Induktor und ein Netzwerk von nur sieben Schaltern S1, S2, S3, S4, S7, S8 und S9. Ein Eingangskondensator Cin ist zwischen dem Eingangsknoten 702 und Masse vorgesehen und ein Ausgangskondensator Cout ist zwischen dem Ausgangsknoten 704 und Masse vorgesehen.
Der erste fliegende Kondensator C1 ist über den Schalter S4 mit Masse und über die Schalter S1 und S9 mit dem Eingangsknoten 702 gekoppelt. In ähnlicher Weise ist der zweite fliegende Kondensator C2 über den Schalter S8 mit Masse und über den Schalter S9 mit dem Eingangsknoten 202 gekoppelt. Der erste fliegende Kondensator C1 hat einen ersten Anschluss, der über S2 mit dem Ausgang gekoppelt ist, und einen zweiten Anschluss, der über S3 mit dem Induktor L gekoppelt ist. Der zweite fliegende Kondensator C2 ist über den Schalter S7 mit L gekoppelt. Der Induktor L hat einen ersten Anschluss an dem Knoten 714 und einen zweiten Anschluss, der mit dem Ausgangsknoten 704 gekoppelt ist. 7th shows a diagram of a DC-DC converter 700 , which is designed to minimize conduction losses, when operated with an output voltage $\frac{V_{Out}}{V_{in}}\le \frac{1}{3}.$

The converter 700 includes two flying capacitors C1 and C2, an inductor and a network of just seven switches S1, S2, S3, S4, S7, S8 and S9. An input capacitor Cin is between the input node 702 and ground and an output capacitor Cout is between the output node 704 and mass provided.
The first flying capacitor C1 is connected to ground via switch S4 and to the input node via switches S1 and S9 702 coupled. Similarly, the second flying capacitor C2 is connected to ground via switch S8 and to the input node via switch S9 202 coupled. The first flying capacitor C1 has a first terminal, which is coupled to the output via S2, and a second terminal, which is coupled to the inductor L via S3. The second flying capacitor C2 is coupled to L via switch S7. The inductor L has a first connection at the node 714 and a second connector connected to the output node 704 is coupled.

) Zustand D. In dem Zustand A' sind die Schalter S2, S4, S7 und S9 geschlossen, während die übrigen Schalter S1, S3 und S8 offen sind. Der Eingangsknoten 702 ist über einen ersten Pfad, der S9, C2, S7 und L aufweist, mit dem Ausgangsknoten 704 gekoppelt. Die Masse ist über einen zweiten Pfad, der S4, C1, S2 aufweist, mit dem Ausgangsknoten 704 gekoppelt und umgeht somit L.The converter 700 operates with a sequence of states including: State A ', State E, and (for a conversion ratio $\frac{V_{Out}}{V_{in}}<\frac{1}{3}$

) State D. In state A ', switches S2, S4, S7 and S9 are closed, while the other switches S1, S3 and S8 are open. The entry node 702 is via a first path comprising S9, C2, S7 and L to the output node 704 coupled. The ground is connected to the output node via a second path comprising S4, C1, S2 704 coupled and thus bypasses L.

The in terms of the 2 to 7th described DC-DC converters have been described as step-down converters, which are also referred to as buck converters. It will be apparent that these converters can be operated in reverse as boost converters (i.e. the input is used as an output and the output is used as an input) to achieve an upconversion.

8th shows the simulations of the inductor current ripples of a two-stage buck converter 810 , a three-stage buck converter 820 and the converter according to the disclosure 830 as a function of transformation ratios. The converter of the disclosure extends the maximum converter output current beyond the maximum rated current of the inductor and reduces inductor conduction loss via an output current path that bypasses the inductor (up to 33% of Iout). The converter of the disclosure also provides an additional conversion range with reduced inductor current ripple by the ratio V _out / V _in = 0.33.

9 shows the simulations of the inductor core power loss of a two-stage buck converter 910 , a three-stage buck converter 920 and the converter according to the disclosure 930 as a function of transformation ratios. A typical inductor power loss, based on the Steinmetz equation, is proportional to the frequency of the inductor current ripple and the square of the current ripple amplitude I _pp .
For a conversion ratio V _out / V _in > 0.5, the frequency of the inductor current ripple of a three-stage buck converter and the converter of the disclosure is approximately twice the frequency of the two-stage buck converter. The inductor current ripple is twice the frequency of the two-stage buck converter below V _out / V _in = 0.5 for the three-stage buck converter and below V _out / V _in = 0.33 for the converter topology of the invention. 9 shows the normalized loss comparison based on these two parameters.

The normalized inductor core loss is in an extended conversion range from V _out / V _in > 0.25 to Vout / Vin> 0.5 930 of a converter according to the disclosure less than ~ 5% of the losses 910 a conventional two-stage buck converter. The three-stage buck converter implements low inductor core loss 920 only about V _out / V _in ~ 0.5.

10 FIG. 13 is a flowchart of a method for converting an input voltage provided at an input terminal to an output voltage provided at an output terminal with a target conversion ratio. In step 1010 an inductor is provided. In step 1020 a first flying capacitor is provided which is selectively coupled to the inductor. In step 1030 a second flying capacitor is provided which is selectively coupled to the inductor. In step 1040 a network of switches is provided. In step 1050 For example, the converter is operated in a first mode associated with a first range of conversion ratios by actuating the network of switches with a first sequence of states during a drive period. The first sequence of states has a first state and a second state. In the first state, one of the input terminal and a ground terminal is coupled to the output terminal via a first path that includes the second flying capacitor and which bypasses the inductor, and the remaining terminal of the input terminal and the ground terminal is coupled to the output terminal via a second path that includes the having the first flying capacitor and the inductor coupled to the output terminal. In the second state, the ground connection is coupled to the output connection via a third path, which has the first flying capacitor, the second flying capacitor and the inductor.

It will be apparent to those skilled in the art that variations of the disclosed arrangements are possible without departing from the disclosure. For example, the flying capacitors can be implemented as single or multiple capacitors connected in series and / or in parallel. Alternatively, a capacitor network can be used. Such a capacitor network can change the configuration during operation of the converter. Accordingly, the above description of the specific embodiment is exemplary only and not for purposes of limitation. It will be apparent to those skilled in the art that minor modifications can be made without materially changing the operation described.

Claims (22)

A power converter having a ground connection, an input connection for receiving an input voltage and an output connection for providing an output voltage with a target conversion ratio, the power converter comprising an inductor; a first flying capacitor selectively coupled to the inductor; a second flying capacitor selectively coupled to the inductor; a network of switches; and a driver configured to operate the converter in a first mode associated with a first range of conversion ratios; wherein in the first mode the driver is configured to drive the network of switches with a first sequence of states during a drive period, the first sequence of states having a first state and a second state, wherein in the first state one of the input terminal and the Ground terminal is coupled to the output terminal via a first path which has the second flying capacitor and which bypasses the inductor, and wherein the remaining terminal of the input terminal and the ground terminal via a second path which has the first flying capacitor and the inductor with is coupled to the output port; wherein in the second state the ground connection is coupled to the output connection via a third path comprising the first flying capacitor, the second flying capacitor and the inductor. The power converter according to Claim 1 wherein the driver is configured to operate the converter in a second mode associated with a second range of conversion ratios; wherein in the second mode the driver is configured to control the network of switches with a second sequence of states, wherein the second sequence of states has the first state, the second state and a third state, wherein in the third state the input terminal has a A path including the first flying capacitor coupled to the output terminal, and wherein the ground terminal is coupled to the output terminal via a path including the second flying capacitor and the inductor. The power converter according to Claim 2 wherein in the third state, the input terminal is coupled to the output terminal via a path including the first flying capacitor and the inductor. The power converter according to Claim 2 or 3 wherein the driver is configured to operate the converter in a third mode associated with a third range of conversion ratios; wherein in the third mode the driver is configured to control the network of switches with a third sequence of states, the third sequence of states having the first state and the third state. The power converter according to claim 1, wherein the first sequence has a demagnetization state in which the ground connection is coupled to the output connection via a demagnetization path having the inductor. The power converter according to Claim 4 , wherein the third sequence has a magnetization state in which the input connection is coupled to the output connection via a magnetization path having the inductor. The power converter according to one of the Claims 2 to 6 wherein the driver is configured to change a first duration of the first state, a second duration of the second state, and a third duration of the third state based on the target conversion ratio. The power converter according to Claim 6 or 7th wherein the driver is configured to change a duration of the magnetization state based on the target conversion ratio. The power converter according to Claim 5 wherein the driver is configured to change a duration of the demagnetization state based on the target conversion ratio. The power converter of any preceding claim, wherein the network comprises switches a first input switch coupled to the input port; a second input switch for coupling the first flying capacitor to the input terminal via the first input switch; a first ground switch for coupling the first flying capacitor to ground; and a second ground switch for coupling the second flying capacitor to ground; wherein the inductor has a first terminal and a second terminal, the second terminal being coupled to the output terminal. The power converter according to Claim 10 wherein each of the first flying capacitor and the second flying capacitor has a first terminal selectively coupled to the input terminal and a second terminal selectively coupled to ground; wherein the network of switches includes a first capacitor switch coupled to the first terminal of the first flying capacitor; a second capacitor switch coupled to the second terminal of the first flying capacitor; and a fourth capacitor switch coupled to the second terminal of the second flying capacitor. The power converter according to Claim 11 wherein the first terminal of the inductor is coupled to the first flying capacitor via the second capacitor switch and to the second flying capacitor via the fourth capacitor switch; and wherein the first capacitor switch is coupled to the output terminal. The power converter according to Claim 11 wherein the network of switches includes a third capacitor switch coupled to the first terminal of the second flying capacitor. The power converter according to Claim 13 wherein the network of switches includes a third input switch for coupling the second flying capacitor to the input port through the first input switch. The power converter according to Claim 14 wherein the first terminal of the inductor is coupled to the first flying capacitor via the first capacitor switch and the second capacitor switch; wherein the first terminal of the inductor is coupled to the second flying capacitor via the third capacitor switch; and wherein the fourth capacitor switch is coupled to the output terminal. The power converter according to Claim 15 , wherein the first flying capacitor is coupled to the second terminal of the inductor via an output switch. A method for converting an input voltage provided at an input terminal into an output voltage provided at an output terminal, the method comprising: Providing an inductor; Providing a first flying capacitor selectively coupled to the inductor; Providing a second flying capacitor selectively coupled to the inductor; Providing a network of switches; Operating the converter in a first mode associated with a first range of conversion ratios by driving the network of switches with a first sequence of states during a driving period, the first sequence of states having a first state and a second state, wherein in the first state one of the input terminal and a ground terminal is coupled to the output terminal via a first path which includes the second flying capacitor and which bypasses the inductor, and wherein the remaining terminal of the input terminal and the ground terminal is coupled to the output terminal via a second path which the first flying capacitor and the inductor coupled to the output terminal; wherein in the second state the ground connection is coupled to the output connection via a third path comprising the first flying capacitor, the second flying capacitor and the inductor. The procedure according to Claim 17 wherein the first sequence has a demagnetization state in which the ground connection is coupled to the output connection via a demagnetization path having the inductor. The procedure according to Claim 17 or 18th comprising operating the converter in a second mode associated with a second range of conversion ratios by actuating the network of switches with a second sequence of states, the second sequence of states the first state, the second state, and a third state, wherein in the third state the input terminal is coupled to the output terminal via a path including the first flying capacitor, and wherein the ground terminal is coupled to the output terminal via a path including the second flying capacitor and the inductor . The procedure according to Claim 19 wherein in the third state, the input terminal is coupled to the output terminal via a path including the first flying capacitor and the inductor. The procedure according to Claim 19 or 20th comprising operating the converter in a third mode associated with a third range of conversion ratios by driving the network of switches with a third sequence of states, the third sequence of states including the first state and the third state. The procedure according to Claim 21 , wherein the third sequence has a magnetization state in which the input connection is coupled to the output connection via a magnetization path having the inductor.

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