CZ308051B6 - Low frequency signals amplifier - Google Patents

Abstract

The low-frequency signals amplifier consists of an input differential stage, a voltage amplifier connected to it and a connected output stage. The voltage amplifier is connected as a crank cascade composed of transistors (T9) and (T11), supplied by a current source as a current mirror composed of two identical transistors (T10) and (T12). The supply current is defined by the resistance values (R21 to R26). Linking the reference point between resistors (R22) and (R23) simultaneously introduces local negative feedback to linearize the cascade response to the excitation signal.

Description

Amplifier of low frequency signals

Field of technology

The invention relates to an amplifier of low-frequency signals with a voltage amplifier which is connected as a folded cascode.

Current state of the art

Since the technology of this type has been developed for more than a century, there are a huge number of solutions for low-frequency amplifiers. There are different directions of development. Some rely on an integrated solution in the form of an operational amplifier assembled on a chip, other approaches are based on the concept of an operational amplifier assembled from discrete components. Here, the designer has the opportunity to influence a significantly larger number of parameters, not only by choosing a suitable circuit topology, but also by choosing the components used, including semiconductors.

At present, a more or less stable topology is used in the design of low-frequency signal amplifiers, examples of which are shown in the diagrams in Fig. 1 a) to c). Typical examples of coupling solutions between the differential stage (transistors Tri to Tr3) and the following voltage amplifier (transistor T4) are presented here. Variant a) represents a differential stage with a symmetrical load on the collectors of transistors Tr2 and Tr3, variant b) represents a solution with omitted collector resistance of transistor Tr3, and finally variant c) represents a solution with a current mirror that contributes to the symmetrization of quiescent currents of the differential stage transistors. However, in all cases, unlike the proposed solution, the base of transistor Tr4 is connected to the collector of transistor Tr2.

Fig. 2 a) to f) shows 6 diagrams of examples of voltage amplifier solutions, which are commonly used in the construction of low-frequency amplifiers:

a) The amplifier consists of one transistor powered by a current source. The dominant pole of the transfer function is defined by the capacity Cdom. The output is followed by the final stage of the amplifier. This is the simplest solution of this part of the amplifier of low-frequency signals.

b) Here, compared to variant a), the current source is replaced by a simple resistor. In order to achieve greater voltage amplification, positive feedback from the amplifier output (so-called bootstrap) is introduced.

c) The solution presented as variant a) can be extended by adding an emitter follower. The resulting Darlington connection contributes to an increase in gain.

d) A cascode made up of two NPN bipolar transistors is used here. This solution further increases the effective collector resistance of the amplifying transistor, which also contributes to increasing the gain of the amplifier. However, this is a different cascode solution than the one that is the subject of the invention application.

e) This variant uses a voltage follower added to connection a) in order to impedance match the output of the amplifier to the next stage.

f) An alternative to case e) reducing the number of current sources in the circuit.

The essence of the invention

The aim of the construction (wiring) of the amplifier of low-frequency signals according to the invention was to create such a topology that would enable the implementation of sufficiently

- 1 CZ 308051 B6 high-quality low-frequency amplifier, capable of competing with expensive operational amplifiers built on the basis of an integrated circuit. The aim of the solution was also to optimize the amplifier for processing audio signals with maximum use of local feedback, the size of which cannot be influenced by the designer who uses the finished monolithic operational amplifier.

The amplifier of low-frequency signals according to the invention consists of an input differential stage, a voltage amplifier connected to it and an output stage connected to it.

The essence of the invention is that the voltage amplifier (see Fig. 3) is connected as a twisted cascode made of transistors T9 and Til, which is fed by a current source realized as a current mirror made of two identical transistors TIP and T12. The magnitude of the supplied current is defined values of resistors R21 to R26. Binding the reference point between resistors R22 and R23 also introduces local negative feedback to linearize the cascode response to the drive signal.

The basis of the input differential stage (see Fig. 3) can be transistors T2 and T8. The input of the amplifier is at the base of transistor T2, while global negative feedback is introduced into the base of transistor T8, determining the overall gain of the circuit and stabilizing the quiescent voltages in the circuit. The input differential stage operates with a quiescent current defined by a Widlar current mirror consisting of transistors TI, T5, T6. The value of the quiescent current in the input differential stage is preferably defined by the ratio of the resistances of the resistor R9 and the sum of the resistances of the resistors RI and R4. To evenly distribute the quiescent currents of transistors T2 and T8, their collectors can be loaded with a current mirror consisting of transistors T4 and T7, which must have the same magnitude of the factor β.

The final stage (see Fig. 3) consisting of transistors T13 to T15 can be an emitter follower driven by transistor T14, with transistors T13 and T15 operating as a constant current load. Capacitor C14 advantageously establishes a coupling between the collector of transistor T14 and the base of transistor T15 for alternating signals to partially modulate the quiescent current of the output stage in rhythm with the excitation voltage.

The circuit can be implemented at low cost, because thanks to the internal linearization by local feedback, even the use of lower quality components is fully functional.

Of course, it would be possible to integrate this structure on the chip as well, it does not necessarily have to be assembled only from discrete components.

Clarification of drawings

For a more detailed explanation of the essence of the invention, the attached drawings are used, where Fig. 1 - exemplary diagrams of a solution for connecting a voltage amplifier to a differential stage, Fig. 2 - diagrams of typical current solutions of a voltage amplifier in the structures of operational amplifiers, Fig. 3 - a diagram of the basic layout of a low-frequency signal amplifier according to the invention, Fig. 4 - diagram of an application example - two low-frequency amplifiers according to the invention form a preamplifier block for a magnetodynamic portable.

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Examples of implementation of the invention

Example 1

An exemplary circuit diagram of the amplifier of low-frequency signals according to the invention is shown in Fig. 3. This is a basic building block enabling the amplification of an analog low-frequency signal in the range of up to 40 dB (taking into account the gain of the amplifier with disconnected feedback). The input resistance of this amplifier can be changed by choosing the value of resistor R19 in the range of tens of kQ.

The input differential stage operates with a quiescent current of approximately 500 μΑ. This current is defined by a Widlar current mirror consisting of transistors TI, T5 and T6. At a given supply voltage (typically 24 V), the size of the quiescent current is defined by the ratio of the resistances of the resistor R9 and the sum of the resistances of the resistors RI and R4. By a suitable combination of the values of the mentioned resistors, it can be achieved that the amount of quiescent current does not depend on the temperature of the transistors. The ratio of resistances of resistors RI and R4 also defines the bias voltage at the input of the amplifier (transistor T2), which should correspond to half of the supply voltage (typically 12 V). Resistor R19 introduces positive feedback from the output to the input, increasing the input impedance of the circuit. The basis of the differential stage are transistors T2 and T8. The input of the amplifier is at the base of transistor T2, while global negative feedback is introduced into the base of transistor T8, which determines the overall gain of the circuit and stabilizes the quiescent voltages in the circuit. In order for the quiescent currents of transistors T2 and T8 to be evenly distributed, their collectors are loaded with a current mirror consisting of transistors T4 and T7, which must have the same magnitude of the factor β. The linearization of the differential stage is realized by local feedback made up of resistors R8 and R10.

The output of the differential stage is solved as a current mirror consisting of transistors T3 and T9. Through simulations, it was found that this solution contributes to the elimination of harmonic distortion of the amplifier. In addition, it is thus possible to achieve a small dependence of the quiescent currents of the transistors on the temperature. The amount of quiescent current flowing through transistor T9 is defined by resistor R20. At the same time, it introduces local feedback reducing the overall gain of the voltage amplification stage. For AC currents, this feedback is reduced to the required level by R18 + C9.

The voltage amplifier is designed as a cascode composed of transistors T9 and Til. This cascode is powered by a current source of approx. 1.2 mA, which is realized as a current mirror assembled from two identical transistors TIP and T12. The magnitude of the supplied current is defined by the values of resistors R21 to R26. These are chosen with regard to the temperature stability of the quiescent current and the fact that the binding of the reference point between resistors R22 and R23 introduces at the same time a local negative feedback that linearizes the response of the cascode to the excitation signal. This solution makes it possible to achieve good frequency and phase characteristics in the entire range of acoustic signals. A typical cut-off frequency of the amplifier is 250 kHz with a gain of approx. 33 dB.

The Nyquist stability condition of the amplifier with global feedback is satisfied by the use of capacitor C1, which defines the dominant pole of the amplifier's transfer function, and capacitor C12, which reduces the effect of local feedback in the cascode in the region of the highest frequencies, which partially compensates for the phase shift caused by the capacitances of the individual transistors. Another intervention in the resulting transfer function is realized by the compensating RC element R14 + C6. The latter appropriately suppresses the effect of global feedback at high frequencies. The amount of global feedback is given by resistors R17 and R15, while for direct voltage the effect of resistor R15 is eliminated by including capacitor C7. This contributes to the stabilization of the working voltages of individual transistors. The base of transistor Til is clamped to a constant bias voltage defined by resistors R27 and R28, which is about 20 V at a supply voltage of 24 V. This value was chosen with regard to the fact that the voltage amplification stage achieves the greatest possible value of voltage swing. The output voltage swing is greater than 16 V at a supply voltage of 24 V.

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The output stage consisting of transistors T13 to T15 is an emitter follower driven by transistor T14, with transistors T13 and T15 operating as constant current loads. The size of the quiescent current together with the size of the amplitude of the output signal indicate the size of the load that the amplifier can drive. For transistors in the TO92 case, the quiescent current can be selected in the order of tens of mA. The capacitor C14 establishes a coupling between the collector of the transistor T14 and the base of the transistor T15 for alternating signals, which makes it possible to partially modulate the quiescent current of the output stage in rhythm with the excitation voltage. This solution contributes to reducing the harmonic distortion of the output stage.

Capacitors C2, C3, C4, C5, C8, CIO, C13, C15 block the voltage in the respective nodes and thus eliminate interference penetrating the amplifier after the supply voltage is applied.

The advantage of this connection of the amplifier of low-frequency signals is the low cost of implementation, when good results can be achieved even with very cheap transistors. The individual blocks are linearized by local feedback, which makes it possible to achieve low harmonic distortion and a good response to excitation signals of an impulse character, and the final stage assembled from transistors T13, T14 and T15 differs from the usual structures implemented on the chip in that it works with a high quiescent current ( class A), which eliminates the occurrence of transient distortion and, depending on the gain set by the global feedback, a bandwidth of the order of hundreds of kHz is achieved. Typical values achieved with BC549 transistors are 250kHz bandwidth at 33dB gain.

Example 2

By connecting two amplifiers according to the diagram shown in Fig. 3 and supplementing them with a RIAA correction element, a preamplifier for a magnetodynamic portable was assembled. An illustration of the overall wiring diagram of the preamplifier for a magnetodynamic portable using two separate blocks of the amplifier of low-frequency signals according to the invention is shown in Fig. 4.

Claims (22)

1. An amplifier of low-frequency signals, consisting of an input differential stage, a voltage amplifier connected to it and an output stage connected to it, characterized by the fact that the voltage amplifier is connected as a twisted cascode made up of transistors (T9) and (Til), which is powered by a source current implemented as a current mirror made up of two identical transistors (T10) and (T12), with the fact that the magnitude of the supplied current is defined by the values of the resistors (R21 to R26) and that the binding of the reference point between the resistors (R22) and (R23) introduces at the same time a local negative feedback to linearize the cascode response to the drive signal. 3 drawings List of relationship tags Element in Fig. 3 Electrical component The value of the component in the base. implementation according to Fig. 3 Note Cl Electrolytic capacitor 10pF / 25V C2 Ceramic capacitor 33nF/50V C3 Electrolytic capacitor 47pF / 16V C4 Electrolytic capacitor 22pF/50V -4CZ 308051 B6 C5 Ceramic capacitor 33nF/50V C6 Ceramic capacitor 1nF/50V 1) C7 Electrolytic capacitor 470pF / 16V 2) C8 Electrolytic capacitor 47pF / 50V C9 Electrolytic capacitor 470pF / 10V 3) CIO Ceramic capacitor 33nF/50V Cll Ceramic capacitor 470pF / 50V 4) C12 Ceramic capacitor 1nF/50V 5) C13 Electrolytic capacitor 10pF / 25V 6) C14 Electrolytic capacitor 470pF / 25V C15 Electrolytic capacitor 100pF / 25V RI Metal oxide resistor 0.5 W 10 kΩ 7) R2 Metal oxide resistor 0.5 W 10 kΩ R3 Metal oxide resistor 0.5 W 10 kΩ R4 Metal oxide resistor 0.5 W 12 kΩ R5 Metal oxide resistor 0.5 W 3.3 kΩ R6 Metal oxide resistor 0.5 W 22 kΩ 8) R7 Metal oxide resistor 0.5 W 3.3 kΩ 8) R8 Metal oxide resistor 0.5 W 33 kΩ 8) R9 Metal oxide resistor 0.5 W 33 Ω RIO Metal oxide resistor 0.5 W 33 kΩ 8) Rll Metal oxide resistor 0.5 W 3.3 kΩ R12 Metal oxide resistor 0.5 W 22 kΩ 8) R13 Metal oxide resistor 0.5 W 3.3 kΩ 8) R14 Metal oxide resistor 0.5 W 100 Ω 9) R15 Metal oxide resistor 0.5 W 470 Ω 10) R16 Metal oxide resistor 0.5 W 330 Ω 11) R17 Metal oxide resistor 0.5 W 22 kΩ R18 Metal oxide resistor 0.5 W 220 Ω 12) R19 Metal oxide resistor 0.5 W 220 kΩ 13) R20 Metal oxide resistor 0.5 W 1.5 kΩ 14) R21 Metal oxide resistor 0.5 W 470 Ω R22 Metal oxide resistor 0.5 W 10 kΩ R23 Metal oxide resistor 0.5 W 2.2 kΩ R24 Metal oxide resistor 0.5 W 47 Ω R25 Metal oxide resistor 0.5 W 2.2 kΩ R26 Metal oxide resistor 0.5 W 47 kΩ 15) R27 Metal oxide resistor 0.5 W 10 kΩ R28 Metal oxide resistor 0.5 W 47 kΩ R29 Metal oxide resistor 0.5 W 47 kΩ 16) R30 Metal oxide resistor 0.5 W 4.7 kΩ R31 Metal oxide resistor 0.5 W 10 Ω R32 Metal oxide resistor 0.5 W 1 kΩ R33 Metal oxide resistor 0.5 W 470 Ω R34 Metal oxide resistor 0.5 W 150 Ω 17) T1 Bipolar transistor BC 560 18) T2 Bipolar transistor BC 560 19) T3 Bipolar transistor BC 550 20) T4 Bipolar transistor BC 550 21) T5 Bipolar transistor BC 560 18) -5 CZ 308051 B6 T6 Bipolar transistor BC 560 18) T7 Bipolar transistor BC 550 21) T8 Bipolar transistor BC 560 19) T9 Bipolar transistor BC 550 20) T10 Bipolar transistor BC 560 22) Till Bipolar transistor BC 557 T12 Bipolar transistor BC 557 22) T13 Bipolar transistor 2N3904 T14 Bipolar transistor 2N3904 T15 Bipolar transistor 2N3904 Length Silicon diode 1N4148 XI Input connector According to need 1) Phase delay compensation when total feedback is introduced. Adjust according to the gain setting so that the frequency characteristic is balanced. 2) Global feedback blocking for DC voltages (stabilization of operating point). Affects the lower cutoff frequency. 3) For nf signals, it eliminates the feedback introduced by resistor R20. 4) Defines the dominant transmission pole of the voltage amplification stage. 5) Frequency compensation of the cascode of the voltage amplification stage. 6) Til transistor base bias blocking, it should have a low ESR value. 7) Together with R9, it determines the temperature compensation of the Widlar current source and the size of the quiescent current by the differential stage. 8) They define the local feedback linearizing the differential stage. 9) Global feedback phase compensation component. 10) Together with R17, it determines the amount of global feedback. 11) To reduce sensitivity to interference in the power distribution, a suitable choke can be connected in series. 12) Defines the size of the cascode feedback loop of the voltage amplifier stage. 13) Introduces positive feedback to increase the input impedance. Allows you to adjust the input impedance to the signal source. 14) Defines the basic setting of the quiescent working current by the cascade of the voltage amplification stage. 15) Introduces local feedback for the entire cascode block of the voltage amplification stage (transistors T9, T10, Til, T12). 16) Defines the auxiliary current necessary for the correct operation of the current source of the final stage. The temperature stability of the quiescent current through the output stage depends on the setting of this current. 17) Defines the size of the quiescent current of the output stage. 18) Match transistors TI, T5, T6 to the same gain factor β. 19) Pair transistors T2, T8 to the same amplification factor β. 20) Pair transistors T3, T9 to the same gain factor β. 21) Pair transistors T4, T7 to the same amplification factor β. 22) Match transistors TI0, TI2 to the same amplification factor β.