CN212677086U - LLC resonant converter and control circuit thereof - Google Patents

Abstract

The application discloses an LLC resonant converter and a control circuit thereof. The LLC resonant converter comprises a control circuit, a first transformer and a second transformer, wherein a first driving winding and a second driving winding of the second transformer are magnetically coupled with a load winding, and the control circuit is used for controlling the conduction state of a path between a homonymous terminal and a heteronymous terminal of the first driving winding, so that the conduction state of a first bipolar transistor and a second bipolar transistor is controlled to adjust the resonant frequency. The control circuit of the LLC resonant converter can realize consistent turn-off time of the two bipolar transistors and reduce the size of a control switch tube.

Description

LLC resonant converter and control circuit thereof

Technical Field

The utility model relates to a power technical field, more specifically relates to LLC resonant converter and control circuit thereof.

Background

The resonant type switching converter is a power converter which obtains square wave voltage by adopting a control switching tube and realizes energy transmission by adopting a resonant circuit for resonance. The LLC resonant converter is a third-order resonant network consisting of an inductor and a capacitor, and can realize the regulation of a load from full load to no load in a narrow frequency range. The LLC resonant converter has a high power density and a small number of electronic components, has a smooth Current waveform, is advantageous to improve electromagnetic interference, and can realize Zero Voltage Switching (ZVS) and Zero Current Switching (ZCS) of the control Switching tube in the entire operating range, which is conducive to obtaining extremely high efficiency, and thus is widely used.

The LLC resonant converter comprises a first power tube and a second power tube which are alternately switched on and off, and converts a direct current input voltage into a square wave voltage. The square wave voltage is input to the resonant tank to generate a resonant current at the resonant frequency. The first power transistor and the second power transistor are, for example, MOSFETs (metal oxide semiconductor field effect transistors) or bipolar transistors. The MOSFET is used as a power transistor to obtain excellent switching performance, but the control circuit is complicated, which results in high circuit cost. The bipolar transistor is adopted as the power tube, so that the circuit cost can be obviously reduced.

In an LLC resonant Converter, a Self-Oscillating driver (SOC) is implemented by obtaining a drive current of a bipolar transistor using, for example, a transformer coupled to a resonant tank. In order to regulate the switching frequency of the bipolar transistors, a control circuit is used to provide an additional path between the base-emitter of one bipolar transistor, thereby forcing an early switching of the on and off states of the bipolar transistor, so that the LLC resonant converter can regulate the resonant current to obtain the desired dc output voltage/current.

However, in the above-described conventional LLC resonant converter, the control circuit provides a path for only one bipolar transistor, the other bipolar transistor being switched off by magnetic coupling of the two drive windings of the transformer. Therefore, the turn-off time of the two bipolar transistors is not consistent, which causes the asymmetry problem of the system and further causes the magnetic bias problem of the current transformer. In addition, the bipolar transistor works in a saturation region in an on state, and only exits the saturation region when current is extracted from the base of the bipolar transistor in a short-circuit mode to control the turn-off of the bipolar transistor. In order to achieve fast turn-off, the on-resistance of the control switch tube needs to be small to obtain a large current, resulting in a correspondingly large size of the control switch tube.

It is desirable to further improve the control scheme of LLC resonant converters to achieve consistent off-times of the two bipolar transistors and to reduce the size of the control switch tube.

SUMMERY OF THE UTILITY MODEL

In view of the above problems, the present application provides an LLC resonant converter and a control circuit thereof, wherein the control circuit is configured to control the conduction state of the path between the dotted terminal and the dotted terminal of the first driving winding to control the conduction states of the two bipolar transistors, so as to achieve consistent turn-off time of the two bipolar transistors and reduce the size of the control switch tube.

According to the utility model discloses an aspect provides a LLC resonant converter, include: the first transformer comprises a primary winding and a secondary winding; a second transformer comprising a load winding, and first and second drive windings magnetically coupled to the load winding; and a first bipolar transistor and a second bipolar transistor connected in series between a positive power supply end and a negative power supply end of a direct current input voltage, wherein a load winding, a resonant element and a primary winding of the second transformer are connected between a middle node of the first bipolar transistor and the second bipolar transistor and the negative power supply end, the resonant element and the primary winding are connected to form a resonant circuit to generate a resonant current, and a resonant output voltage is provided at two ends of the primary winding.

Preferably, the first driving winding includes a homonymous terminal, a synonym terminal, and a tap terminal, the second driving winding includes a homonymous terminal and a synonym terminal, a base of the first bipolar transistor is connected to the tap terminal of the first driving winding to receive a first driving current generated according to an induced current of the resonant current, and a base of the second bipolar transistor is connected to the synonym terminal of the second driving winding to receive a second driving current generated according to an induced current of the resonant current.

Preferably, the first bipolar transistor and the second bipolar transistor are NPN bipolar transistors, an emitter of the first bipolar transistor and a collector of the second bipolar transistor are commonly connected to the intermediate node, the different-name end of the first driving winding is connected to the emitter of the first bipolar transistor, and the same-name end of the second driving winding is connected to the emitter of the second bipolar transistor.

Preferably, the method further comprises the following steps: the control circuit generates a clock signal according to a current sampling signal of the resonant current and/or a voltage feedback signal of the resonant output voltage, and generates a first gate drive signal and a second gate drive signal according to the clock signal, so that the first control switch tube and the second control switch tube are controlled to periodically short-circuit the homonymous end and the heteronymous end of the first drive winding, and the resonant period of the LLC resonant converter follows the clock signal.

Preferably, the first control switch tube and the second control switch tube are respectively N-type MOSFETs, a drain of the first control switch tube and a drain of the second control switch tube are connected to each other, a source of the first control switch tube is connected to a dotted terminal of the first driving winding, a source of the second control switch tube is connected to a dotted terminal of the first driving winding, and a source of the second control switch tube is grounded.

Preferably, the gate of the first control switch tube receives a first gate driving signal, and the gate of the second control switch tube receives a second gate driving signal.

Preferably, the first control switch tube and the second control switch tube are respectively N-type MOSFETs, a source of the first control switch tube and a source of the second control switch tube are connected to each other, a drain of the first control switch tube is connected to a dotted terminal of the first drive winding, a drain of the second control switch tube is connected to a dotted terminal of the first drive winding, and a middle node between the first control switch tube and the second control switch tube is grounded.

Preferably, the gate of the first control switch tube receives a first gate driving signal, and the gate of the second control switch tube receives a second gate driving signal.

Preferably, the method further comprises the following steps: and a sampling resistor connected in series with the primary winding of the first transformer, and having one end connected to a middle node of the first bipolar transistor and the second bipolar transistor, wherein a current sampling signal of the resonant current is obtained at both ends of the sampling resistor.

Preferably, the control circuit includes: the clock signal generating module is used for generating the clock signal with a corresponding clock period according to a current sampling signal of the resonant current and/or a voltage feedback signal of the resonant output voltage; the logic module is connected with the clock signal generating module and generates a first control signal and a second control signal according to the clock signal; the first driving module is connected with the logic module and generates the first grid driving signal according to the first control signal; and the second driving module is connected with the logic module and generates the second grid driving signal according to the second control signal.

Preferably, the control circuit further comprises: a first comparator to compare a current sampling signal of the resonant current with a first current threshold to obtain a first comparison signal; and a second comparator, configured to compare a current sampling signal of the resonant current with a second current threshold to obtain a second comparison signal, wherein the logic module is connected to the first comparator to obtain the first comparison signal, and is connected to the second comparator to obtain the second comparison signal, and the logic module controls the first gate driving signal to be in an active state when the clock signal is invalid or the current sampling signal is smaller than the first current threshold, and controls the second gate driving signal to be in an active state when the clock signal is valid or the current sampling signal is larger than the second current threshold.

Preferably, the control circuit further comprises: the first current source and the switch tube are connected in series between a power supply end and a same-name end of the first driving winding; and a one-shot module, a first input end of which is connected to the first comparator to obtain the first comparison signal, a second input end of which is connected to the second comparator to obtain the second comparison signal, and an output end of which is connected to the switch tube to provide a start signal, wherein during start-up of the LLC resonant converter, the current sampling signal is greater than the first current threshold and less than the second current threshold, the start signal is valid, so that the switch tube is turned on, the first current source injects excitation current to the first drive winding via the switch tube, and the first control switch tube and the second control switch tube are turned off.

Preferably, the logic module comprises: a NOT gate for obtaining an inverted signal of the clock signal; a first or gate, a first input terminal receiving an inverted signal of the clock signal, a second input terminal receiving the first comparison signal; the first input end of the first AND gate is connected to the output end of the first OR gate, and the second input end of the first AND gate receives the starting signal; a second or gate, the first input terminal receiving the clock signal, the second input terminal receiving the second comparison signal; and a second and gate, a first input terminal connected to an output terminal of the second or gate, a second input terminal receiving the enable signal, wherein the enable signal is active at low level.

Preferably, the first driving module is connected to a homonymous terminal of the first driving winding, and performs level shift on the first control signal with respect to the homonymous terminal to obtain the first gate driving signal, and the second driving module is connected to a synonym terminal of the first driving winding, and performs level shift on the second control signal with respect to the synonym terminal to obtain the second gate driving signal.

Preferably, the clock signal generating module includes: the circuit comprises a compensation module and a first capacitor connected to the output end of the compensation module, wherein compensation signals are generated at two ends of the first capacitor; the oscillator generates an oscillation signal with corresponding frequency according to the compensation signal; and a frequency division module generating the clock signal with a 50% duty ratio according to the oscillation signal, wherein the compensation module compares the current sampling signal with a current reference signal and/or compares the voltage feedback signal with a voltage reference signal to generate the compensation signal.

Preferably, the control circuit and the first and second control switch tubes are integrated into a single chip.

According to a second aspect of the present invention, there is provided a control circuit for an LLC resonant converter, said LLC resonant converter comprising a first drive winding coupled with a resonant tank, said control circuit comprising: the clock signal generating module generates a clock signal with a corresponding clock period according to a current sampling signal of the resonant current and/or a voltage feedback signal of the resonant output voltage; the logic module is connected with the clock signal generating module and generates a first control signal and a second control signal according to the clock signal; the first driving module is connected with the logic module and generates a first grid driving signal according to the first control signal; and the second driving module is connected with the logic module and generates a second gate driving signal according to the second control signal, wherein the control circuit controls the first control switching tube and the second control switching tube to periodically short-circuit the homonymous end and the synonym end of the first driving winding by adopting the first gate driving signal and the second gate driving signal, and generates a driving current at the tap end of the first driving winding, so that the resonance period of the LLC resonant converter follows the clock signal.

Preferably, the method further comprises the following steps: a first comparator to compare a current sampling signal of the resonant current with a first current threshold to obtain a first comparison signal; and a second comparator, configured to compare a current sampling signal of the resonant current with a second current threshold to obtain a second comparison signal, wherein the logic module is connected to the first comparator to obtain the first comparison signal, and is connected to the second comparator to obtain the second comparison signal, and the logic module controls the first gate driving signal to be in an active state when the clock signal is invalid or the current sampling signal is smaller than the first current threshold, and controls the second gate driving signal to be in an active state when the clock signal is valid or the current sampling signal is larger than the second current threshold.

Preferably, the method further comprises the following steps: the first current source and the switch tube are connected in series between a power supply end and a same-name end of the first driving winding; and a one-shot module, a first input end of which is connected to the first comparator to obtain the first comparison signal, a second input end of which is connected to the second comparator to obtain the second comparison signal, and an output end of which is connected to the switch tube to provide a start signal, wherein during start-up of the LLC resonant converter, the current sampling signal is greater than the first current threshold and less than the second current threshold, the start signal is valid, so that the switch tube is turned on, the first current source injects excitation current to the first drive winding via the switch tube, and the first control switch tube and the second control switch tube are turned off.

Preferably, the logic module comprises: a NOT gate for obtaining an inverted signal of the clock signal; a first or gate, a first input terminal receiving an inverted signal of the clock signal, a second input terminal receiving the first comparison signal; the first input end of the first AND gate is connected to the output end of the first OR gate, and the second input end of the first AND gate receives the starting signal; a second or gate, the first input terminal receiving the clock signal, the second input terminal receiving the second comparison signal; and a second and gate, a first input terminal connected to an output terminal of the second or gate, a second input terminal receiving the enable signal, wherein the enable signal is active at low level.

Preferably, the first driving module is connected to a homonymous terminal of the first driving winding, and performs level shift on the first control signal with respect to the homonymous terminal to obtain the first gate driving signal, and the second driving module is connected to a synonym terminal of the first driving winding, and performs level shift on the second control signal with respect to the synonym terminal to obtain the second gate driving signal.

Preferably, the clock signal generating module includes: the circuit comprises a compensation module and a first capacitor connected to the output end of the compensation module, wherein compensation signals are generated at two ends of the first capacitor; the oscillator generates an oscillation signal with corresponding frequency according to the compensation signal; and a frequency division module that generates the clock signal with a 50% duty cycle from the oscillation signal, wherein the compensation module compares the current sampling signal with a current reference signal and/or compares the voltage feedback signal with a voltage reference signal to generate the compensation signal such that the frequency of the clock signal is adjusted in relation to the resonant current and/or the resonant output voltage.

Preferably, the first control switch tube and the second control switch tube are respectively N-type MOSFETs, a drain of the first control switch tube and a drain of the second control switch tube are connected to each other, a source of the first control switch tube is connected to a dotted terminal of the first drive winding, a source of the second control switch tube is connected to a dotted terminal of the first drive winding, and a source of the second control switch tube is grounded, wherein a gate of the first control switch tube receives a first gate drive signal, and a gate of the second control switch tube receives a second gate drive signal.

Preferably, the first control switch tube and the second control switch tube are respectively N-type MOSFETs, a source of the first control switch tube and a source of the second control switch tube are connected to each other, a drain of the first control switch tube is connected to a dotted terminal of the first drive winding, a drain of the second control switch tube is connected to a dotted terminal of the first drive winding, and a middle node of the first control switch tube and the second control switch tube is grounded, wherein a gate of the first control switch tube receives a first gate drive signal, and a gate of the second control switch tube receives a second gate drive signal.

Preferably, the control circuit and the first and second control switch tubes are integrated into a single chip.

According to the utility model discloses LLC resonant converter, control circuit control the on-state of the route between the dotted terminal and the synonym terminal of first drive winding to control first bipolar transistor with the on-state of second bipolar transistor is in order to adjust resonant frequency. The LLC resonant converter can realize consistent turn-off time of two bipolar transistors and reduce the size of a control switch tube.

In a preferred embodiment, the first bipolar transistor obtains a first driving current generated according to an induced current of the resonant current from a tap terminal of the first driving winding, and the second bipolar transistor obtains a second driving current generated according to an induced current of the resonant current from a synonym terminal of the second driving winding, so as to implement Self-Oscillating driving (SOC). Under the control of the self-oscillation driving signal, the first bipolar transistor and the second bipolar transistor are alternately turned on and off to convert the direct-current input voltage into a square-wave voltage. The square wave voltage is input to the resonant tank to generate a resonant current at the resonant frequency. Thus, the electrical energy is transferred from the primary side of the first transformer to the secondary side of the first transformer through the resonant tank. The LLC resonant converter generates the driving currents of the first bipolar transistor and the second bipolar transistor based on the induction current of the first driving winding, and the consistency of the turn-off time of the two bipolar transistors can be further improved.

In a preferred embodiment, the control circuit generates a compensation signal according to a current sampling signal of the resonant current and/or a voltage feedback signal of the resonant output voltage to adjust a clock period of the clock signal, so as to control switching periods of the first control switch tube and the second control switch tube, so as to provide an additional path and an excitation current for the first driving winding, and further controls conducting states of the first bipolar transistor and the second bipolar transistor through magnetic coupling of the first driving winding and the second driving winding, so as to realize consistent turn-off time of the two bipolar transistors, thereby adjusting the resonant frequency. Therefore, the LLC resonant converter can adjust the resonant current in a clock period adjusting mode to obtain the expected DC output voltage/current, thereby realizing constant current or constant voltage control, simplifying a control circuit and reducing the circuit cost.

Drawings

The above and other objects, features and advantages of the present invention will become more apparent from the following description of the embodiments of the present invention with reference to the accompanying drawings, in which:

fig. 1 shows a schematic circuit diagram of an LLC resonant converter according to the prior art.

Fig. 2 shows a schematic circuit diagram of an LLC resonant converter according to a first embodiment of the invention.

Fig. 3 shows a schematic circuit diagram of a control circuit in the LLC resonant converter shown in fig. 2.

Fig. 4 shows an equivalent circuit diagram of the LLC resonant converter shown in fig. 2 during start-up.

Fig. 5 shows a waveform diagram of the LLC resonant converter shown in fig. 2 operating in a self-oscillation mode.

Fig. 6a to 6j show equivalent circuit diagrams of the LLC resonant converter shown in fig. 2 at different stages of operation in the self-oscillation mode.

Fig. 7 shows a schematic circuit diagram of an LLC resonant converter according to a second embodiment of the invention.

Fig. 8 shows a schematic circuit diagram of a control circuit in the LLC resonant converter shown in fig. 7.

Detailed Description

Various embodiments of the present invention will be described in more detail below with reference to the accompanying drawings. Like elements in the various figures are denoted by the same or similar reference numerals. For purposes of clarity, the various features in the drawings are not necessarily drawn to scale.

Fig. 1 shows a schematic circuit diagram of an LLC resonant converter according to the prior art. The LLC resonant converter 10 includes a first transformer T1, a second transformer T2, bipolar transistors Q1 and Q2, diodes D1 and D2, a resonant capacitor Cr, an output capacitor Co, and a resonant inductor Lr.

On the primary side of the first transformer T1, the primary winding Lp of the first transformer T1, the resonant capacitor Cr, and the resonant inductor Lr form a resonant tank. Between the positive and negative input terminals of the LLC resonant converter 10, bipolar transistors Q1 and Q2 are connected in series, with the intermediate node of the two being connected to the resonant tank. The second transformer T2 includes three windings around the same core, i.e., a load winding W1, drive windings W2 and W3. In the resonant tank, the load winding W1 is connected in series with the primary winding Lp. Also, drive windings W2 and W3 are coupled to the bases of bipolar transistors Q1 and Q2, respectively, but in opposite directions. That is, the dotted terminal of drive winding W2 is connected to the base of bipolar transistor Q1, and the dotted terminal of drive winding W3 is connected to the base of bipolar transistor Q2.

On the secondary side of the first transformer T1, diodes D1 and D2 constitute a rectifying circuit. The two ends of the secondary winding are respectively connected with the anodes of the diodes D1 and D2, and the middle tap of the secondary winding Ls is grounded. An output capacitor Co is connected between the cathodes of diodes D1 and D2 and ground, and provides a dc output voltage across it to the load Rd.

The LLC resonant converter 10 is a self-oscillating LLC half-bridge driven topology. The winding of the second transformer T2 is used to provide a driving current to drive the bases of bipolar transistors Q1 and Q2 to achieve Self-Oscillating driving (SOC). Under the control of the self-oscillating drive signal, bipolar transistors Q1 and Q2 are alternately turned on and off, converting the dc input voltage into a square wave voltage. The square wave voltage is input to the resonant tank to generate a resonant current at the resonant frequency. Thus, electrical energy is transferred from the primary side of the first transformer T1 to the secondary side of the first transformer T1 through the resonant tank.

The operating frequency of the LLC resonant converter 10 according to the prior art is the natural SOC oscillation frequency and thus regulation of the dc output voltage/current cannot be achieved.

Fig. 2 shows a schematic circuit diagram of an LLC resonant converter according to a first embodiment of the invention. The LLC resonant converter 20 includes a first transformer T1, a second transformer T2, bipolar transistors Q1 and Q2, diodes D1 and D2, a resonant capacitor Cr, a resonant inductor Lr, control switching transistors M1 and M2, and a control circuit 100.

On the primary side of the first transformer T1, the primary winding Lp of the first transformer T1, the resonant capacitor Cr, and the resonant inductor Lr form a resonant tank. Between the positive and negative input terminals of the LLC resonant converter 20, bipolar transistors Q1 and Q2 are connected in series, with the intermediate node of the two being connected to the resonant tank. In the resonant tank, a sampling resistor Rs is connected in series with the primary winding Lp, so that a sampling signal for characterizing the inductor current flowing through the primary winding Lp can be obtained.

On the secondary side of the first transformer T1, diodes D1 and D2 constitute a rectifying circuit. Two ends of the secondary winding are respectively connected with anodes of diodes D1 and D2, and the middle tap of the secondary winding is grounded. An output capacitor Co is connected between the cathodes of diodes D1 and D2 and ground, and provides a dc output voltage across it to the load Rd.

The second transformer T2 includes three windings around the same core, i.e., a load winding W1, drive windings W2 and W3. In the resonant tank, the load winding W1 is connected in series with the primary winding Lp. Also, drive windings W2 and W3 are coupled to the bases of bipolar transistors Q1 and Q2, respectively, but in opposite directions. For example, drive winding W2 has a homonymous terminal, a synonym terminal, and a tap terminal therebetween, and drive winding W3 has a homonymous terminal and a synonym terminal. The dotted terminal DR1 of the drive winding W2 is connected to a current terminal of the control circuit 100 to obtain an excitation current I1.

The control switching tubes M1 and M2 are connected in series in opposite directions between the dotted terminal DR1 and the dotted terminal DR2 of the drive winding W2 of the second transformer T2. The control switch transistors M1 and M2 are, for example, N-type MOSFETs (Metal-Oxide-Semiconductor Field Effect transistors), respectively, and include anti-parallel body diodes, respectively. The drains of the control switch transistors M1 and M2 are connected to each other, the source of the control switch transistor M1 is connected to the dotted terminal DR1 of the drive winding W2, and the source of the control switch transistor M2 is connected to the dotted terminal DR2 of the drive winding W2.

The intermediate node of bipolar transistors Q1 and Q2 is connected to the synonym terminal DR2 of drive winding W2. Specifically, the base and emitter of bipolar transistor Q1 are connected to the tap and alias terminals DR2, respectively, of drive winding W2. The base and emitter of bipolar transistor Q2 are connected to the synonym and synonym terminals, respectively, of drive winding W3.

The turn ratio of the second transformer T2 is selected according to the electrical characteristics of the bipolar transistors Q1 and Q2. In this embodiment, the load winding W1 of the second transformer T2, the drive winding W2: the turn ratio of drive winding W3 is 1: 18: 6, wherein the tap end of the drive winding W2 divides the drive winding W2 into a first sub-winding between the dotted end DR1 and the tap end and a second sub-winding between the tap end and the unlike end DR2, and the first sub-winding: the turn ratio of the second sub-winding is 12: 6. in an alternative embodiment, the load winding W1 of the second transformer T2, the drive winding W2: the turn ratio of drive winding W3 is 1: 24: 6 or 2: 36: 12. the fixed turn ratio of the first sub-winding to the second sub-winding of drive winding W2 is such that the drive voltage of bipolar transistor Q1 is in a fixed proportional relationship with the terminal voltage of drive winding W2 (i.e., the voltage between the dotted terminal DR1 and the dotted terminal DR 2).

For example, in the state where only one of the control switches M1 and M2 is conducting, the terminal voltage of the driving winding W2 is the forward voltage drop of the body diode of the control switch, so that the terminal voltage of the driving winding W2 is 0.7V, and the driving signal clamp of the bipolar transistor Q1 is at 0.7/3V, so that the bipolar transistor Q1 maintains the off state.

The control circuit 100 obtains a current sampling signal CS of the resonant current from the sampling resistor Rs of the resonant converter 20, obtains a voltage feedback signal FB of the resonant output voltage from the auxiliary winding La of the first transformer T1 of the resonant converter 20, generates a gate driving signal VG1 for controlling the switching tube M1 and a gate driving signal VG2 for controlling the switching tube M2 according to the current sampling signal CS and the voltage feedback signal FB, and excites the current I1. In this embodiment, the ground terminal of the control circuit 100, the intermediate node of the bipolar transistors Q1 and Q2, and the dotted terminal of the auxiliary winding La are all grounded together.

Under the control of the control circuit 100, the control switch tubes M1 and M2 have three working modes, i.e., double-off, double-on, and single-on. During the start-up phase of the LLC resonant converter, the switching tubes M1 and M2 are controlled to be in a double-off state, and the drive winding W2 of the second transformer T2 of the resonant converter 20 injects an excitation current, thereby starting the self-oscillation mode of operation. In the self-oscillation working phase of the LLC resonant converter, the switching tubes M1 and M2 are controlled to be in a double-conduction state or a single-conduction state, so that the conduction states and the cut-off states of the bipolar transistors Q1 and Q2 are forced to be switched in advance, and the LLC resonant converter can obtain the expected direct current output voltage/current by adopting a mode of regulating resonant current.

According to the LLC resonant Converter 20 of this embodiment, the drive windings W2 and W3 are used to provide drive current to drive the bases of bipolar transistors Q1 and Q2 to achieve Self-Oscillating drive (SOC). Under the control of the self-oscillating drive signal, bipolar transistors Q1 and Q2 are alternately turned on and off, converting the dc input voltage into a square wave voltage. The square wave voltage is input to the resonant tank to generate a resonant current at the resonant frequency. Thus, electrical energy is transferred from the primary side of the first transformer T1 to the secondary side of the first transformer T1 through the resonant tank.

The control circuit 100 controls the conduction states of the control switch transistors M1 and M2 to provide an additional path and excitation current for the drive winding W2, and further controls the conduction states of the bipolar transistors Q1 and Q2 via magnetic coupling of the drive windings W2 and W3 to achieve consistent off times of the two bipolar transistors to adjust the resonant frequency. Therefore, the LLC resonant converter 20 can adjust the resonant frequency to obtain the desired dc output voltage/current, thereby achieving constant current control, simplifying the control circuit and reducing the circuit cost.

Fig. 3 shows a schematic circuit diagram of a control circuit in the LLC resonant converter shown in fig. 2. The control circuit 100 includes a clock signal generating module 110, a logic module 120, comparators a12 and a13, a one-shot module 104, driving modules 105 and 106, a current source Is1, and a switch M3. The clock signal generation module 110 includes a compensation module 101, an oscillator 111, a frequency division module 103, and a capacitor Cc. The logic module 120 includes NOT gates A14, OR gates A15 and A16, and gates A17 and A18.

The compensation module 101 receives the current sampling signal CS and/or the voltage feedback signal FB, and generates a compensation signal COMP at two ends of the capacitor Cc according to a comparison result between the current sampling signal CS and the current reference signal, and/or according to a comparison result between the voltage feedback signal FB and the voltage reference signal. In the constant-current control mode, if the direct-current output current is too small, the compensation signal COMP is increased, and if the direct-current output current is too large, the compensation signal COMP is decreased. In the constant voltage control mode, if the dc output voltage is too small, the compensation signal COMP increases, and if the dc output voltage is too large, the compensation signal COMP decreases.

The oscillator 111 further includes a comparator a11, a current source Is2, and a capacitor C11. The current source Is2 and the capacitor C11 are connected in series between the supply terminal and the ground terminal, and the intermediate node provides a triangular wave signal. The comparator a11 has a non-inverting input terminal receiving the triangular wave signal, an inverting input terminal receiving the compensation signal COMP, and an output terminal providing the oscillation signal OSC. The frequency of the oscillation signal OSC is related to the compensation signal COMP, and the frequency of the oscillation signal OSC is low when the compensation signal COMP is high, and the frequency of the oscillation signal OSC is high when the compensation signal COMP is low. The frequency divider module 103 has an input terminal receiving the oscillation signal OSC, and an output terminal providing the clock signal CLK. The frequency division module 103 generates a clock signal CLK with a 50% duty cycle according to the oscillation signal OSC.

The comparator a12 compares the current sample signal CS of the resonant current with a first current threshold (e.g., -100mV) to obtain a first comparison signal, and the comparator a13 compares the current sample signal CS of the resonant current with a second current threshold (e.g., +100mV) to obtain a second comparison signal. The one-shot module 104 determines whether the current sampling signal CS is in a predetermined current interval according to the first comparison signal and the second comparison signal, and generates a start signal Ts.

During start-up of the LLC resonant converter 20, the current sampling signal CS is in the predetermined current interval, and the start-up signal Ts is active at a low level.

In this embodiment, the control switches M1 and M2 are, for example, N-type MOSFETs, and the switch M3 is, for example, P-type MOSFETs. The enable signal Ts is asserted low, causing the switch M3 to conduct. Meanwhile, the enable signal Ts is used as one of the input signals of the and gates a17 and a18, so that the gate driving signals VG1 and VG2 provided by the driving modules 105 and 106 are disabled, and the control switch tubes M1 and M2 are turned off. A current source Is1 Is connected in series with switch M3 between the supply terminal and the dotted terminal DR1 of drive winding W2.

Therefore, during start-up of the LLC resonant converter 20, the current source Is1 injects the excitation current I1 via the switching tube M3 to the dotted terminal DR1 of the drive winding W2, so that the LLC resonant converter 20 starts to operate in a self-oscillating manner. An equivalent circuit diagram of the LLC resonant converter 20 during start-up is shown in fig. 4.

Further, after the LLC resonant converter 20 starts operating in the self-oscillation mode, the current sampling signal CS of the resonant current exceeds the above-mentioned predetermined current interval for a fixed time of a plurality of switching cycles, and therefore, the start signal Ts maintains the inactive state of the high level. Since the switching tube M3 Is turned off, the current source Is1 stops injecting the excitation current I1 to the dotted terminal DR1 of the drive winding W2 via the switching tube M3.

The control switch tubes M1 and M2 control their conducting states according to the clock signal CLK. As shown, a first input terminal of the or gate a15 is connected to the output terminal of the frequency dividing module 103 via an inverter a14 to receive an inverted signal of the clock signal CLK, and a second input terminal is connected to the output terminal of the comparator a12 to receive a first comparison signal; the or gate a16 has a first input connected to the output of the divider block 103 to receive the clock signal CLK and a second input connected to the output of the comparator a13 to receive the second comparison signal. The AND gate A17 has a first input connected to the output of the OR gate A15, a second input receiving the enable signal Ts, and an output connected to the driver module 105. The AND gate A18 has a first input coupled to the output of the OR gate A16, a second input receiving the enable signal Ts, and an output coupled to the driver module 106. Therefore, the control switch M1 is turned off only when the clock signal CLK is active and the current sampling signal CS is greater than the first current threshold, and is turned on otherwise. In the case that the clock signal CLK is inactive and the current sampling signal CS is smaller than the second current threshold, the control switch M2 is turned off, and the rest is turned on.

The driving modules 105 and 106 are used for generating a gate driving signal VG1 for controlling the switch tube M1 and a gate driving signal VG2 for controlling the switch tube M2. The driving modules 105 and 106 are, for example, level shift circuits, respectively. Further, the driving module 105 is connected to the dotted terminal DR1 of the driving winding W2, and level-shifts the output signal of the and gate a17 with respect to the dotted terminal DR1 to obtain the gate driving signal VG 1. The driving module 106 is connected to the dotted terminal DR2 of the driving winding W2, and level-shifts the control signal with respect to the dotted terminal DR2 to obtain the gate driving signal VG 2. Therefore, even in the case where a positive voltage or a negative voltage exists at the dotted terminal DR1 and the dotted terminal DR2 of the driving winding W2, the level shift circuit is used to provide the gate driving signal VG1 with an appropriate level between the gate and the source of the control switch M1 and to provide the gate driving signal VG2 with an appropriate level between the gate and the source of the control switch M2.

Fig. 5 shows a waveform diagram of the LLC resonant converter shown in fig. 2 operating in a self-oscillation mode. The graph shows the time-dependent change relationship among the resonant current CR obtained by the control circuit 100, the excitation current CT of the second transformer T2, and the clock signal CLK.

The resonant current CR is a current flowing through the primary winding Lp of the first transformer T1, and a current flowing to the same-name terminal of the primary winding Lp of the first transformer T1 is defined as a positive current in this application. The exciting current CT is an exciting current of the load winding W1 of the second transformer T2, and in the present application, the field current of the same-name terminal flowing to the load winding W1 of the second transformer T2 is defined as a positive current.

Further, the current sampling signal CS of the resonance current is a voltage signal obtained at the resistance Rs, and a voltage drop in the direction of the current flowing to the intermediate node of the bipolar transistors Q1 and Q2 is defined as a positive voltage, a negative voltage signal of the current sampling signal CS is obtained when the resonance current is a positive current, and a positive voltage signal of the current sampling signal CS is obtained when the resonance current is a negative current.

The resonant current CR periodically intersects the excitation current CT of the second transformer T2 at point A, B, C, D. The clock signal CLK has two states of high and low (1, 0), and the resonant current CR also has two states of positive and negative (>0, <0), which are combined in pairs to have four different states, thereby generating different circuit stages.

In the low level period of the clock signal CLK, the resonance current CR is converted from a negative current to a positive current. In the high level period of the clock signal CLK, the resonance current CR is converted from a positive current to a negative current. The resonant current CR presents a predetermined current interval defined by a first current threshold (e.g., -100mV) and a second current threshold (e.g., +100mV) around zero current. In the period in which the clock signal CLK is low (0), the resonance current CR sequentially passes through 3 circuit stages of a period t1 greater than the second current threshold, a period t2 within a predetermined current region, and a period t3 less than the first current threshold. In the period in which the clock signal CLK is high (1), the resonance current CR sequentially passes through 3 circuit stages of a period t4 smaller than the first current threshold, a period t5 within the predetermined current region, and a period t6 larger than the second current threshold. Thus, in each switching cycle of the LLC resonant converter, there are 6 circuit stages as shown in the following table.

TABLE 1 LLC resonant converter in different circuit phases of the switching cycle

Fig. 6a to 6c show equivalent circuit diagrams of the LLC resonant converter shown in fig. 2 in a first phase t 1.

In the first phase t1, the clock signal CLK is low, the resonant current CR is a negative current, and the current sampling signal CS of the resonant current is greater than 100 mV. As can be seen from the operation of the control circuit shown in fig. 3, the first comparison signal generated by the comparator a12 is inactive, the inverted signal generated by the clock signal CLK via the inverter a14 is active, and the gate driving signal VG1 generated according to the first comparison signal and the inverted signal of the clock signal CLK is active, so that the control switch M1 is in a conducting state. Meanwhile, the second comparison signal generated by the comparator a13 is active, the clock signal CLK is inactive, and the gate driving signal VG2 generated according to the second comparison signal and the clock signal CLK is active, so that the control switch transistor M2 is in a conducting state. Therefore, the control switches M1 and M2 are in a double-conduction state, and form a bidirectional path between the same-name terminal and the different-name terminal of the driving winding W2 of the second transformer T2 via the control switches M1 and M2.

As shown in fig. 6a, before point a, the resonance current CR and the excitation current CT are negative currents, respectively, and the absolute value of the resonance current CR is larger than the absolute value of the excitation current CT. The net current is the difference of the resonant current CR minus the excitation current CT, and the net current direction at this stage is still negative. The bipolar transistor Q1 is in the off state under the base-emitter bias voltage supplied from the drive winding W2 of the second transformer T2, and the bipolar transistor Q2 is brought from the saturated state to the off state under the base-emitter bias voltage supplied from the drive winding W3 of the second transformer T2. The induced current of the driving winding W2 of the second transformer T2 is a forward current, i.e. flows from the different-name terminal to the same-name terminal via the control switching tubes M1 and M2. Since the bipolar transistor Q2 is turned off, no induced current is generated in the drive winding W3 of the second transformer T2. Since the bipolar transistors Q1 and Q2 are both in the off state, the resonance current charges the capacitor Cm, and the node voltage Vmid gradually rises.

As shown in fig. 6b, as the node voltage Vmid rises, the PN junction of the base-collector of the bipolar transistor Q1 is turned on, and thus enters a reverse amplification state from the off state, and the bipolar transistor Q2 is maintained in the off state by the base-emitter bias voltage supplied from the drive winding W3 of the second transformer T2. The induced current of the driving winding W2 of the second transformer T2 is a forward current, i.e. flows from the different-name terminal to the same-name terminal via the control switching tubes M1 and M2. Since the bipolar transistor Q2 is turned off, no induced current is generated in the drive winding W3 of the second transformer T2.

As shown in fig. 6c, after point a, the resonance current CR and the excitation current CT are negative currents, respectively, and the absolute value of the resonance current CR is smaller than the absolute value of the excitation current CT. Therefore, the net current direction of the resonance current CR and the excitation current CT becomes a positive current. At this time, the bipolar transistor Q1 is maintained in the reverse amplification state, and the bipolar transistor Q2 is maintained in the off state. The induced current of the driving winding W2 of the second transformer T2 becomes a reverse current, i.e., flows from the homonymous terminal to the synonym terminal via the control switching tubes M1 and M2. Since the bipolar transistor Q2 is turned off, no induced current is generated in the drive winding W3 of the second transformer T2.

Fig. 6d and 6e show equivalent circuit diagrams of the LLC resonant converter shown in fig. 2 in the second stage t 2.

In the second phase t2, the clock signal CLK is maintained at a low level, the resonance current CR is changed from a negative current to a positive current, and the current sampling signal CS of the resonance current is greater than-100 mV and less than 100 mV. As can be seen from the operation of the control circuit shown in fig. 3, the first comparison signal generated by the comparator a12 is inactive, the inverted signal generated by the clock signal CLK via the inverter a14 is active, and the gate driving signal VG1 generated according to the first comparison signal and the inverted signal of the clock signal CLK is active, so that the control switch M1 is in a conducting state. Meanwhile, the second comparison signal generated by the comparator a13 is inactive, the clock signal CLK is inactive, and the gate driving signal VG2 generated according to the second comparison signal and the clock signal CLK is inactive, so that the control switch tube M2 is in an off state. Therefore, the control switch transistors M1 and M2 are in a single conduction state, and a unidirectional path is formed from the different-name terminal to the same-name terminal of the drive winding W2 of the second transformer T2 via the control switch transistors M1 and M2.

As shown in fig. 6d, when the resonant current CR is a negative current, that is, the current sampling signal CS of the resonant current is greater than 0 and less than 100mV, the resonant current CR and the excitation current CT are negative currents, respectively, and the absolute value of the resonant current CR is smaller than the absolute value of the excitation current CT. Therefore, the net current direction of the resonant current CR and the excitation current CT is maintained as a positive current. At this time, the bipolar transistor Q1 is in a reverse saturation state, and the bipolar transistor Q2 is maintained in an off state. In the case where the driving winding W2 of the second transformer T2 allows a bidirectional current flow, the induced current is a reverse current, i.e., flows from the homonymous terminal to the heteronymous terminal via the control switching tubes M1 and M2. However, the driving winding W2 of the second transformer T2 has only a unidirectional path from the different name terminal to the same name terminal, and therefore, the induced current generated by the driving winding W2 of the second transformer T2 provides the driving current of the bipolar transistor Q1. Since the bipolar transistor Q2 is turned off, no induced current is generated in the drive winding W3 of the second transformer T2.

As shown in fig. 6e, when the resonant current CR changes from a negative current to a positive current, that is, the current sampling signal CS of the resonant current is less than 0 and greater than-100 mV, the resonant current CR is a positive current, and the excitation current CT is a negative current. Therefore, the net current direction of the resonant current CR and the excitation current CT is maintained as a positive current. At this time, the bipolar transistor Q1 is in the forward on state, and the bipolar transistor Q2 is maintained in the off state. The base-emitter PN junction of bipolar transistor Q1 is turned on, forming an additional path from the tap terminal to the alias terminal of drive winding W2 of second transformer T2. Therefore, the induced current of the driving winding W2 of the second transformer T2 is a reverse current, i.e., flows from the tap terminal to the alias terminal through the PN junction with the base-emitter of the bipolar transistor Q1. Since the bipolar transistor Q2 is turned off, no induced current is generated in the drive winding W3 of the second transformer T2.

Fig. 6e shows an equivalent circuit diagram of the LLC resonant converter shown in fig. 2 in a third stage t 3.

In the third stage t3, the clock signal CLK is maintained low, the resonant current CR is maintained at a positive current, and the current sample signal CS for the resonant current is less than-100 mV. As can be seen from the operation of the control circuit shown in fig. 3, the first comparison signal generated by the comparator a12 is asserted, the inverted signal generated by the clock signal CLK via the inverter a14 is asserted, and the gate driving signal VG1 generated according to the first comparison signal and the inverted signal of the clock signal CLK is asserted, so that the control switch M1 is turned on. Meanwhile, the second comparison signal generated by the comparator a13 is inactive, the clock signal CLK is inactive, and the gate driving signal VG2 generated according to the second comparison signal and the clock signal CLK is inactive, so that the control switch tube M2 is in an off state. Therefore, the control switch transistors M1 and M2 are in a single conduction state, and a unidirectional path is formed from the different-name terminal to the same-name terminal of the drive winding W2 of the second transformer T2 via the control switch transistors M1 and M2.

In the third phase T3, the induced current state in the drive windings W2 and W3 of the second transformer T2 remains unchanged, as shown in fig. 6 e.

Fig. 6f to 6h show equivalent circuit diagrams of the LLC resonant converter shown in fig. 2 in a fourth phase t 4.

In the fourth stage t4, the clock signal CLK is high, the resonant current CR is a positive current, and the current sampling signal CS for the resonant current is less than-100 mV. As can be seen from the operation of the control circuit shown in fig. 3, the first comparison signal generated by the comparator a12 is asserted, the inverted signal generated by the clock signal CLK via the inverter a14 is de-asserted, and the gate driving signal VG1 generated according to the first comparison signal and the inverted signal of the clock signal CLK is asserted, so that the switch transistor M1 is turned on. Meanwhile, the second comparison signal generated by the comparator a13 is inactive, the clock signal CLK is active, and the gate driving signal VG2 generated according to the second comparison signal and the clock signal CLK is active, so that the switch transistor M2 is in a conducting state. Therefore, the switching tubes M1 and M2 are in a double-conduction state, and form a bidirectional path between the same-name terminal and the different-name terminal of the driving winding W2 of the second transformer T2 via the control switching tubes M1 and M2.

As shown in fig. 6f, before point B, the resonance current CR and the excitation current CT are positive currents, respectively, and the absolute value of the resonance current CR is larger than the absolute value of the excitation current CT. The net current is the difference of the resonant current CR minus the excitation current CT, and the net current direction at this stage is still positive. The bipolar transistor Q2 is in the off state under the base-emitter bias voltage supplied from the drive winding W3 of the second transformer T2, and the bipolar transistor Q1 is brought from the saturated state to the off state under the base-emitter bias voltage supplied from the drive winding W2 of the second transformer T2. The induced current of the driving winding W2 of the second transformer T2 is a reverse current, i.e. flows from the homonymous terminal to the synonym terminal via the control switching tubes M1 and M2. Since the bipolar transistor Q2 is turned off, no induced current is generated in the drive winding W3 of the second transformer T2. Since the bipolar transistors Q1 and Q1 are both off, the resonant current CR discharges the capacitor Cm, and the node voltage Vmid gradually decreases.

As shown in fig. 6g, as the node voltage Vmid decreases, when the node voltage Vmid is lower than the ground potential, the PN junction of the base-collector of the bipolar transistor Q2 is turned on, and thus enters the reverse amplification state from the off state, and the bipolar transistor Q1 is maintained in the off state by the base-emitter bias voltage supplied from the drive winding W2 of the second transformer T2. The induced current of the driving winding W2 of the second transformer T2 is a reverse current, i.e. flows from the homonymous terminal to the synonym terminal via the control switching tubes M1 and M2. Since the bipolar transistor Q2 reversely amplifies the state, no induced current is generated in the drive winding W3 of the second transformer T2.

As shown in fig. 6h, after point B, the resonance current CR and the excitation current CT are positive currents, respectively, and the absolute value of the resonance current CR is smaller than the absolute value of the excitation current CT. Therefore, the net current direction of the resonance current CR and the excitation current CT becomes a negative current. At this time, the bipolar transistor Q2 is maintained in the reverse amplification state, and the bipolar transistor Q1 is maintained in the off state. The induced current of the driving winding W2 of the second transformer T2 becomes a forward current, i.e., flows from the different-name terminal to the same-name terminal via the control switching tubes M1 and M2. Since the bipolar transistor Q2 is turned off, no induced current is generated in the drive winding W3 of the second transformer T2.

Fig. 6i and 6j show equivalent circuit diagrams of the LLC resonant converter shown in fig. 2 in a fifth stage t 5.

In the fifth phase t5, the clock signal CLK is maintained at a high level, the resonance current CR is changed from a positive current to a negative current, and the current sampling signal CS of the resonance current is greater than-100 mV and less than 100 mV. As can be seen from the operation of the control circuit shown in fig. 3, the first comparison signal generated by the comparator a12 is inactive, the inverted signal generated by the clock signal CLK via the inverter a14 is inactive, and the gate driving signal VG1 generated according to the first comparison signal and the inverted signal of the clock signal CLK is inactive, so that the switch tube M1 is turned off. Meanwhile, the second comparison signal generated by the comparator a13 is inactive, the clock signal CLK is active, and the gate driving signal VG2 generated according to the second comparison signal and the clock signal CLK is active, so that the switch transistor M2 is in a conducting state. Therefore, the switching tubes M1 and M2 are in a single conduction state, and form a unidirectional path from the dotted terminal to the different terminal of the driving winding W2 of the second transformer T2 via the control switching tubes M1 and M2.

As shown in fig. 6i, when the resonant current CR is a positive current, that is, the current sampling signal CS of the resonant current is less than 0 and greater than-100 mV, the resonant current CR and the excitation current CT are positive currents, respectively, and the absolute value of the resonant current CR is less than the absolute value of the excitation current CT. Therefore, the net current direction of the resonant current CR and the excitation current CT is maintained as a negative current. At this time, the bipolar transistor Q2 is in a reverse saturation state, and the bipolar transistor Q1 is maintained in an off state. In the case where the drive winding W2 of the second transformer T2 allows bidirectional current flow, the induced current is a forward current, i.e., flows from the different-name terminal to the same-name terminal via the control switching tubes M1 and M2. However, the driving winding W2 of the second transformer T2 has only a unidirectional path from the dotted terminal to the dotted terminal, and thus the driving winding W2 of the second transformer T2 generates substantially no induced current. Since the bipolar transistor Q2 is in a reverse saturation state, the induced current generated in the drive winding W3 of the second transformer T2 is a forward current, i.e., flows from the different-name terminal to the same-name terminal via the base-collector PN junction of the bipolar transistor Q2.

As shown in fig. 6j, when the resonant current CR changes from a positive current to a negative current, that is, the current sampling signal CS of the resonant current is greater than 0 and less than 100mV, the resonant current CR is a negative current, and the excitation current CT is a positive current. Therefore, the net current direction of the resonant current CR and the excitation current CT is maintained as a negative current. At this time, the bipolar transistor Q2 is in the forward on state, and the bipolar transistor Q1 is maintained in the off state. In the case where the drive winding W2 of the second transformer T2 allows bidirectional current flow, the induced current is a forward current, i.e., flows from the different-name terminal to the same-name terminal via the control switching tubes M1 and M2. However, the driving winding W2 of the second transformer T2 has only a unidirectional path from the dotted terminal to the dotted terminal, and thus the driving winding W2 of the second transformer T2 generates substantially no induced current. Since the bipolar transistor Q2 is in the forward conduction state, the induced current generated in the drive winding W3 of the second transformer T2 is a forward current, i.e., flows from the different-name terminal to the same-name terminal via the base-emitter PN junction of the bipolar transistor Q2.

Fig. 6j shows an equivalent circuit diagram of the LLC resonant converter shown in fig. 2 in a sixth phase t 6.

In the sixth phase t6, the clock signal CLK is maintained at a high level, the resonance current CR is maintained at a negative current, and the current sampling signal CS of the resonance current is greater than 100 mV. As can be seen from the operation of the control circuit shown in fig. 3, the first comparison signal generated by the comparator a12 is inactive, the inverted signal generated by the clock signal CLK via the inverter a14 is inactive, and the gate driving signal VG1 generated according to the first comparison signal and the inverted signal of the clock signal CLK is inactive, so that the switch tube M1 is in the off state. Meanwhile, the second comparison signal generated by the comparator a13 is asserted, the clock signal CLK is asserted, and the gate driving signal VG2 generated according to the second comparison signal and the clock signal CLK is asserted, so that the switch transistor M2 is in a conducting state. Therefore, the control switch transistors M1 and M2 are in a single conduction state, and a unidirectional path is formed from the dotted terminal to the different-name terminal of the drive winding W2 of the second transformer T2 via the control switch transistors M1 and M2.

In the sixth phase T6, the induced current state in the drive windings W2 and W3 of the second transformer T2 remains unchanged, as shown in fig. 6 j.

Fig. 7 shows a schematic circuit diagram of an LLC resonant converter according to a second embodiment of the invention. The LLC resonant converter 30 includes a first transformer T1, a second transformer T2, bipolar transistors Q1 and Q2, diodes D1 and D2, a resonant capacitor Cr, a resonant inductor Lr, control switching transistors M1 and M2, and a control circuit 200.

On the primary side of the first transformer T1, the primary winding Lp of the first transformer T1, the resonant capacitor Cr, and the resonant inductor Lr form a resonant tank. Between the positive and negative input terminals of the LLC resonant converter 30, bipolar transistors Q1 and Q2 are connected in series, the intermediate node of which is connected to the resonant tank. In the resonant tank, a sampling resistor Rs is connected in series with the primary winding Lp, so that a sampling signal for characterizing the inductor current flowing through the primary winding Lp can be obtained.

On the secondary side of the first transformer T1, diodes D1 and D2 constitute a rectifying circuit. Two ends of the secondary winding are respectively connected with anodes of diodes D1 and D2, and the middle tap of the secondary winding is grounded. An output capacitor Co is connected between the cathodes of diodes D1 and D2 and ground, and provides a dc output voltage across it to the load Rd.

The second transformer T2 includes three windings around the same core, i.e., a load winding W1, drive windings W2 and W3. In the resonant tank, the load winding W1 is connected in series with the primary winding Lp. Also, drive windings W2 and W3 are coupled to the bases of bipolar transistors Q1 and Q2, respectively, but in opposite directions. For example, drive winding W2 has a homonymous terminal, a synonym terminal, and a tap terminal therebetween, and drive winding W3 has a homonymous terminal and a synonym terminal. The dotted terminal DR1 of the drive winding W2 is connected to the current terminal of the control circuit 200 to obtain the excitation current I1.

The control switching tubes M1 and M2 are connected in series in opposite directions between the dotted terminal DR1 and the dotted terminal DR2 of the drive winding W2 of the second transformer T2. The control switch transistors M1 and M2 are, for example, N-type MOSFETs, respectively, and include body diodes connected in anti-parallel, respectively. Unlike the first embodiment, the sources of the control switches M1 and M2 are connected to each other, the drain of the control switch M1 is connected to the dotted terminal DR1 of the driving winding W2, and the drain of the control switch M2 is connected to the dotted terminal DR2 of the driving winding W2. Further, the intermediate node of the control switch transistors M1 and M2 is grounded.

The intermediate node of bipolar transistors Q1 and Q2 is connected to the synonym terminal DR2 of drive winding W2. Specifically, the base and emitter of the bipolar transistor Q1 are connected to the tap and alias terminals of the drive winding W2, respectively. The base and emitter of bipolar transistor Q2 are connected to the synonym and synonym terminals, respectively, of drive winding W3.

The turn ratio of the second transformer T2 is selected according to the electrical characteristics of the bipolar transistors Q1 and Q2. In this embodiment, the load winding W1 of the second transformer T2, the drive winding W2: the turn ratio of drive winding W3 is 1: 18: 6, wherein the tap end of the drive winding W2 divides the drive winding W2 into a first sub-winding between the dotted end DR1 and the tap end and a second sub-winding between the tap end and the unlike end DR2, and the first sub-winding: the turn ratio of the second sub-winding is 12: 6. in an alternative embodiment, the load winding W1 of the second transformer T2, the drive winding W2: the turn ratio of drive winding W3 is 1: 24: 6 or 2: 36: 12. the fixed turn ratio of the first sub-winding to the second sub-winding of drive winding W2 is such that the drive voltage of bipolar transistor Q1 is in a fixed proportional relationship with the terminal voltage of drive winding W2 (i.e., the voltage between the dotted terminal DR1 and the dotted terminal DR 2). For example, in the state where only one of the control switches M1 and M2 is conducting, the terminal voltage of the driving winding W2 is the forward voltage drop of the body diode of the control switch, so that the terminal voltage of the driving winding W2 is 0.7V, and the driving signal clamp of the bipolar transistor Q1 is at 0.7/3V, so that the bipolar transistor Q1 maintains the off state.

The control circuit 200 obtains a current sampling signal CS of the resonant current from the sampling resistor Rs of the resonant converter 30, obtains a voltage feedback signal FB of the resonant output voltage from the auxiliary winding La of the first transformer T1 of the resonant converter 30, generates a gate driving signal VG1 for controlling the switching tube M1 and a gate driving signal VG2 for controlling the switching tube M2 according to the current sampling signal CS and the voltage feedback signal FB, and excites the current I1. In this embodiment, the ground terminal of the control circuit 200, the intermediate node of the control switch transistors M1 and M2, and the homonymous terminal of the auxiliary winding La are grounded together.

Under the control of the control circuit 200, the control switch tubes M1 and M2 have three working modes, i.e., double-off, double-on, and single-on. During the start-up phase of the LLC resonant converter, the switching tubes M1 and M2 are controlled to a double-off state, and the drive winding W2 of the second transformer T2 of the resonant converter 30 injects an excitation current, thereby starting the self-oscillation mode of operation. In the self-oscillation working phase of the LLC resonant converter, the switching tubes M1 and M2 are controlled to be in a double-conduction state or a single-conduction state, so that the conduction states and the cut-off states of the bipolar transistors Q1 and Q2 are forced to be switched in advance, and the LLC resonant converter can obtain the expected direct current output voltage/current by adopting a mode of regulating resonant current.

According to the LLC resonant converter 30 of this embodiment, the drive windings W2 and W3 are used to provide drive currents to drive the bases of bipolar transistors Q1 and Q2 to achieve self-oscillating drive. Under the control of the self-oscillating drive signal, bipolar transistors Q1 and Q2 are alternately turned on and off, converting the dc input voltage into a square wave voltage. The square wave voltage is input to the resonant tank to generate a resonant current at the resonant frequency. Thus, electrical energy is transferred from the primary side of the first transformer T1 to the secondary side of the first transformer T1 through the resonant tank.

The control circuit 200 controls the conduction states of the switching transistors M1 and M2 to provide an additional path and excitation current for the drive winding W2, and further controls the conduction states of the bipolar transistors Q1 and Q2 via magnetic coupling of the drive windings W2 and W3 to achieve consistent off times of the two bipolar transistors to adjust the resonant frequency. Therefore, the LLC resonant converter 30 can adjust the resonant frequency to obtain the desired dc output voltage/current, thereby achieving constant current control, simplifying the control circuit and reducing the circuit cost.

In the first and second embodiments, the control switching transistors M1 and M2 are, for example, N-type MOSFETs, and the control switching transistors M1 and M2 are connected in series in reverse between the dotted terminal DR1 and the dotted terminal DR2 of the driving winding W2 of the second transformer T2. In the first embodiment, the gate driving voltage of the control switching tubes M1 and M2 is, for example, a negative voltage, and the anodes of the parasitic diodes of the control switching tubes M1 and M2 are connected to each other to constitute an internal parasitic PNP transistor. In the case where the control transistors M1 and M2 are single-on, the parasitic PNP transistor may also be turned on. For this reason, a special silicon-on-insulator (abbreviated as SOI) process is required in an integrated circuit process and level shifting is performed in a control circuit. In the second embodiment, the intermediate node between the control switches M1 and M2 is grounded, so that the gate driving voltages of the control switches M1 and M2 are both positive voltages, thereby simplifying the control circuit. Cathodes of the parasitic diodes of the control switching tubes M1 and M2 are connected to each other, and form an internal parasitic NPN transistor. In the case where the control transistors M1 and M2 are single conducting, the parasitic NPN transistors do not turn on. Therefore, in the second embodiment, a special silicon-on-insulator (abbreviated as SOI) process is not required in the integrated circuit process, and level shifting is not required in the control circuit.

Fig. 8 shows a schematic circuit diagram of a control circuit in the LLC resonant converter shown in fig. 7. The control circuit 200 includes a clock signal generating module 110, a logic module 120, comparators a12 and a13, a one-shot module 104, driving modules 105 and 106, a current source Is1, and a switch M3. The clock signal generation module 110 includes a subtraction module 201, a compensation module 202, an oscillator 111, a frequency division module 103, and a capacitor Cc. The logic module 120 includes NOT gates A14, OR gates A15 and A16, and gates A17 and A18.

Two input terminals of the subtraction module 201 are connected to two terminals of the sampling resistor RS shown in fig. 7. In this embodiment, the sampling resistor RS has one end connected to the dotted end of the load winding W1 of the second transformer T2 and the other end connected to the dotted end DR2 of the drive winding W2 of the second transformer T2, i.e., the intermediate node of the bipolar transistors Q1 and Q2. Therefore, the two input terminals of the subtraction module 201 obtain the current sampling signal CS and the node voltage Vmid, respectively, and perform subtraction to obtain the voltage drop across the sampling resistor RS, thereby obtaining the modified current sampling signal CS1 proportional to the resonant current.

The compensation module 202 receives the modified current sampling signal CS1 and/or the voltage feedback signal FB, and generates a compensation signal COMP across the capacitor Cc according to a comparison of the modified current sampling signal CS1 with the current reference signal and/or according to a comparison of the voltage feedback signal FB with the voltage reference signal. In the constant-current control mode, if the direct-current output current is too small, the compensation signal COMP is increased, and if the direct-current output current is too large, the compensation signal COMP is decreased. In the constant voltage control mode, if the dc output voltage is too small, the compensation signal COMP increases, and if the dc output voltage is too large, the compensation signal COMP decreases.

The oscillator 111 further includes a comparator a11, a current source Is2, and a capacitor C11. A current source Is2 and a capacitor C11 are connected in series between the supply terminal and internal ground, the intermediate node of which provides a triangular wave signal. The comparator a11 has a non-inverting input terminal receiving the triangular wave signal, an inverting input terminal receiving the compensation signal COMP, and an output terminal providing the oscillation signal OSC. The frequency of the oscillation signal OSC is related to the compensation signal COMP, and the frequency of the oscillation signal OSC is low when the compensation signal COMP is high, and the frequency of the oscillation signal OSC is high when the compensation signal COMP is low. The frequency divider module 103 has an input terminal receiving the oscillation signal OSC, and an output terminal providing the clock signal CLK. The frequency division module 103 generates a clock signal CLK with a 50% duty cycle according to the oscillation signal OSC.

The comparator a12 compares the modified current sample signal CS1 of the resonant current with a first current threshold (e.g., -100mV) to obtain a first comparison signal, and the comparator a13 compares the modified current sample signal CS1 of the resonant current with a second current threshold (e.g., +100mV) to obtain a second comparison signal. The one-shot module 104 determines whether the modified current sampling signal CS1 is in the predetermined current interval according to the first comparison signal and the second comparison signal, and generates the enable signal Ts.

During start-up of the LLC resonant converter 30, the modified current sampling signal CS1 is in the predetermined current interval, and the start-up signal Ts is active at low level.

In this embodiment, the control switches M1 and M2 are, for example, N-type MOSFETs, and the switch M3 is, for example, P-type MOSFETs. The enable signal Ts is asserted low, causing the switch M3 to conduct. Meanwhile, the enable signal Ts is one of the input signals of the and gates a17 and a18, so that the gate driving signal VG1 provided by the driving module 106 and the gate driving signal VG2 provided by the driving module 105 are disabled, and the control switches M1 and M2 are turned off. A current source Is1 Is connected in series with switch M3 between the supply terminal and the dotted terminal DR1 of drive winding W2.

Therefore, during start-up of the LLC resonant converter 30, the current source Is1 injects the excitation current I1 via the switching tube M3 to the dotted terminal DR1 of the drive winding W2, so that the LLC resonant converter 30 starts to operate in a self-oscillating manner.

Further, after the LLC resonant converter 30 starts operating in the self-oscillation mode, the corrected current sampling signal CS1 of the resonant current exceeds the above-mentioned predetermined current interval for a fixed time of a plurality of switching cycles, and therefore, the start signal Ts maintains the inactive state of the high level. Since the switching tube M3 Is turned off, the current source Is1 stops injecting the excitation current I1 to the dotted terminal DR1 of the drive winding W2 via the switching tube M3.

The control switch tubes M1 and M2 control their conducting states according to the clock signal CLK. As shown, a first input terminal of the or gate a15 is connected to the output terminal of the frequency dividing module 103 via an inverter a14 to receive an inverted signal of the clock signal CLK, and a second input terminal is connected to the output terminal of the comparator a12 to receive a first comparison signal; the or gate a16 has a first input connected to the output of the divider block 103 to receive the clock signal CLK and a second input connected to the output of the comparator a13 to receive the second comparison signal. The AND gate A17 has a first input connected to the output of the OR gate A15, a second input receiving the enable signal Ts, and an output connected to the driver module 105. The AND gate A18 has a first input coupled to the output of the OR gate A16, a second input receiving the enable signal Ts, and an output coupled to the driver module 106. Therefore, when the clock signal CLK is asserted and the modified current sampling signal CS1 is greater than the first current threshold, the control switch M1 is turned off, and the rest is turned on. When the clock signal CLK is inactive and the modified current sampling signal CS1 is less than the second current threshold, the control switch M2 is turned off, and the rest is turned on.

The driving module 106 is used for generating a gate driving signal VG1 for controlling the switch transistor M1, and the driving module 105 is used for generating a gate driving signal VG2 for controlling the switch transistor M2. The driving modules 105 and 106 are, for example, a plurality of inverters connected in series, respectively.

In the above embodiment, it is described that in the LLC resonant converter, the bipolar transistors are controlled to be alternately turned on by controlling the short circuit of the driving winding of the upper bipolar transistor and releasing the short-circuit state at an appropriate time, so that the switching period of the bipolar transistors follows the period of the switching control signal inside the control circuit, and further the frequency of the switching control signal is controlled according to the negative feedback of the resonant current, so as to achieve the constancy of the output current or voltage. However, the present invention is not limited thereto. It will be appreciated that the same technical effect can be achieved by controlling the current path to the drive winding of the lower bipolar transistor of the LLC resonant converter based on similar operating principles.

In the first embodiment described above, it is described that the intermediate node of two bipolar transistors in the LLC resonant converter is grounded, and the compensation module in the control circuit directly obtains the voltage drop of the sampling resistor as the current sampling signal, the first driving module is connected to the dotted terminal of the first driving winding, and performs level shift on the first control signal with respect to the dotted terminal to obtain the first gate driving signal; the second driving module is connected to the synonym terminal of the second driving winding, and performs level shift on the second control signal relative to the synonym terminal to obtain a second gate driving signal.

In the second embodiment described above, it is described that the middle node of the two control switch tubes in the LLC resonant converter is grounded, and the two input terminals of the subtraction module in the control circuit are connected to the two ends of the sampling resistor to obtain the voltage drop of the sampling resistor as the modified current sampling signal, and the first driving module generates the first gate driving signal with respect to the middle node of the control switch tubes M1 and M2; the second driving module generates a second gate driving signal with respect to a middle node of the control switch transistors M1 and M2.

In accordance with the embodiments of the present invention as set forth above, these embodiments are not exhaustive and do not limit the invention to the precise embodiments described. Obviously, many modifications and variations are possible in light of the above teaching. The embodiments were chosen and described in order to best explain the principles of the invention and its practical application, to thereby enable others skilled in the art to best utilize the invention and its various embodiments with various modifications as are suited to the particular use contemplated. The present invention is limited only by the claims and their full scope and equivalents.

Claims (25)

1. An LLC resonant converter, comprising:

the first transformer comprises a primary winding and a secondary winding;

a second transformer comprising a load winding, and first and second drive windings magnetically coupled to the load winding; and

a first bipolar transistor and a second bipolar transistor connected in series between a positive power supply terminal and a negative power supply terminal of a DC input voltage, a load winding, a resonant element and the primary winding of the second transformer connected between a middle node of the first bipolar transistor and the second bipolar transistor and the negative power supply terminal, the resonant element and the primary winding being connected to form a resonant tank to generate a resonant current, a resonant output voltage being provided at both ends of the primary winding,

wherein the LLC resonant converter further includes a control circuit for controlling the conduction state of a path between the dotted terminal and the dotted terminal of the first drive winding, thereby controlling the conduction states of the first and second bipolar transistors to adjust the resonant frequency.

2. The LLC resonant converter of claim 1, wherein said first drive winding includes a homonymous terminal, a heteronymous terminal, and a tap terminal, said second drive winding includes a homonymous terminal and a heteronymous terminal,

a base of the first bipolar transistor is connected to a tap terminal of the first driving winding to receive a first driving current generated according to an induced current of the resonant current,

the base of the second bipolar transistor is connected to the synonym terminal of the second driving winding to receive a second driving current generated according to the induced current of the resonant current.

3. The LLC resonant converter of claim 2, wherein said first and second bipolar transistors are NPN bipolar transistors, an emitter of said first bipolar transistor and a collector of said second bipolar transistor being commonly connected to said intermediate node, an opposite terminal of said first drive winding being connected to an emitter of said first bipolar transistor, and a like terminal of said second drive winding being connected to an emitter of said second bipolar transistor.

4. The LLC resonant converter of claim 2, further comprising:

a first control switch tube and a second control switch tube which are connected in series in reverse between the homonymous terminal and the synonym terminal of the first drive winding,

the control circuit generates a clock signal according to a current sampling signal of the resonant current and/or a voltage feedback signal of the resonant output voltage, and generates a first gate driving signal and a second gate driving signal according to the clock signal, so that the first control switch tube and the second control switch tube are controlled to periodically short-circuit the homonymous end and the synonym end of the first driving winding, and the resonant period of the LLC resonant converter follows the clock signal.

5. The LLC resonant converter according to claim 4, wherein said first control switch tube and said second control switch tube are N-type MOSFETs respectively, a drain of said first control switch tube and a drain of said second control switch tube are connected to each other, a source of said first control switch tube is connected to a homonymous terminal of said first driving winding, a source of said second control switch tube is connected to a synonym terminal of said first driving winding, and a source of said second control switch tube is grounded.

6. The LLC resonant converter of claim 5, wherein the gate of the first control switch receives a first gate drive signal and the gate of the second control switch receives a second gate drive signal.

7. The LLC resonant converter according to claim 4, wherein said first control switch tube and said second control switch tube are N-type MOSFETs respectively, a source of said first control switch tube and a source of said second control switch tube are connected to each other, a drain of said first control switch tube is connected to a homonymous terminal of said first driving winding, a drain of said second control switch tube is connected to a synonym terminal of said first driving winding, and an intermediate node of said first control switch tube and said second control switch tube is grounded.

8. The LLC resonant converter of claim 7, wherein the gate of the first control switch receives a first gate drive signal and the gate of the second control switch receives a second gate drive signal.

9. The LLC resonant converter of claim 4, further comprising: and a sampling resistor connected in series with the primary winding of the first transformer, and having one end connected to a middle node of the first bipolar transistor and the second bipolar transistor, wherein a current sampling signal of the resonant current is obtained at both ends of the sampling resistor.

10. The LLC resonant converter of claim 4, wherein the control circuit comprises:

the clock signal generating module is used for generating the clock signal with a corresponding clock period according to a current sampling signal of the resonant current and/or a voltage feedback signal of the resonant output voltage;

the logic module is connected with the clock signal generating module and generates a first control signal and a second control signal according to the clock signal;

the first driving module is connected with the logic module and generates the first grid driving signal according to the first control signal; and

and the second driving module is connected with the logic module and generates the second grid driving signal according to the second control signal.

11. The LLC resonant converter of claim 10, wherein the control circuit further comprises:

a first comparator to compare a current sampling signal of the resonant current with a first current threshold to obtain a first comparison signal; and

a second comparator to compare a current sample signal of the resonant current with a second current threshold to obtain a second comparison signal,

wherein the logic module is connected with the first comparator to obtain the first comparison signal, and connected with the second comparator to obtain the second comparison signal,

the logic module controls the first gate driving signal to be in an active state under the condition that the clock signal is invalid or the current sampling signal is smaller than the first current threshold value,

and under the condition that the clock signal is effective or the current sampling signal is greater than the second current threshold, the logic module controls the second gate driving signal to be in an effective state.

12. The LLC resonant converter of claim 11, wherein the control circuit further comprises:

the first current source and the switch tube are connected in series between a power supply end and a same-name end of the first driving winding; and

a one-shot module, a first input end of which is connected with the first comparator to obtain the first comparison signal, a second input end of which is connected with the second comparator to obtain the second comparison signal, and an output end of which is connected with the switch tube to provide a starting signal,

during starting of the LLC resonant converter, the current sampling signal is greater than the first current threshold and less than the second current threshold, the starting signal is effective to turn on the switch tube, the first current source injects excitation current to the first drive winding through the switch tube, and the first control switch tube and the second control switch tube are turned off.

13. The LLC resonant converter of claim 12, wherein the logic module comprises:

a NOT gate for obtaining an inverted signal of the clock signal;

a first or gate, a first input terminal receiving an inverted signal of the clock signal, a second input terminal receiving the first comparison signal;

the first input end of the first AND gate is connected to the output end of the first OR gate, and the second input end of the first AND gate receives the starting signal;

a second or gate, the first input terminal receiving the clock signal, the second input terminal receiving the second comparison signal; and

a second AND gate, a first input terminal connected to an output terminal of the second OR gate, a second input terminal receiving the enable signal,

wherein the enable signal is active at a low level.

14. The LLC resonant converter of claim 10, wherein the first drive module is connected to a homonymous terminal of the first drive winding, level shifts the first control signal relative to the homonymous terminal to obtain the first gate drive signal, an

The second driving module is connected to the synonym terminal of the first driving winding, and performs level shift on the second control signal relative to the synonym terminal to obtain the second gate driving signal.

15. The LLC resonant converter of claim 10, wherein the clock signal generation module includes:

the circuit comprises a compensation module and a first capacitor connected to the output end of the compensation module, wherein compensation signals are generated at two ends of the first capacitor;

the oscillator generates an oscillation signal with corresponding frequency according to the compensation signal; and

a frequency division module generating the clock signal with a 50% duty ratio according to the oscillation signal,

wherein the compensation module compares the current sampling signal with a current reference signal and/or compares the voltage feedback signal with a voltage reference signal to generate the compensation signal.

16. The LLC resonant converter of claim 4, wherein said control circuit and said first and second control switches are integrated as a single chip.

17. A control circuit for an LLC resonant converter, said LLC resonant converter comprising a first drive winding coupled with a resonant tank, characterized in that said control circuit comprises:

the clock signal generating module generates a clock signal with a corresponding clock period according to a current sampling signal of the resonant current and/or a voltage feedback signal of the resonant output voltage;

the logic module is connected with the clock signal generating module and generates a first control signal and a second control signal according to the clock signal;

the first driving module is connected with the logic module and generates a first grid driving signal according to the first control signal; and

a second driving module connected with the logic module and generating a second gate driving signal according to the second control signal,

the control circuit controls a first control switch tube and a second control switch tube to periodically short-circuit the homonymous end and the synonym end of the first drive winding by adopting the first grid drive signal and the second grid drive signal, and generates a drive current at the tap end of the first drive winding, so that the resonance period of the LLC resonant converter follows the clock signal.

18. The control circuit of claim 17, further comprising:

a first comparator to compare a current sampling signal of the resonant current with a first current threshold to obtain a first comparison signal; and

a second comparator to compare a current sample signal of the resonant current with a second current threshold to obtain a second comparison signal,

wherein the logic module is connected with the first comparator to obtain the first comparison signal, and connected with the second comparator to obtain the second comparison signal,

the logic module controls the first gate driving signal to be in an active state under the condition that the clock signal is invalid or the current sampling signal is smaller than the first current threshold value,

and under the condition that the clock signal is effective or the current sampling signal is greater than the second current threshold, the logic module controls the second gate driving signal to be in an effective state.

19. The control circuit of claim 18, further comprising:

the first current source and the switch tube are connected in series between a power supply end and a same-name end of the first driving winding; and

a one-shot module, a first input end of which is connected with the first comparator to obtain the first comparison signal, a second input end of which is connected with the second comparator to obtain the second comparison signal, and an output end of which is connected with the switch tube to provide a starting signal,

during starting of the LLC resonant converter, the current sampling signal is greater than the first current threshold and less than the second current threshold, the starting signal is effective to turn on the switch tube, the first current source injects excitation current to the first drive winding through the switch tube, and the first control switch tube and the second control switch tube are turned off.

20. The control circuit of claim 19, wherein the logic module comprises:

a NOT gate for obtaining an inverted signal of the clock signal;

a first or gate, a first input terminal receiving an inverted signal of the clock signal, a second input terminal receiving the first comparison signal;

the first input end of the first AND gate is connected to the output end of the first OR gate, and the second input end of the first AND gate receives the starting signal;

a second or gate, the first input terminal receiving the clock signal, the second input terminal receiving the second comparison signal; and

a second AND gate, a first input terminal connected to an output terminal of the second OR gate, a second input terminal receiving the enable signal,

wherein the enable signal is active at a low level.

21. The control circuit of claim 17, wherein the first driving module is connected to a homonymous terminal of the first driving winding, level-shifts the first control signal relative to the homonymous terminal to obtain the first gate driving signal, and

the second driving module is connected to the synonym terminal of the first driving winding, and performs level shift on the second control signal relative to the synonym terminal to obtain the second gate driving signal.

22. The control circuit of claim 17, wherein the clock signal generation module comprises:

the circuit comprises a compensation module and a first capacitor connected to the output end of the compensation module, wherein compensation signals are generated at two ends of the first capacitor;

the oscillator generates an oscillation signal with corresponding frequency according to the compensation signal; and

a frequency division module generating the clock signal with a 50% duty ratio according to the oscillation signal,

wherein the compensation module compares the current sampling signal with a current reference signal and/or compares the voltage feedback signal with a voltage reference signal to generate the compensation signal such that the frequency of the clock signal is adjusted in relation to the resonant current and/or the resonant output voltage.

23. The control circuit according to claim 17, wherein the first control switch tube and the second control switch tube are N-type MOSFETs respectively, a drain of the first control switch tube and a drain of the second control switch tube are connected to each other, a source of the first control switch tube is connected to a dotted terminal of the first driving winding, a source of the second control switch tube is connected to a dotted terminal of the first driving winding, and a source of the second control switch tube is grounded,

the grid electrode of the first control switch tube receives a first grid electrode driving signal, and the grid electrode of the second control switch tube receives a second grid electrode driving signal.

24. The control circuit according to claim 17, wherein the first control switch tube and the second control switch tube are N-type MOSFETs respectively, a source of the first control switch tube and a source of the second control switch tube are connected to each other, a drain of the first control switch tube is connected to a homonymous terminal of the first driving winding, a drain of the second control switch tube is connected to a synonym terminal of the first driving winding, and an intermediate node of the first control switch tube and the second control switch tube is grounded,

the grid electrode of the first control switch tube receives a first grid electrode driving signal, and the grid electrode of the second control switch tube receives a second grid electrode driving signal.

25. The control circuit of claim 17, wherein the control circuit and the first and second control switches are integrated into a single chip.

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