CN210535825U - Small broadband dual-polarized antenna based on non-uniform hyperplane - Google Patents

Abstract

The utility model provides a small-size broadband dual polarized antenna based on inhomogeneous hyperplane contains hyperplane and dipole antenna, and the hyperplane is for distributing 4X 4 square becket units on the dielectric plate, and dipole antenna contains the radiator, coaxial line and reflector, and four square becket units are put for the field font, constitute the dipole, and the coaxial line inner conductor is connected to the lower surface of the reflector of transmitter behind the outer conductor. The utility model discloses have stable radiation pattern, it is insensitive to the distance between radiator and the radiator. In other words, a stable radiation pattern can be obtained even if the antenna height is low. Simulation and actual measurement results show that the system realizes stable radiation direction, stable gain, low cross polarization and high port isolation in a working frequency band. The proposed antenna would be very promising in base station applications.

Description

Small broadband dual-polarized antenna based on non-uniform hyperplane

Technical Field

The invention relates to the field of antennas, in particular to a base station dipole antenna.

Background

In modern wireless communication systems, a ± 45 ° dual-polarized antenna is widely used in a base station because it can provide polarization diversity to reduce the side effects of multipath fading and increase channel capacity. In recent years dual polarized antennas have found a lot of applications, most of which are in the form of crossed dipoles with a metallic ground plane as radiator. Due to the reflective nature of the metal surface, the height of the antenna is typically about 0.25 λ 0, where λ 0 is the wavelength of the center operating frequency [3,4 ]. The base station antenna mainly works in two frequency bands, namely a low frequency band (0.69-0.96GHz) and a high frequency band (1.7-2.7GHz), and the normal height of the base station antenna working in the lower frequency band is about 90 mm.

However, with the rapid development of mobile communication systems, such a height has not been able to satisfy the requirement for miniaturization of base station antennas in many cases. The difficulty in further reducing the height of the base station antenna mainly lies in: (1) the radiation pattern deteriorates after the height is reduced. Since the distance between the radiator and the radiator is less than a quarter of a wavelength, the reflected wave no longer superimposes in phase with the wave radiated directly to the boresight. (2) Broadband impedance matching is difficult to achieve. The reduced height brings the radiator of the dipole antenna closer to the reflector, making the mutual coupling between the two considerably stronger, and the input impedance of the antenna becomes more reactive over a wide frequency band, making broadband impedance matching difficult to achieve.

In order to solve the above problems, an Artificial Magnetic Conductor (AMC) surface is used to change the reflection characteristics of a metal radiator. By placing the AMC surface between the radiator and the radiator, the antenna radiation pattern is kept stable and a good impedance matching is achieved after a reduction in height. However, since the operating bandwidth of AMC is relatively narrow, the final operating bandwidth (22%) of an antenna using AMC is not wide enough. In addition, loading the AMC results in bulky antenna structures and high manufacturing costs, since the layout of the AMC building blocks typically requires more space. A similar design can also be seen. In modern mobile communication systems, miniaturized, low-cost broadband dual-polarized antennas are highly desirable.

When the height of the antenna is 90mm (0.25 λ 0), it corresponds to an open-ended flared transmission line, and the input impedance is purely resistive and easily matched to a 50 Ω transmission line.

Disclosure of Invention

In order to overcome the defects of the prior art, the invention provides a small dual-polarized antenna based on a non-uniform hyperplane.

The technical scheme adopted by the invention for solving the technical problems is as follows:

a small dual-polarized antenna based on a non-uniform hyperplane comprises a hyperplane and a dipole antenna, wherein 4 x 4 square metal ring units are distributed on a dielectric plate on the hyperplane, 16 metal ring units are divided into 4 inner ring metal ring units and 12 outer ring metal ring units, the width of the metal ring on the outer ring of the hyperplane is Rw1, the width of the metal ring on the inner ring of the hyperplane is Rw2, and the antenna efficiency at two resonant frequencies is adjusted by adjusting the width of the metal ring;

the dipole antenna comprises a radiator, a coaxial line and a reflector, wherein four square metal plates are arranged in a field shape to form two crossed dipoles, namely, every two diagonally arranged square metal plates form a dipole, and the structure formed by the two dipoles is called as a radiator; the coaxial line comprises an inner conductor and an outer conductor, wherein the inner conductor is two metal pieces with U-shaped structures and is respectively called CL1 and CL2, the U-shaped part of each inner conductor is divided into three parts, the 1 st part and the 3 rd part are vertical cylinders with the same diameter, the length of the 3 rd part is greater than that of the 1 st part, the 2 nd part is a middle cross bar for connecting the 1 st part and the 3 rd part, and in order to avoid the cross bars of the two coaxial line inner conductors from colliding after crossing, the CL1 is integrally in a convex shape with an opening at the lower part, namely the middle section of the cross bar of the 2 nd part of the CL1 is upwards convex; the CL2 is integrally in a concave shape with an opening at the lower part, namely the middle section of the cross bar of the 2 nd part of the CL2 is concave downwards; placing CL1 and CL2 at the center of the lower part of the radiator after crossing vertically, the 2 nd part of CL1/CL2 crosses the radiator and the coaxial line outer conductor part, therefore, the antenna-shaped notch and the outer conductor on the radiator near CL1 and CL2 need to be cut off to allow the 2 nd parts of CL1 and CL2 to connect the 1 st and 3 rd parts through the notch of the radiator and the coaxial line outer conductor; the outer conductor is a hollow metal tube, the two coaxial line inner conductors are connected to the radiator after crossing, and two No. 3 parts of the coaxial line inner conductors are connected to the lower surface of the reflector of the emitter after passing through the outer conductor, and CL1 is connected to the port 1 end of the reflector, and CL2 is connected to the port 2 end of the reflector.

The dielectric board of the hyperplane is a printed circuit board based on FR4, and the thickness St is 0.8 mm, and the dielectric constant is 4.4.

The invention has the advantages that the invention has stable radiation pattern and is insensitive to the distance between the radiators. In other words, a stable radiation pattern can be obtained even if the antenna height is low. Simulation and actual measurement results show that the system realizes stable radiation direction, stable gain, low cross polarization and high port isolation in a working frequency band. The proposed antenna would be very promising in base station applications.

Drawings

Fig. 1 is a schematic diagram of S parameters of port 1 and port 2, where fig. 1(a) is a schematic diagram of S11 parameters, fig. 1(b) is a schematic diagram of S22 parameters, and fig. 1(c) is a schematic diagram of S21 parameters.

Fig. 2 is a simulated and measured HPBW of the present invention.

FIG. 3 shows simulated and measured visual axis gain achieved by the present invention.

Fig. 4 is a simulated and actual normalized radiation pattern of the present invention, wherein the three patterns are simulated and actual normalized radiation patterns of the antenna at frequencies of 0.69GHz, 0.825GHz, and 0.96GHz, respectively.

Fig. 5(a) is a dipole antenna geometry, fig. 5(b) antenna internal CL1 and CL2 geometries, fig. 5(c) hyper-plan view, fig. 5(d) antenna side view, fig. 5(e) assembly schematic.

Fig. 6 shows the relationship between HPBW and frequency for different antenna heights according to the present invention.

Fig. 7 is a diagram showing an electric field distribution between a radiator and a reflector of an antenna of different heights according to the present invention, where fig. 7(a) is a diagram showing an electric field distribution between a radiator and a reflector when Ah is 90mm, and fig. 7(b) is a diagram showing an electric field distribution between a radiator and a reflector when Ah is 35 mm.

Fig. 8 is a schematic diagram of the mutual coupling and the terminal load effect of the present invention, fig. 8(a) is a schematic diagram equivalent to an open-ended transmission line when the antenna height is 90mm (0.25 λ 0), and fig. 8(b) is a schematic diagram of the mutual coupling between the radiator and the reflector being very strong and the terminal load effect being introduced when the antenna height is very low (0.1 λ 0).

FIG. 9 is a schematic view of S11 under different Uw of the present invention.

FIG. 10 is a schematic view of S11 under different Rw1 and Rw2 in accordance with the present invention.

Detailed Description

The invention is further illustrated with reference to the following figures and examples.

The invention provides a small-sized dual-polarized antenna based on a non-uniform hyperplane, which comprises a hyperplane and a dipole antenna, wherein 4 multiplied by 4 square metal ring units are distributed on a dielectric plate on the hyperplane, 16 metal ring units are divided into 4 inner ring metal ring units and 12 outer ring metal ring units, the width of the metal ring on the outer ring on the hyperplane is Rw1, the width of the metal ring on the inner ring on the hyperplane is Rw2, and the antenna efficiency at two resonant frequencies is adjusted by adjusting the width of the metal ring; as shown in fig. 10, the outer ring unit width Rw1 becomes wider, the depth of the lower resonance frequency S11 becomes deeper, and the antenna efficiency becomes higher; the width Rw2 of the inner ring unit is widened, the depth of S11 at two resonant frequencies is deepened, but the higher resonant frequency is more sensitive, and good impedance matching can be achieved on a broadband.

The dipole antenna comprises a radiator, a coaxial line and a reflector, as shown in fig. 5(e), four square metal plates are arranged in a field shape to form two crossed dipoles, namely, every two diagonally arranged square metal plates form a dipole, and the structure formed by the two dipoles is called as a radiator; the coaxial line comprises an inner conductor and an outer conductor, wherein the inner conductor is two metal pieces with U-shaped structures and is respectively called CL1 and CL2, the U-shaped part of each inner conductor is divided into three parts, the 1 st part and the 3 rd part are vertical cylinders with the same diameter, the length of the 3 rd part is greater than that of the 1 st part, the 2 nd part is a middle cross bar for connecting the 1 st part and the 3 rd part, and in order to avoid the cross bars of the two coaxial line inner conductors from colliding after crossing, the CL1 is integrally in a convex shape with an opening at the lower part, namely the middle section of the cross bar of the 2 nd part of the CL1 is upwards convex; the CL2 is integrally in a concave shape with an opening at the lower part, namely the middle section of the cross bar of the 2 nd part of the CL2 is concave downwards; placing CL1 and CL2 at the center of the lower part of the radiator after crossing vertically, the 2 nd part of CL1/CL2 crosses the radiator and the coaxial line outer conductor part, therefore, the antenna-shaped notch and the outer conductor on the radiator near CL1 and CL2 need to be cut off to allow the 2 nd parts of CL1 and CL2 to connect the 1 st and 3 rd parts through the notch of the radiator and the coaxial line outer conductor; the outer conductor is a hollow metal tube, the two coaxial line inner conductors are connected to the radiator after crossing, and two No. 3 parts of the coaxial line inner conductors are connected to the lower surface of the reflector of the emitter after passing through the outer conductor, and CL1 is connected to the port 1 end of the reflector, and CL2 is connected to the port 2 end of the reflector, and the internal structures of CL1 and CL2 are shown in FIG. 5 (b).

The dielectric board of the hyperplane is a printed circuit board based on FR4, and the thickness St is 0.8 mm, and the dielectric constant is 4.4.

The two U-shaped coaxial line inner conductors penetrating through the reflectors are vertically and crossly arranged, and when the two ports are respectively excited, the dipole antenna radiates-45-degree and + 45-degree linear polarized waves.

The invention adopts the non-uniform hyperplane to realize the broadband impedance matching. The hyperplane is composed of 16 square metal ring units, and is arranged according to 4 multiplied by 4 layout, and each metal ring unit is a rectangular ring; to achieve better impedance matching at the operating band (0.69-0.96GHz), the outer 12 elements have a wider annular band than the inner 4 elements, as shown in fig. 5 (c). In the invention, the hyperplane is attached to a dielectric plate with the thickness St of 0.8 mm and the dielectric constant of 4.4. The dielectric plate is placed on top of the radiator of the dipole, very close to the radiator. Four plastic cylinders with the thickness of 2mm are adopted to support the hyperplane and the antenna, the positions are at four outer corners between the hyperplane and the radiator, the joints are connected by adhesive, and the assembly diagram of the invention is shown in fig. 5 (e). The dimensions of the final design are shown in the following table (unit: mm).

The present embodiment is uniquely determined when the center frequency is 0.825GHz and the operating frequency band is 0.69-0.96GHz, and the dimensions in the following table will change if the present embodiment operates at other frequencies.

Pll	Plg	Gl	Pd	d1	d2	Rw1	Rw2	Ug	Uw
										85	4	320	34	10	4.3	5	1	1	43
Plt	St	Cg	Ag	Ah	P1	P2	P3	Ct	Ch
										2.5	0.8	8.5	2	37.8	45	48	30	2.5	2.5

In the table, the definition of the respective dimensions is as follows:

pll is the side length of a single square metal plate forming a dipole;

plg is the distance between two adjacent square metal plates forming a dipole;

gl is the side length of the square reflector;

pd is the horizontal distance between the vertical parts of the two coaxial line inner conductors;

d1 is the coaxial outer conductor diameter;

d2 is the diameter of the coaxial line inner conductor;

rw1 is the hyperplane outer ring element width;

rw2 is the width of the hyperplane inner circle unit;

ug is the distance between the inner ring unit and the outer ring unit of the hyperplane;

uw is the length of the hyperplane unit;

plt is the radiator thickness;

st is the hyperplane thickness;

cg is the distance between the highest position of CL1 and the lowest position of CL 2;

ag is the distance between the hyperplane and the radiator;

ah is the overall height of the invention;

p1 is coaxial line inner conductor part 3 height;

p2 is coaxial line inner conductor 2 part length;

p3 is the height of the 1 st part of the coaxial line inner conductor;

ct is the thickness of the 2 nd part of the coaxial line inner conductor;

ch is the upward convex thickness of part 2 of coaxial line inner conductor CL1, and is also the downward concave thickness of part 2 of CL 2.

The common height of the prior dipole antenna is 0.25 lambda₀，λ₀For the central frequency wavelength of the antenna, the problems of deteriorated radiation pattern, difficult realization of broadband impedance matching and the like will occur after the height of the antenna is reduced. The metamaterial is applied to the position 2mm above the dipole antenna with the reduced height, so that the performance of the antenna can be consistent with that of the dipole antenna before the height is reduced, and some parameters are even superior to those of the original dipole antenna. In the invention, the total height of the hyperplane and dipole antenna is 0.1 lambda 0, which is greatly reduced from the original antenna height of 0.25 lambda 0.

The working frequency of the small-sized dual-polarized antenna based on the non-uniform hyperplane covers the low frequency band (0.69-0.96GHz) of the base station antenna, in order to realize broadband impedance matching after the height of the antenna is reduced, a hyperplane which is non-uniform and has the same area with the radiator is arranged in parallel at the position of 2mm (0.005 wavelength) right above the radiator, and the total size of the antenna is almost not changed after the hyperplane is added because the distance between the hyperplane and the radiator is negligible relative to the size of the antenna. Since the hyperplane itself has an inductive character, it is possible to supplement the capacitive loading due to the reduced height of the antenna, and thus the addition of the hyperplane will introduce two adjacent resonant frequencies within the operating frequency band.

The simulated and measured S parameters for ports 1 and 2 are shown in fig. 1. The measurement results and simulation results of S11 and S22 are consistent, and are respectively shown in FIG. 1(a) and FIG. 1 (b); where S11 in fig. 1(a) is the input reflection coefficient, i.e., the input return loss, S22 in fig. 1(b) is the output reflection coefficient, i.e., the output return loss, and S21 in fig. 1(c) is the forward transmission coefficient, i.e., the gain. The measured result shows that S11 is lower than-12.5 dB and S22 is lower than-12 dB on the working frequency band (0.69-0.96 GHz). As shown in fig. 1(c), S21 indicates that the isolation between the two ports is higher than 25 dB. In addition to manufacturing errors, the differences between the measurements and simulations at certain frequencies are mainly due to the low level of simulated S21 (about-30 dB), which makes accurate measurements by the Vector Network Analyzer (VNA) difficult.

The change of the HPBW in the operating band is shown in fig. 2. The results show that the simulated HPBW is between 59 and 67 degrees, and the measured HPBW is between 61 and 68 degrees after the hyperplane is added, and the radiation pattern is very stable.

Simulated and measured boresight realized gains of 8.7-10.1dBi and 7.4-9.2dBi, respectively, as shown in FIG. 3. Note: the co-polarisation is-45 deg. linear polarisation because only port 1 is energised in the above case. As shown, the cross-polarization isolation (XPI) of the visual axis is higher than 28dB over the entire frequency band, indicating good polarization.

The normalized radiation pattern in the xOz plane is shown in fig. 4. It can be seen that a stable radiation pattern is achieved over the entire operating frequency band.

The effect of antenna height reduction on antenna performance in terms of radiation pattern and impedance matching was investigated by analyzing the Half Power Beamwidth (HPBW) and S11 over the operating band (0.69-0.96 GHz). To do this, we gradually reduced the antenna height from 90mm to 35mm and observed the changes in HPBW and S11.

Fig. 6 shows HPBW versus frequency for different antenna heights. When the height of the antenna is 90mm, the variation range of the HPBW in the working frequency band is 63-69 degrees, and the HPBW is slightly narrowed along with the reduction of the height of the antenna. The HPBW varies from 60 degrees to 67 degrees when the antenna height is reduced to 35 millimeters. Note that only port 1 is in the energized state, as is the case hereinafter.

The height reduction does not significantly change the radiation pattern of the antenna. To explain this phenomenon, radiators of antennas of different heights and electric field distribution between the radiators are plotted, as shown in fig. 7. When the antenna height (Ah) is 90mm, the wave radiated to the radiator by the radiator propagates in the form of a TEM wave, and therefore, the direction of the E field is the same as the direction of the wave radiated to the boresight direction (+ z axis) in the xy plane shown in fig. 7 (a). However, when the height of the antenna is reduced to 35mm, since the radiators are four square plates, they constitute a relatively large conductor surface (0.48 λ)₀*0.48λ₀) Thus the conductor surface forms a cavity together with the radiator, as shown in fig. 7(b), the cavity area being surrounded by a dotted line. In this case, the waves in the cavity are in the TM₁₀A mode exists in which the direction of the electric field is along the Z-axis, as shown in fig. 7 (b). At the same time, a portion of the wave radiates outward from the cavity edge in phase with the wave radiated by the radiator in the boresight direction. Thus, the radiation pattern of the dipole antenna proposed herein is highly insensitive to the antenna.

For impedance matching after antenna height reduction, S11 for different antenna heights is shown in fig. 8. It can be seen that S11 is below-10 dB over the operating band when the antenna height is 90 mm. As the antenna height decreased from 90mm to 35mm, S11 increased above-2.5 dB, indicating a very poor impedance match. This is because when the normal height of the antenna is 90mm (0.25 λ)₀) It is equivalent to the split-flared transmission line [16 ] shown in FIG. 8(a)]The input impedance is purely resistive and is easily matched to a 50 Ω transmission line. However, when the antenna height is very low, the mutual coupling between the radiator and the radiator becomes very strong, as shown in fig. 8(b), and a terminal load effect is introduced. Therefore, the input impedance becomes more reactive and it is difficult to match with a 50 Ω transmission line.

It is not difficult to see that the end-loading effect is capacitive, so we propose to add an inductive hyperplane on top of the antenna, as shown in fig. 8 (b). The meta-surface consists of 16 cells arranged in a 4 x 4 layout. Each cell is an inductive rectangular loop. As shown in fig. 9, the addition of the hyperplane introduces two resonant frequencies, the frequencies being related to the size of the unit (Uw), which was then determined in the simulation to achieve good impedance matching throughout the operating band (0.69-0.96 GHz). Here the antenna height is set to 35 mm. Due to the structural symmetry between port 1 and port 2, only S11 for port 1 is shown here.

After Uw is determined, the widths of the inner 4 cells (Rw1) and the outer 12 cells (Rw2) are further independently optimized to achieve better impedance matching in the operating band, as shown in fig. 10.

As shown in fig. 10, despite the reduced antenna height (35mm), S11 remained below-10 dB over the operating band after the addition of the hyperplane, indicating that the reactance of the input impedance has been cancelled by the hyperplane. Specifically, near 0.71GHz (low tilt) and 0.925GHz (high tilt), there are two tilt angles introduced by the hyperplane. By comparing the four curves for the different Rw1 and Rw2 we can find that Rw1 is the depth of low inclination and Rw2 is the depth of high inclination. Therefore, we can appropriately select the values of Rw1 and Rw2 to achieve better impedance matching. Therefore, in the simulation, S11 may remain below-12.5 dB, as shown in fig. 10 (Rw1 ═ 1 mm, Rw2 ═ 5 mm). As for the radiation pattern after the addition of the hyperplane, there was little change in HPBW.

Claims (2)

1. A small-size broadband dual polarized antenna based on inhomogeneous hyperplane which characterized in that:

the small dual-polarized antenna based on the non-uniform hyperplane comprises a hyperplane and a dipole antenna, wherein 4 x 4 square metal ring units are distributed on a dielectric plate on the hyperplane, 16 metal ring units are divided into 4 inner ring metal ring units and 12 outer ring metal ring units, the width of the metal ring on the outer ring of the hyperplane is Rw1, the width of the metal ring on the inner ring of the hyperplane is Rw2, and the antenna efficiency at two resonant frequencies is adjusted by adjusting the width of the metal ring;

the dipole antenna comprises a radiator, a coaxial line and a reflector, wherein four square metal plates are arranged in a field shape to form two crossed dipoles, namely, every two diagonally arranged square metal plates form a dipole, and the structure formed by the two dipoles is called as a radiator; the coaxial line comprises an inner conductor and an outer conductor, wherein the inner conductor is two metal pieces with U-shaped structures and is respectively called CL1 and CL2, the U-shaped part of each inner conductor is divided into three parts, the 1 st part and the 3 rd part are vertical cylinders with the same diameter, the length of the 3 rd part is greater than that of the 1 st part, the 2 nd part is a middle cross bar for connecting the 1 st part and the 3 rd part, and in order to avoid the cross bars of the two coaxial line inner conductors from colliding after crossing, the CL1 is integrally in a convex shape with an opening at the lower part, namely the middle section of the cross bar of the 2 nd part of the CL1 is upwards convex; the CL2 is integrally in a concave shape with an opening at the lower part, namely the middle section of the cross bar of the 2 nd part of the CL2 is concave downwards; placing CL1 and CL2 at the center of the lower part of the radiator after crossing vertically, the 2 nd part of CL1/CL2 crosses the radiator and the coaxial line outer conductor part, therefore, the antenna-shaped notch and the outer conductor on the radiator near CL1 and CL2 need to be cut off to allow the 2 nd parts of CL1 and CL2 to connect the 1 st and 3 rd parts through the notch of the radiator and the coaxial line outer conductor; the outer conductor is a hollow metal tube, the two coaxial line inner conductors are connected to the radiator after crossing, and two No. 3 parts of the coaxial line inner conductors are connected to the lower surface of the reflector of the emitter after passing through the outer conductor, and CL1 is connected to the port 1 end of the reflector, and CL2 is connected to the port 2 end of the reflector.

2. The small-sized broadband dual-polarized antenna based on the non-uniform hyperplane as claimed in claim 1, wherein:

the dielectric board of the hyperplane is a printed circuit board based on FR4, and the thickness St is 0.8 mm, and the dielectric constant is 4.4.

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