CN117639581A - Permanent magnet synchronous motor sensorless control method - Google Patents
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Abstract
The invention discloses a control method of a permanent magnet synchronous motor without a position sensor, which comprises the following steps: constructing a discrete sliding mode observer; based on a discrete sliding mode observer, acquiring back electromotive force; performing phase compensation on the counter electromotive force; a phase-locked loop based on flux linkage compensation is adopted to carry out phase locking on the back electromotive force after compensation, so that the rotor speed and the rotor position are obtained; and obtaining a driving control signal of the inverter according to the rotor speed and the rotor position, and further controlling the motor. The control method for the permanent magnet synchronous motor without the position sensor can eliminate errors caused by the quasi-sliding mode movement, improve the position estimation precision of a sliding mode observer under the working condition of low carrier ratio, and inhibit the rotor position estimation errors in the acceleration and deceleration process while keeping the quasi-steady state performance.
Description
Technical Field
The invention relates to a control method for a permanent magnet synchronous motor without a position sensor, and belongs to the field of electric transmission.
Background
The permanent magnet synchronous motor is used as an alternating current motor, and has the advantages of compact structure, small volume, light weight, high efficiency, high power density, reliable operation, convenient maintenance and the like, so that the permanent magnet synchronous motor is widely applied to the fields of industrial production, electric automobiles, aerospace, wind power generation and the like. In order to overcome the influence of using a position sensor on a system, a position sensor-free control strategy gradually replaces the use of the position sensor, wherein the sliding mode observer is widely applied due to the advantages of strong robustness, insensitivity to system parameter change and external interference, easiness in implementation and the like.
In practical applications, there are still many errors in the sliding mode observer under the working condition of low carrier ratio, including:
(1) The problem of discretization error is solved, the traditional method usually designs a position observer under a continuous domain, and then the discretization observer can cause estimated position error under the working condition of low carrier ratio;
(2) Errors exist in the sliding mode observer, and estimated position errors can be caused by the quasi-sliding mode movement;
(3) The error caused by a conventional phase-locked loop is typically combined with the phase-locked loop to obtain estimated rotor position and speed information after the back EMF is obtained by a sliding mode observer.
The traditional phase-locked loop control method is simple, the response speed is high, accurate and smooth rotor position and speed information can be obtained in the constant-speed running process, but when the motor is in the acceleration and deceleration process, even if the rotating speed can be accurately estimated, the rotor position still has hysteresis error, and the error is increased along with the increase of the acceleration.
In order to eliminate the discretization error of the sliding-mode observer under the working condition of low carrier ratio, document 'Discrete-time SMO based Sensorless Control of CSC-fed PMSM Drives with Low Switching Frequency' proposes a low sampling frequency position-free sensor control method based on a Discrete-time sliding-mode observer, a Discrete-domain sliding-mode observer is designed, and the estimation precision and the robustness to parameter mismatch are improved by comparing and analyzing the traditional sliding-mode observer and the Discrete-time sliding-mode observer. But does not analyze and compensate for the estimated position error caused by the quasi-sliding mode motion and the estimated position error caused by the conventional phase-locked loop during motor acceleration and deceleration.
In order to reduce the estimated position error generated in the dynamic process of the traditional phase-locked loop, the document 'Adaptive PLL-Based Sensorless Control for Improved Dynamics of High-Speed PMSM' proposes a self-Adaptive quadrature phase-locked loop (AQPLL), and the bandwidth during the acceleration of a motor is adjusted in real time, so as to achieve the purposes of reducing the dynamic error of the system and improving the dynamic performance. But does not take into account the problems of discretization errors and estimated position errors caused by quasi-sliding mode motion.
Therefore, a more intensive study on a control method of the permanent magnet synchronous motor without a position sensor is necessary to solve the problems of discretization errors of a traditional sliding mode observer, errors caused by quasi-sliding mode movement and dynamic errors caused by a traditional phase-locked loop.
Disclosure of Invention
In order to overcome the problems, the inventor of the present invention has conducted intensive studies and devised a sensorless control method of a permanent magnet synchronous motor, comprising the steps of:
s1, constructing a discrete sliding mode observer;
s2, based on a discrete sliding mode observer, acquiring back electromotive force;
s3, performing phase compensation on the counter electromotive force;
s4, phase locking is carried out on the back electromotive force after compensation by adopting a phase-locked loop based on flux linkage compensation, so that the rotor speed and the rotor position are obtained;
s5, obtaining a driving control signal of the inverter according to the rotor speed and the rotor position, and further controlling the motor.
In a preferred embodiment, in S1, the discrete sliding mode observer is represented as:
wherein a, b and c are coefficient parameters,to observe the stator current, u s For the measured voltage vector, L smo For the sliding mode coefficient, Z is a switching function, and the superscript k indicates the kth sampling time.
In a preferred embodiment, the coefficient parameters are set to:
wherein T is sc Represents the sampling step size, R s The stator resistance is represented, and L the inductance is represented.
In a preferred embodiment, the slip form coefficient L smo The method comprises the following steps:
wherein k is g A tuning parameter greater than 0 and less than 1.
The switching function Z is set to:
wherein Z is 0 For the current error boundary value, set as:
ψ f indicating motor flux linkage omega e Indicating rotor speed, f s Represents the sampling frequency, k z Is a tuning parameter greater than 1.
In a preferred embodiment, in S2, the back emf is expressed as:
wherein E is k Representing the back emf,the current error at the kth sampling time is represented, j represents the imaginary part of the complex number, ω e Indicating the rotor speed.
In a preferred embodiment, in S3, the back emf after phase compensation is expressed as:
wherein,is the back electromotive force after phase compensation.
In a preferred embodiment, in S4, the phase-locked loop based on flux compensation is expressed as:
wherein,representing the estimated rotor speed obtained by the phase-locked loop, < >>Representing the estimated rotor position obtained by the phase-locked loop, h being the phase-locked loop parameter, ψ f Is the motor flux linkage, delta phi f Is a flux linkage compensation term, E d For counter-electromotive force after phase compensation->Projection on d-axis, E q For counter-electromotive force after phase compensation->Projection on q-axis, s denotes Laplace operator.
In a preferred embodiment, the flux linkage compensation term Δψ f Can be obtained by E d Obtained through a PI controller, expressed as:
wherein k is p 、k i And s is a Laplace operator and is a PI controller parameter.
The invention also provides an electronic device, comprising:
at least one processor; and
a memory communicatively coupled to the at least one processor; wherein the memory stores instructions executable by the at least one processor to enable the at least one processor to perform the method of any one of the preceding claims.
The invention also provides a computer readable storage medium storing computer instructions for causing the computer to perform the method of any one of the above.
The invention has the beneficial effects that:
(1) The method of the invention not only can eliminate errors caused by the quasi-sliding mode movement and improve the position estimation precision of the sliding mode observer under the working condition of low carrier ratio, but also has simple structure and small calculated amount;
(2) Compared with a traditional sliding mode observer combined with a traditional phase-locked loop, the method and the device have the advantages that the steady-state performance is maintained, and meanwhile, the rotor position estimation error in the acceleration and deceleration process is restrained.
Drawings
FIG. 1 is a flow chart of a permanent magnet synchronous motor sensorless control method according to a preferred embodiment of the present invention;
FIG. 2 shows simulation results of rotor position in example 1;
FIG. 3 shows simulation results of the rotor position in comparative example 1
FIG. 4 shows rotor position error in example 1;
FIG. 5 shows the rotor position error in comparative example 1;
fig. 6 shows the motor dynamic process in embodiment 2;
fig. 7 shows the motor dynamic process in comparative example 2.
Detailed Description
The invention is further described in detail below by means of the figures and examples. The features and advantages of the present invention will become more apparent from the description.
The word "exemplary" is used herein to mean "serving as an example, embodiment, or illustration. Any embodiment described herein as "exemplary" is not necessarily to be construed as preferred or advantageous over other embodiments. Although various aspects of the embodiments are illustrated in the accompanying drawings, the drawings are not necessarily drawn to scale unless specifically indicated.
The invention discloses a control method of a permanent magnet synchronous motor without a position sensor, which is characterized by comprising the following steps:
s1, constructing a discrete sliding mode observer;
s2, based on a discrete sliding mode observer, acquiring back electromotive force;
s3, performing phase compensation on the counter electromotive force;
s4, phase locking is carried out on the back electromotive force after compensation by adopting a phase-locked loop based on flux linkage compensation, so that the rotor speed and the rotor position are obtained;
s5, obtaining a driving control signal of the inverter according to the rotor speed and the rotor position, and further controlling the motor.
Preferably, in S1, the discrete sliding mode observer is constructed according to a stator voltage vector and a stator current, and expressed as:
wherein a, b and c are coefficient parameters,to observe the stator current, u s For the measured voltage vector, L smo For the sliding mode coefficient, Z is a switching function, and the superscript k indicates the kth sampling time.
The traditional sliding mode observer has large error under the working condition of low carrier ratio, and analysis shows that the error is mainly formed by discretization error of the traditional sliding mode observer and error caused by quasi-sliding film movement.
In a preferred embodiment, the coefficient parameter is set to:
wherein T is sc Represents the sampling step size, R s The stator resistance is represented, and L the inductance is represented.
In a more preferred embodiment, the sliding mode coefficient L smo The method comprises the following steps:
wherein k is g A tuning parameter greater than 0 and less than 1.
The switching function Z is set to:
wherein Z is 0 For the current error boundary value, set as:
ψ f indicating motor flux linkage omega e Indicating rotor speed, f s Represents the sampling frequency, k z Is a tuning parameter greater than 1.
L provided in the invention smo And Z, such that the current tracking error is limited to within the boundary, thereby making the sliding mode observer more stable and the current error relatively smaller. According to the invention, in S2, the back emf is expressed as:
wherein E is k Representing the back emf,the current error at the kth sampling time is represented, j represents the imaginary part of the complex number, ω e Indicating the rotor speed.
Further, current errorExpressed as:
wherein,representing the stator current observed by the kth sampling instant discrete sliding mode observer,/for the stator current>Representing the stator current measured by the current sensor at the kth sampling instant.
According to the present invention, in S3, the back electromotive force after phase compensation is expressed as:
wherein,for the back EMF after phase compensation, E k To compensate for the pre-back emf, i.e. the back emf obtained in S2.
According to the present invention, in S4, the phase-locked loop based on flux compensation is expressed as:
wherein,representing the estimated rotor speed obtained by the phase-locked loop, < >>Representing the estimated rotor position obtained by the phase-locked loop, h being the phase-locked loop parameter, ψ f Is the motor flux linkage, delta phi f Is a flux linkage compensation term, E d For counter-electromotive force after phase compensation->Projection on d-axis, E q For counter-electromotive force after phase compensation->Projection on q-axis, s denotes Laplace operator.
The estimated position of the traditional phase-locked loop has large error in the dynamic change process, and the phase-locked loop based on the flux linkage compensation provided by the invention has significantly reduced rotor position error in the dynamic change process, and has smaller dynamic position estimated error while keeping the similar steady state performance of the traditional phase-locked loop.
In a preferred embodiment, the phase compensated back emfProjection E on d-axis d Can be expressed as:
E d ＝-ω e ψ f sin(Δθ e )
back electromotive force after phase compensationProjection E on q-axis q Can be expressed as:
E q ＝ω e ψ f cos(Δθ e )
wherein,θ e representing the actual rotor position.
Preferably, the flux linkage compensation term Δψ f Can be obtained by E d Obtained through a PI controller, expressed as:
wherein k is p 、k i And s is a Laplace operator and is a PI controller parameter.
According to a preferred embodiment of the present invention, in S5, the driving control signal of the inverter is obtained using a vector control technique and an SVPWM technique. The vector control technology and the SVPWM technology are common technologies in motor control, and specific processes thereof are not described in detail in the present invention.
The various embodiments of the methods described above in this invention may be implemented in digital electronic circuitry, integrated circuit systems, field Programmable Gate Arrays (FPGAs), application Specific Integrated Circuits (ASICs), application Specific Standard Products (ASSPs), systems On Chip (SOCs), load programmable logic devices (CPLDs), computer hardware, firmware, software, and/or combinations thereof. These various embodiments may include: implemented in one or more computer programs, the one or more computer programs may be executed and/or interpreted on a programmable system including at least one programmable processor, which may be a special purpose or general-purpose programmable processor, that may receive data and instructions from, and transmit data and instructions to, a storage system, at least one input device, and at least one output device.
It should be appreciated that various forms of the flows shown above may be used to reorder, add, or delete steps. For example, the steps described in the present disclosure may be performed in parallel, sequentially, or in a different order, so long as the desired result of the technical solution of the present disclosure is achieved, and the present disclosure is not limited herein.
Examples
Example 1
The simulation experiment is carried out by adopting a permanent magnet synchronous motor sensorless control method, and the method comprises the following steps:
s1, constructing a discrete sliding mode observer;
s2, based on a discrete sliding mode observer, acquiring back electromotive force;
s3, performing phase compensation on the counter electromotive force;
s4, phase locking is carried out on the back electromotive force after compensation by adopting a phase-locked loop based on flux linkage compensation, so that the rotor speed and the rotor position are obtained;
s5, obtaining a driving control signal of the inverter according to the rotor speed and the rotor position, and further controlling the motor.
In S1, the discrete sliding mode observer is expressed as:
the coefficient parameters are set as follows:
the sliding mode coefficient L smo The method comprises the following steps:
the switching function Z is set to:
in S2, the back emf is expressed as:
in S3, the back emf after phase compensation is expressed as:
wherein,is the back electromotive force after phase compensation.
In S4, the phase-locked loop based on flux linkage compensation is expressed as:
flux linkage compensation term Deltapsi f Can be composed of d Obtained through a PI controller, expressed as:
in the simulation process, the switching frequency is set to be 1000Hz, the motor rotating speed is 100Hz, namely the carrier ratio is 10, and the simulation results are shown in figures 2, 4 and 6.
Fig. 2 is a simulation result of a rotor position when the carrier ratio is 10, wherein a blue waveform is an actual rotor position, and a red waveform is an estimated rotor position;
fig. 4 shows the rotor position error at a carrier ratio of 10.
Example 2
The same experiment as in example 1 was performed, except that during the simulation, the motor speed was stepped from 20Hz to 100Hz at 1s of motor start.
The motor dynamic process in the simulation process is shown in fig. 6.
Comparative example
Comparative example 1
The same simulation experiment as in example 1 was performed, except that a conventional sliding mode observer was used, the specific structure of which is described in literature (wish to be new sun, zeng Guohui, huang Bo, etc. permanent magnet synchronous motor vector control [ J ] information and control of the improved sliding mode observer, 2020,49 (06): 708-713+721.) fig. 3 is a simulation result of the rotor position at a carrier ratio of 10, blue waveform is an actual rotor position, and red waveform is an estimated rotor position;
fig. 5 shows the rotor position error at a carrier ratio of 10.
Comparative example 2
The same simulation experiment as in example 2 was performed, except that a conventional phase-locked loop was used, the specific structure of which is described in the literature (zhu Xinyang, zeng Guohui, huang Bo, etc.. Permanent magnet synchronous motor vector control [ J ] information and control for improved sliding mode observer, 2020,49 (06): 708-713+721.).
Fig. 7 is a motor dynamic process in a simulation process.
Comparing fig. 2 and 3, i.e. the rotor positions in example 1 and comparative example 1, it can be seen that under the working condition of carrier ratio of 10, the estimated rotor position of the conventional sliding mode observer in comparative example 1 has a more obvious delay error (fig. 3) compared with the actual rotor position, while in example 1, the blue waveform and the red waveform are well matched (fig. 2), and the error of the discrete sliding mode observer is obviously reduced.
Comparing fig. 4 and fig. 5, i.e., the rotor position errors in example 1 and comparative example 1, it can be seen that the error of the conventional sliding mode observer in comparative example 1 is large and is up to 16 degrees (fig. 5) under the working condition that the carrier ratio is 10, while the error of the discrete sliding mode observer in example 1 is small and is always about 0.3 degrees (fig. 4).
Comparing fig. 6 and fig. 7, i.e. the motor dynamic process in the embodiment 2 and the comparative example 2, it can be seen that the estimated position of the conventional pll in the comparative example 2 has a large error in the dynamic process (fig. 7), and the rotor position error of the pll based on flux linkage compensation in the embodiment 2 in the dynamic process is significantly reduced.
In the description of the present invention, it should be noted that the positional or positional relationship indicated by the terms such as "upper", "lower", "inner", "outer", "front", "rear", etc. are based on the positional or positional relationship in the operation state of the present invention, are merely for convenience of describing the present invention and simplifying the description, and do not indicate or imply that the apparatus or elements referred to must have a specific orientation, be constructed and operated in a specific orientation, and thus should not be construed as limiting the present invention. Furthermore, the terms "first," "second," "third," "fourth," and the like are used for descriptive purposes only and are not to be construed as indicating or implying relative importance.
In the description of the present invention, it should be noted that, unless explicitly specified and limited otherwise, the terms "mounted," "connected," and "connected" are to be construed broadly, and may be, for example, fixedly connected, detachably connected, or integrally connected in common; can be mechanically or electrically connected; can be directly connected or indirectly connected through an intermediate medium, and can be communication between two elements. The specific meaning of the above terms in the present invention will be understood in specific cases by those of ordinary skill in the art.
The invention has been described above in connection with preferred embodiments, which are, however, exemplary only and for illustrative purposes. On this basis, the invention can be subjected to various substitutions and improvements, and all fall within the protection scope of the invention.
Claims (10)
1. The sensorless control method of the permanent magnet synchronous motor is characterized by comprising the following steps of:
s1, constructing a discrete sliding mode observer;
s2, based on a discrete sliding mode observer, acquiring back electromotive force;
s3, performing phase compensation on the counter electromotive force;
s4, phase locking is carried out on the back electromotive force after compensation by adopting a phase-locked loop based on flux linkage compensation, so that the rotor speed and the rotor position are obtained;
s5, obtaining a driving control signal of the inverter according to the rotor speed and the rotor position, and further controlling the motor.
2. The sensorless control method of permanent magnet synchronous motor of claim 1,
in S1, the discrete sliding mode observer is expressed as:
wherein a, b and c are coefficient parameters,to observe the stator current, u s For the measured voltage vector, L smo For the sliding mode coefficient, Z is a switching function, and the superscript k indicates the kth sampling time.
3. The sensorless control method of permanent magnet synchronous motor according to claim 2, characterized in that,
the coefficient parameters are set as follows:
wherein T is sc Represents the sampling step size, R s The stator resistance is represented, and L the inductance is represented.
4. The sensorless control method of permanent magnet synchronous motor according to claim 2, characterized in that,
the sliding mode coefficient L smo The method comprises the following steps:
wherein k is g Setting parameters of more than 0 and less than 1;
the switching function Z is set to:
wherein Z is 0 For the current error boundary value, set as:
ψ f indicating motor flux linkage omega e Indicating rotor speed, f s Represents the sampling frequency, k z Is a tuning parameter greater than 1.
5. The sensorless control method of permanent magnet synchronous motor of claim 1,
in S2, the back emf is expressed as:
wherein E is k Representing the back emf,the current error at the kth sampling time is represented, j represents the imaginary part of the complex number, ω e Indicating the rotor speed.
6. The sensorless control method of permanent magnet synchronous motor of claim 1,
in S3, the back emf after phase compensation is expressed as:
wherein,is the back electromotive force after phase compensation.
7. The sensorless control method of permanent magnet synchronous motor of claim 1,
in S4, the phase-locked loop based on flux linkage compensation is expressed as:
wherein,representing the estimated rotor speed obtained by the phase-locked loop, < >>Representing the estimated rotor position obtained by the phase-locked loop, h being the phase-locked loop parameter, ψ f Is the motor flux linkage, delta phi f Is a flux linkage compensation term, E d For counter-electromotive force after phase compensation->Projection on d-axis, E q For counter-electromotive force after phase compensation->Projection on q-axis, s denotes Laplace operator.
8. The sensorless control method of permanent magnet synchronous motor of claim 7,
flux linkage compensation term Deltapsi f Can be obtained by E d Obtained through a PI controller, expressed as:
wherein k is p 、k i And s is a Laplace operator and is a PI controller parameter.
9. An electronic device, comprising:
at least one processor; and
a memory communicatively coupled to the at least one processor; wherein the memory stores instructions executable by the at least one processor to enable the at least one processor to perform the method of any one of claims 1-8.
10. A computer readable storage medium storing computer instructions for causing the computer to perform the method of any one of claims 1-8.
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| 辛振;赵仁德;郭宝玲;马帅;: "基于二阶广义积分器-锁频环的异步电机同步角频率估计方法", 电工技术学报, no. 01, 26 January 2014 (2014-01-26) * |
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