CN117613526A - Linear topology structure with phase compensation characteristic, design and superconducting filter - Google Patents

Abstract

The embodiment of the application relates to the technical field of filter design and discloses a linear topological structure with phase compensation characteristics, a design and a superconducting filter, wherein the topological structure is based on a standard linear topological structure comprising a source suspension branch, a linear coupling main path and a load suspension branch, four continuous resonators are arbitrarily selected on the linear coupling main path, a first resonator and a fourth resonator in the four continuous resonators are arranged in a cross coupling mode to form a linear topological structure with phase compensation characteristics, the four continuous resonators form a CQ structure, so that the introduction, independent control and adjustment of a real zero point of phase compensation are realized under the condition of fully utilizing the simplicity of the linear topological structure, the group delay in a passband of the filter is compensated while the sideband suppression capability is improved, and the signal fidelity of a communication system is improved.

Description

Linear topology structure with phase compensation characteristic, design and superconducting filter

Technical Field

The embodiment of the application relates to the technical field of filter design, in particular to a linear topological structure with phase compensation characteristics, a design and a superconducting filter.

Background

With the rapid development of modern communication technology and the increasing crowding of the radio spectrum, its bandwidth utilization and interference immunity have met new challenges. As a management portal for spectrum resources, a microwave filter will need to have a narrower bandwidth and a better sideband suppression capability, and thus a filter with high quality factor (Q-value) characteristics will find good use, such as HTS (high temperature superconducting filter, superconducting filter), cavity filter, high Q-value filter chip, and the like. In particular, the high-temperature superconductive film material has extremely low surface resistance in the microwave frequency band, and the planar microwave filter manufactured by using the high-temperature superconductive film material can well meet the requirements of the insertion loss and the filter order of the filter.

By setting CQ (cross-square cross-coupling element), CT (cross-triangular cross-coupling element), extracted pole, NRNs (non-resonant nodes), etc., the sideband suppression degree, sideband roll-off degree of the filter can be improved by introducing cross-coupling between non-adjacent resonators or adding additional stub units to achieve introducing transmission zero, i.e., adding physical structural coupling structural units of the filter, thereby changing the characteristics of the frequency response.

The other topological structure which is simply, conveniently and effectively introduced with the real-frequency transmission zero point is a linear topological structure, and the linear topological structure is particularly suitable for the design of HTS, because the linear topological structure can realize the miniaturization of the volume of a filter, can be well adapted to the low-temperature working environment of a liquid nitrogen temperature zone, effectively reduces the heat load of a refrigerator, has low design difficulty and is easy to meet the precision requirement of a processing technology.

However, the above improvement of sideband suppression will deteriorate the group delay within the filter passband, especially for narrow band filters, which will directly affect the signal fidelity of the communication system.

Disclosure of Invention

An object of the embodiment of the present application is to provide a linear topology structure, design and superconducting filter with phase compensation characteristics, which can implement introduction, independent control and adjustment of a real zero point of phase compensation under the condition of fully utilizing the simplicity of the linear topology structure, and compensate group delay in a passband of the filter while improving sideband suppression capability, thereby improving signal fidelity of a communication system.

To solve the above technical problems, embodiments of the present application provide a linear topology with phase compensation characteristics, which is based on a standard linear topology including a source suspension branch, a linear coupling main circuit, and a load suspension branch. Four continuous resonators are arbitrarily selected on the linear coupling main circuit, and a first resonator and a fourth resonator in the four continuous resonators are arranged to be cross-coupled to form a linear topological structure with phase compensation characteristics; wherein the source hanging branch is provided with n ^S A resonator, the linear coupling main path is provided with n ^D A plurality of resonators, the load hanging branch is provided with n ^L Resonators, n ^D N is an integer greater than 4 ^S And n ^L Are integers not less than 0, and the four continuous resonators form a CQ structure.

The embodiment of the application also provides a design method of the high-temperature superconductive filter, which comprises the following steps:

step 1: determining the type, topological structure and the number of resonators of a response function of the high-temperature superconducting filter according to a preset performance index; the type of the response function is a bounded GC response or a reduced GC response, the topological structure is the linear topological structure with the phase compensation characteristic, and the performance indexes comprise expected center frequency, bandwidth, relative bandwidth, insertion loss, return loss, order, reflection, sideband suppression and group delay of the high-temperature superconducting filter;

step 2: deducing the transmission coefficient and reflection coefficient multiple patterns and the number of zero poles corresponding to the response function based on the topological structure and the number of resonators, and combining the return loss to synthesize the transmission coefficient and reflection coefficient multiple patterns to obtain expressions and zero pole characteristics of the transmission coefficient and reflection coefficient multiple patterns;

Step 3: increasing the coupling coefficient value of cross coupling in the CQ unit from 0, and adjusting the coupling coefficient of adjacent coupling in the CQ unit by an iterative optimization method until the suppression requirement of the group delay is met;

step 4: based on each coupling coefficient in the CQ unit under the requirement of restraining the group delay, adjusting the coupling coefficients of other adjacent couplings in the topological structure until the requirement of the group delay is met, and obtaining a normalized coupling coefficient matrix corresponding to the topological structure at the moment;

step 5: performing inverse normalization on the normalized coupling coefficient matrix to obtain general parameters required by realizing the physical structure size of the filter, wherein the general parameters comprise the physical coupling coefficient for realizing the filter, the quality factor of an input port and the quality factor of an output port;

step 6: determining a distance between the resonators, a first distance between the resonator connected with the source and the first loading tap structure, and a second distance between the resonator connected with the load and the second loading tap structure according to the general parameters;

step 7: and (3) determining the size of the filter circuit according to the calculation result in the step (6), and processing and packaging the filter circuit.

The embodiment of the application also provides a high-temperature superconductive filter, which is designed according to the design method of the high-temperature superconductive filter, and is manufactured on a three-layer structure film substrate with double-sided yttrium barium copper oxide/magnesium oxide/yttrium barium copper oxide, one side of the three-layer structure film substrate is subjected to circuit etching through a standard process of photoetching and ion etching, the other side of the three-layer structure film substrate is used for grounding, and the high-temperature superconductive filter is packaged in a metal shielding box.

The linear topological structure with the phase compensation characteristic, the design and the superconducting filter provided by the embodiment of the application can realize the introduction, independent control and adjustment of the limited frequency transmission zero point and the phase compensation real zero point without arranging an additional unit, and the design of the filter with the linear phase compensation characteristic of multiple transmission zero points can be realized under the condition of ensuring the maximum simplicity of the filter structure, the minimum complexity of the circuit design and the minimum device volume. The designed high-temperature superconductive filter has two pairs of symmetrical limited frequency transmission zero points, and improves the sideband suppression capability; meanwhile, the pair of real zero points are provided for compensating group delay, namely, the group delay in the passband of the filter is compensated while the sideband suppression capacity is improved, so that the signal fidelity of the communication system is improved. In addition, the results of parameters such as in-band insertion loss, band edge roll-off rate and the like of the designed high-temperature superconducting filter are highly matched with theoretical simulation values, and the high-temperature superconducting filter has a great application prospect in a modern wireless communication system.

In some alternative embodiments, M of the CQ structure _i,i+1 、M _i+1,i+2 、M _i+2,i+3 、M _i,i+3 Satisfy |M _i,i+1 ·M _i+1,i+2 ·M _i+2,i+3 ·M _i,i+3 |>0, namely the coupling coefficients of the four pairs of couplings of the CQ structure are not equal to 0; wherein M is _i,i+1 M is the coupling coefficient between the first resonator and the second resonator in the CQ structure _i+1,i+2 M is the coupling coefficient between the second resonator and the third resonator in the CQ structure _i+2,i+3 M is the coupling coefficient between the third resonator and the fourth resonator in the CQ structure _i,i+3 The first resonator in the CQ structure is the ith resonator in the linear topology with phase compensation characteristic, i is larger than n, and the coupling coefficient between the first resonator and the fourth resonator in the CQ structure ^S And is smaller than n ^D +n ^S -an integer of 2.

In some alternative embodiments, 0.ltoreq.n ^S ≤2，0≤n ^L 2 is less than or equal to 0 and n is less than or equal to ^S +n ^L ≤4。

In some alternative embodiments, the resonators in the source suspension branch and the resonators in the load suspension branch are used to determine the frequency location of a finite transmission zero, i.e., a complex zero, and the resonators in the CQ structure are used to determine the frequency location of a real zero.

In some alternative embodiments, the physical coupling coefficients of the filter have the simplicity of being all positive or negative, i.e. the coupling between all resonators uses the same coupling characteristics.

In some alternative embodiments, the high-temperature superconductive filter includes a plurality of resonators, and the resonators are miniaturized resonators of folded half-wavelength microstrip lines fabricated on a film substrate with a three-layer structure of yttrium barium copper oxide/magnesium oxide/yttrium barium copper oxide.

In some alternative embodiments, the resonant frequency of the resonator is controlled by adjusting the length of the resonator, and the coupling coefficient between the two resonators is adjusted by adjusting the distance between the two resonators.

Drawings

One or more embodiments are illustrated by way of example and not limitation in the figures of the accompanying drawings.

FIG. 1 is a schematic diagram of a standard linear topology;

FIG. 2 is a schematic diagram of a linear topology with phase compensation characteristics provided by an embodiment of the present application;

FIG. 3 is a schematic diagram of another linear topology with phase compensation characteristics provided by an embodiment of the present application;

FIG. 4a is a schematic diagram of the S-parameter frequency response curve of an 8-order bounded GC function topology provided by one embodiment of the application;

FIG. 4b is a graph of a group delay frequency response curve of an 8 th order bounded GC function topology provided by one embodiment of the application;

FIG. 5 is a pole-zero distribution plot of an 8-order filter of two pairs of transmission zeros provided in one embodiment of the present application;

FIG. 6a is a schematic diagram of an S-parameter frequency response curve of a reduced GC-10 topology, according to one embodiment of the application;

FIG. 6b is a graph of a group delay frequency response curve of a reduced GC function topology of order 10 provided by one embodiment of the application;

FIG. 7 is a pole-zero distribution plot of a 10-order filter of two pairs of transmission zeros provided by one embodiment of the present application;

FIG. 8 is a flow chart of a method of designing a high temperature superconducting filter according to one embodiment of the present application;

FIG. 9 is a schematic diagram of a miniaturized resonator based on folded half-wavelength microstrip lines provided by one embodiment of the present application;

FIG. 10a is a graph of the characteristics of coupling between adjacent resonators provided by one embodiment of the present application;

FIG. 10b is a graph of the characteristics of cross coupling between non-adjacent resonators provided by one embodiment of the present application;

FIG. 11 is a schematic diagram of the frequency response S-parameters and group delay characteristics of a designed high temperature superconducting filter according to one embodiment of the present application;

FIG. 12 is a schematic diagram of a fabricated high temperature superconducting filter provided in one embodiment of the present application;

FIG. 13 is a graph of S-parameter characteristics measured at 77K for the high temperature superconducting filter of FIG. 12.

Detailed Description

For the purpose of making the objects, technical solutions and advantages of the embodiments of the present application more apparent, the embodiments of the present application will be described in detail below with reference to the accompanying drawings. However, as will be appreciated by those of ordinary skill in the art, in the various embodiments of the present application, numerous technical details have been set forth in order to provide a better understanding of the present application. However, the technical solutions claimed in the present application can be implemented without these technical details and with various changes and modifications based on the following embodiments. The following embodiments are divided for convenience of description, and should not be construed as limiting the specific implementation of the present application, and the embodiments may be mutually combined and referred to without contradiction.

For convenience of explanation of the technical solution of the present application, the content of the standard linear topology, the feature analysis of the high-temperature superconductive filter (which is simply referred to as a linear filter in the visible) designed based on the standard linear topology, the constraint restriction between the transmission zero and the reflection zero, the characteristics of the bounded GC response (Generalized Chebyshev, generalized chebyshev response) and the reduced GC response filter, and the comprehensive analysis of several typical linear filters are introduced.

1. Standard linear topology structure and characteristic analysis of high-temperature superconducting filter designed based on same

A standard linear topology can be shown in FIG. 1, which consists of n coupled in cascade along a straight line ^P Each resonator node, input (source) node and output (load) node, the source (S) and load (L) will be n in topology ^P The resonator nodes are divided into three units, the open small circles in fig. 1 represent the source and load, the black solid dots represent the resonator nodes, and the line segments between the two resonator nodes represent the coupling between the two resonators. From n ^S The first element of the resonator node can be considered as the source suspension, called n ^S Source hanging branches of the steps; the second unit is composed of n ^D The resonator nodes are formed and are called as linear coupling main paths; from n ^L The third element of the resonator node can be regarded as a suspension of the load, called n ^L Load hanging branches of the order, n ^P ＝n ^s +n ^D +n ^L 。

When n is ^S =0 or n ^L When=0, the standard linear topology changes to a single suspended topology; when n is ^S ＝n ^L At=0, the standard linear topology degenerates to the original linear standard structure. In practical engineering applications, n is usually set ^S Not more than 2 and n ^L Not more than 2, and at the same time setting n _Tz ＝n ^S +n ^L ≤4。

Assume that for n corresponding to source S ^S Pole numberWhere n corresponds to load L ^L Pole->Where the source, load branches are equivalently shorted, thus at this n ^S Pole->、n ^L Pole->The input and output admittances of the filter are:

in the method, in the process of the invention,representing source input admittance,/->Representing source output admittance,/->Representing load input admittance,/->The load output admittance is expressed, in other words can be expressed (noted as satisfying the first condition):

k＝0,1,2

l＝0,1,2

wherein S is ₁₁ 、S ₂₂ 、S ₁₂ Are all S parameters.

At this time, admittance Y ^S Pole and Y of (2) ^L The pole of (2) is exactly the n of the filter _TZs ＝n ^S +n ^L Transmission zeros.Is produced by the source->Generated by a load, at n ^D The transmission zero point does not exist in the linear coupling main circuit formed by the resonator nodes. Thus, pole->And pole->Independent of each other, i.e. k and l may take any value or the same value. Although n ^s And n ^L Is in principle arbitrary but is subject to Y when one branch produces two or more transmission zeros ^S And Y ^L At least one of a pair of adjacent poles or transmission zeros of the filter is represented by Y ^S (or Y) ^L ) The effective attenuation of the sideband amplitude can be well realized only by the generated zero point, especially under the condition that two transmission zero points are relatively close. We therefore limit the transmission zeroes generated by each spur to two or less and when one spur generates two transmission zeroes, it is desirable that the two transmission zeroes cannot be on the same side of the stop band or have the same sign (same positive or same negative). Therefore, it is necessary to let n ^S And n ^L Are all smaller than 2, and the design is convenientAnd has been able to meet practical engineering requirements.

The null distribution of the conventional multiple transmission null frequency response exists in several cases:

1) Single zero point

n ^S ＝1，Or n ^L ＝1，/>

2) Symmetrical zero point

n ^S ＝2，Or n ^L ＝2，/>

3) Two zero points of the same sign

n ^S ＝n ^L ＝1，When->When the two zero points are coincident.

4) Zero points of two different symbols

n ^S ＝n ^L ＝1，When->When the two zero points are symmetrical.

5) Three zero points

n ^S ＝2，n ^L ＝1，Or->

Alternatively, n ^S ＝1，n ^L ＝2，Or->

6) Two pairs of symmetrical zero points

n ^S ＝n ^L ＝2，When->When the two pairs of zero points coincide.

For the synthesis of low-pass prototypes of a linear topology, the characteristic polynomials F(s), P(s) and E(s) can be analyzed, defining the scattering parameters of the filter (noted as second condition) as:

wherein complex frequency variable s=jΩ, represents a conjugate matrix, and P(s) is n _TZ Order normalization polynomial, F(s) and E(s) are n ^P And (5) an order normalization polynomial. The roots of F(s) and P(s) are the Reflection Zero (RZs) and Transmission Zero (TZs), respectively. E(s) is a strict Hurwitz polynomialThe root of which is the pole of the filter. E(s), F(s) and P(s) satisfy: epsilon is the coefficient of the transfer function epsilon _R Is the coefficient of the reflection function. Meanwhile, according to the characteristics of the linear filter, the feature polynomial also satisfies the following constraint (noted as a third constraint):

k＝0,1,2

l＝0,1,2

As such, the constraints of the feature polynomials F(s), P(s), and E(s) at the poles are in one-to-one correspondence with the response of a filter based on a standard linear topology. F(s), P(s) and E(s) can meet the above constraints, and can be implemented based on a standard linear topology, and vice versa.

Filters implemented based on standard straight line topologies require a minimum number of coupling elements and no additional elements to generate the transmission zero and no special settings of the coupling elements or resonators, e.g. the coupling between all resonators can be positive or negative.

2. Constraint restriction between transmission zero and reflection zero

For a standard linear topology, under the condition of meeting the above characteristic function polynomial, if the filter response of the generalized chebyshev is to be realized, the condition that epsilon needs to be met is as follows:

wherein RL denotes echo loss, and RL is n ^P And jΩ _Tz In the S parameter of the generalized chebyshev response of a standard linear topology, epsilon has a fixed constraint relation with the echo loss RL and is jointly determined by F (j) and P (j). Thus, for a standard linear topology, the reflection polynomial F(s) is constrained to the transmission polynomial P(s), i.e. the transmission zero j Ω, because the frequency response needs to meet either the first or third condition _Tz (And) And the determination of the reflection zero point are mutually restricted and are not arbitrarily set. In brief, the generalized Chebyshev filter functions TZs and RZs are no longer free to generate, subject to the physical constraints of a standard linear topology.

3. Characteristics of bounded GC response and reduced GC response filter

The level of the RL and the location of the transmission zero in the linear filter of the GC response interact, which results in that no free allocation of RL is allowed. Therefore, the freedom degree of the transmission zero point can be restored by changing the characteristics of the reflection zero point, thereby meeting the requirements of different out-of-band rejection, i.e. relaxing S ₁₁ And S is ₂₂ Is required by the characteristics of the product. Two typical GC responses are a bounded GC response and a reduced GC response, and according to different practical application scenarios, different response schemes of linear topology can be used:

1) When the transmission zeroes are small or there are no fixed requirements, standard GC responses can be used, i.e. all RZs are purely imaginary.

2) When the value of the transmission zero is not large (the transmission zero is closer to the sideband),RZs can be partially or fully degenerated to an imaginary number with a non-zero real part, where RL ε [ RL _min ，RL _max ]Generates a new response as a bounded GC response.

3) When the transmission zero is large (the transmission zero is far from the sideband), one or two RZs can be degenerated to be pure real numbers, the other RZs is kept to be pure imaginary numbers, and at this time, the number of the pure imaginary numbers RZs is reduced, and the generated response is the reduced GC response.

The bounded GC response widens n by bounding the range of the RL when modeling the polynomial ^P And TZs degrees of freedom, which are typically characterized by RL ε [ RL _min ，RL _max ]Wherein the minimum value in the passband (RL _min ) Sum maximum (RL) _max ) Determined by RZs. Within a certain TZs setting, the RL can be adjusted _min And RL(s) _max So that the polynomial of the standard straight-line topology satisfies the third condition described above. There are two limitations to bounded GC response: on the one hand, the response is only applicable to the case of one or a pair of transmission zeroes mentioned above, and when the number of transmission zeroes is 3 or more, it is difficult to find a RL capable of realizing a bounded GC response _min And RL(s) _max Solution of (2); on the other hand, the freedom of the bounded GC response to TZs release is limited, that is TZs cannot be chosen for any frequency location within the stop band. There are two advantages to bounded GC responses at the same time: on the one hand, the sign of the RZ real part does not affect the magnitude of the S parameter (only the phase is determined), which means that when the third condition is implemented, 2n is present ^P The satisfaction of a different F(s) polynomial; on the other hand, in RL ε [ RL ] _min ，RL _max ]Under the condition that the response of the second condition is satisfied, n ^P The real part of RZs can take different values, and the RZs solution is multiple, which increases the selectivity of RL. In view of the bounded GC response of the standard straight line topology, the RL's bounding and RZs polynomials, an optimization method is typically used to obtain an effective matrix function.

Although the bounded GC response increases the degrees of freedom of RZs and TZs to some extent by releasing the range of the RL, there is still room in these allocationsThere are constraints that when the TZs value is relatively large or when there are significant asymmetries or multiple transmission zeros in the desired response, this can result in a particularly large RL value, which is difficult to do in engineering implementations. Therefore, n can be realized by a method of moving a plurality of reflection zero points to a complex plane and solving a reduced GC response function ^P A second order non-chebyshev equal ripple response, wherein the polynomial F(s) has n ^c RZs, n in general ^c Equal to 1 or 2, while the remaining n _Rzi ＝n ^p -n ^c RZs is purely imaginary. Thus, the characteristic function of the reduced GC response has n in the passband _RZi And RL and TZs can be freely set. Wherein, the number n of RZs ^c The selection of (a) may follow the rule that one RZs allows two additional degrees of freedom that may be used to support up to two jΩ generated from any one suspension branch _Tz Is free to set up. In the case of extracting a single or two TZs from a single suspension branch, the addition of one RZs is sufficient. When one or two freely controllable TZs are extracted from each of the two suspension branches, two RZs need to be added. This typically occurs in several situations:

1)n ^c ＝1，n _TZ Not more than 2, wherein n ^L ＝0，n ^S ＝n _Tz Not more than 2, or n ^S ＝0，n ^L ＝n _TZ ≤2。

2)n ^c ＝2，n _TZ 4 or less, wherein n ^L ＝1，n ^S Not more than 2, or n ^S ＝1，n ^L Not more than 2, or n ^L ＝n ^S ＝2。

Wherein "n ^c ＝1，n ^L ＝0，n ^S ＝n _Tz =2 "and" n ^c ＝2，n ^L ＝2，n ^S =2″ is a structure relatively commonly used in engineering, n _TZ Refers to the number of transmission zeros, under which conditions the introduction of transmission zeros can be achieved.

The reduced GC response of the filter is achieved using a standard linear topology, requiring the following steps:

1) According to n _TZ Preliminary of the numerical value of (2)Evaluation of n ^c Is selected from the value of n ^c =1 or n ^c ＝2。

2) Selecting n ^c After the value of (1), assume a real number jΩ _RZ For a fixed initial value, the value of RZs and F(s) are calculated that determine the GC response, notably where F(s) has complex roots and purely imaginary roots.

3) Selecting and updating coefficients epsilon of the reflection function based on the value of RL _R And calculates the coefficient epsilon of the transfer function based on the conditions epsilon above need to satisfy.

4) Using the formulaE(s) is calculated.

5) It is checked whether the current condition satisfies a third condition.

6) If the current condition does not meet the third condition, returning to the step 3) to update epsilon _R And epsilon; and ending the calculation program if the current condition meets the third condition.

So far, based on a standard linear topology, a linear topology with n ^c Plural zero points sum n _RZi ＝n ^p- n ^c A purely imaginary reflection zero, an in-band RL and TZs, freely settable reduced GC frequency response is obtained and the third condition can be met. In practical engineering application, n _TZ 、n ^c RL and jΩ _Tz There are few typical cases, so an efficient solution is easy to obtain. The main advantage of a filter designed based on a standard linear topology is that it can be implemented without special internal elements. Furthermore, the sign of the coupling does not affect the characteristics of the response. The actual size of the linear filter can therefore be realized in exactly the same way as a direct cascade coupled all-pole filter.

4. Comprehensive analysis of several typical linear filters

The standard linear topological structure can effectively introduce a transmission zero omega of a limited frequency through loading of a source or a load _ITz According to the method, we can synthesize a pair of omega _ITz Bounded GC response of 8 th order filter of (2)The corresponding function polynomials and pole-zero characteristics are shown in tables I and II below, where n ^S ＝2、n ^L ＝0、n ^D ＝0， Rl=22 dB. Similarly, we can easily synthesize a solution with two pairs of Ω according to the design flow above _ITz The reduced GC response of the 10 th order filter of (2) corresponds to a functional polynomial and pole-zero characteristics, as shown in tables III and IV below, where n ^S ＝n ^K ＝2，/>Rl=20 dB. Once the polynomial used to represent the linear topology is determined, it can be used to complete the filter's coupling coefficient synthesis.

Table 1: transfer coefficient and reflection coefficient polynomials for an 8 th order filter with a pair of transmission zeroes, rl=22 dB, epsilon= 0.9707

Table 2: zero characteristic of 8-order filter having a pair of transmission zeros

Table 3: transfer coefficient and reflection coefficient polynomials for a 10 th order filter with two pairs of finite zeros, rl=20 dB, epsilon= 0.9138

Table 4: zero characteristic of 10 th order filter with two pairs of transmission zeros

From the above description of the linear filter, it is clear that although the linear filter can improve the sideband suppression effect, this solution worsens the group delay in the passband of the filter, especially for narrow-band filters, which directly affects the signal fidelity of the communication system.

In order to solve the above technical problem that the group delay in the passband of the filter is relatively poor and the signal fidelity of the communication system is not high, an embodiment of the present application proposes a linear topology structure with phase compensation characteristics. The implementation details of the linear topology with phase compensation characteristics of the present embodiment are specifically described below, and the following is merely provided for understanding the implementation details, and is not a necessity for implementing the present embodiment.

The linear topology structure with phase compensation characteristic in this embodiment is based on a standard linear topology structure including a source suspension branch, a linear coupling main path and a load suspension branch, four continuous resonators are selected on the linear coupling main path, and a first resonator and a fourth resonator of the four continuous resonators are disposed to be cross-coupled to form the linear topology structure with phase compensation characteristic. Wherein the source hanging branch is provided with n ^S The linear coupling main circuit is provided with n resonators ^D A resonator, the load hanging branch is provided with n ^L Resonators, n ^D N is an integer greater than 4 ^S And n ^L All are integers not less than 0, and four continuous resonators selected on the linear coupling main circuit form a CQ structure.

The linear topological structure with the phase compensation characteristic provided by the embodiment can realize the introduction, independent control and adjustment of the limited frequency transmission zero point and the phase compensation real zero point without arranging an additional unit, and realize the design of the filter with the linear phase compensation characteristic of multiple transmission zero points under the condition of ensuring the maximum simplicity of the filter structure, the minimum complexity of the circuit design and the minimum device volume. The high-temperature superconductive filter designed based on the topological structure (can be called as a linear type linear phase filter) has two pairs of symmetrical finite frequency transmission zero points, improves the sideband suppression capability, and simultaneously has a pair of real zero points to compensate group delay, namely, the group delay in the passband of the filter is compensated while the sideband suppression capability is improved, so that the signal fidelity of a communication system is improved. In addition, the results of parameters such as in-band insertion loss, band edge roll-off rate and the like of the high-temperature superconducting filter designed based on the topological structure are highly matched with theoretical simulation values, and the high-temperature superconducting filter has a great application prospect in a modern wireless communication system.

Fig. 2 shows a linear topology with phase compensation characteristics, in which n ^S ＝2、n ^D ＝6、n ^L =2, i.e. the source suspension branch is provided with 2 resonators, the straight coupling main is provided with 6 resonators, and the load suspension branch is provided with 2 resonators. The 2, 3, 4, 5 resonators on the selected straight line coupling main circuit (i.e. the total 4, 5, 6, 7 resonators) form a CQ structure. Based on the linear topological structure with the phase compensation characteristic, the 10-order reduced GC response can be realized.

Fig. 3 shows another linear topology with phase compensation characteristics, in which n ^S ＝2、n ^D ＝6、n ^L =0, i.e. the source suspension branch sets 2 resonators, the straight coupling main path sets 6 resonators, and the load suspension branch sets 0 resonators. The 2, 3, 4, 5 resonators on the selected straight line coupling main circuit (i.e. the total 4, 5, 6, 7 resonators) form a CQ structure. Based on the linear type topology with phase compensation characteristicThe flutter architecture can achieve an 8 th order bounded GC response.

In a specific implementation, M of CQ structure _i,i+1 、M _i+1,i+2 、M _i+2,i+3 、M _i,i+3 Satisfy |M _i,i+1 ·M _i+1,i+2 ·M _i+2,i+3 ·M _i,i+3 |>0, i.e. the coupling coefficients of the four pairs of couplings of the CQ structure are not equal to 0, where M _i,i+1 M is the coupling coefficient between the first resonator and the second resonator in the CQ structure _i+1,i+2 For coupling coefficient between second resonator and third resonator in CQ structure, M _i+2,i+3 M is the coupling coefficient between the third resonator and the fourth resonator in the CQ structure _i,i+3 The first resonator in the CQ structure, i.e. the ith resonator in the linear topology with phase compensation characteristics, i is greater than n, is the coupling coefficient between the first resonator and the fourth resonator in the CQ structure ^S And is smaller than n ^D An integer of +nS-2.

That is, the location of the CQ structure is not fixed at a fixed location on the straight coupling main path, such as n ^D The linear coupling main path=6 may be selected to form a CQ structure, or may be selected to form a CQ structure by selecting the 1 st, 2 nd, 3 rd, 4 th and 5 th resonators on the linear coupling main path, or may be selected to form a CQ structure by selecting the 3 rd, 4 th, 5 th and 6 th resonators on the linear coupling main path.

The linear topologies with phase compensation characteristics of FIGS. 2 and 3 select the total 4, 5, 6, 7 resonators to form a CQ structure, i.e., M ₄₅ 、M ₅₆ 、M ₆₇ And M ₄₇ Forming a CQ structure.

In a specific implementation, 0.ltoreq.n in a linear topology with phase compensation characteristics ^S ≤2，0≤n ^L 2 is less than or equal to 0 and n is less than or equal to ^S +n ^L And is less than or equal to 4. Such an arrangement is easy to implement in practical engineering.

In a specific implementation, the resonators in the source suspension branch and the resonators in the load suspension branch are used to determine the frequency location of the finite transmission zero, i.e., the complex zero, and the resonators in the CQ structure are used to determine the frequency location of the real zero.

The linear topology with phase compensation characteristics proposed in this embodiment is subjected to characteristic analysis as follows.

For the linear topology with phase compensation characteristics shown in fig. 3, based on the function polynomials and zero point characteristics in tables 1 and 2, the frequency response that can be extracted is shown in fig. 4a and 4b, fig. 4a is an S-parameter frequency response curve of an 8-order bounded GC function topology, and fig. 4b is a group delay frequency response curve of an 8-order bounded GC function topology. At this time correspond to M ₄₇ Curves for =0. The coupling coefficient of the integrated topology is shown in FIG. 4a, where M ₄₅ ＝0.532、M ₅₆ ＝0.539、M ₆₇ =0.578, the corresponding pole-zero distribution is shown in fig. 5, and j Ω can be seen from fig. 5 _RZ With a real part instead of a pure imaginary part. On the basis of this, if M ₄₇ The value of (2) is increased from 0 to 0.20, while M in the CQ unit is correspondingly adjusted ₄₅ 、M ₅₆ 、M ₆₇ And keeping the other coupling coefficients unchanged in the topology, the corresponding frequency response polynomials and pole-zero characteristics are shown in tables 5 and 6. As can be seen from fig. 3, Ω of the filter _ITz Is unchanged, and two real transmission zero j omega are added _CTz ＝{-0.5936,0.6198}，jΩ _Rz The value of (2) is hardly changed. In addition, as can be seen from fig. 4a, the sideband roll-off of the near sideband is unchanged, the sideband roll-off of the far sideband is slowed down, rl=22 in the passband is unchanged, and as can be seen from fig. 4b, the group delay in the passband can be well compensated. It follows that the finite frequency transmission zero point positionZero pole j omega in passband _Rz And RL in the passband is determined by the topological response, and the CQ unit can independently regulate and control the real zero omega _CTz Is a frequency of (a) is a frequency of (b). Therefore, based on the standard linear topology, the linear topology with phase compensation characteristic obtained by introducing the phase compensation CQ unit can realize the complex zero F of the bounded GC response _ITz And real zero point omega _CTz Is independently introduced and tuned.

Table 5: transfer coefficient and reflection coefficient polynomials for an 8 th order filter with two pairs of transmission zeroes, rl=22 dB, epsilon= 0.9267

Table 6: zero characteristic of 8-order filter with two pairs of transmission zeros

Similarly, for the linear topology with phase compensation characteristics shown in fig. 3, based on the function polynomials and zero point characteristics in tables 3 and 4, the frequency response that can be extracted is shown in fig. 6a and 6b, where fig. 6a is an S-parameter frequency response curve of a reduced GC function topology of 10 th order, and fig. 6b is a group delay frequency response curve of a reduced GC function topology of 10 th order. At this time correspond to M ₄₇ Curves for =0. The coupling coefficient of the integrated topology is shown in FIG. 6a, where M ₄₅ ＝0.526、M ₅₆ ＝0.521、M ₆₇ =0.524, the corresponding pole-zero distribution is shown in fig. 7, where jΩ can be seen in fig. 7 _RZ All are purely imaginary numbers. On the basis of this, if M ₄₇ The value of (2) is increased from 0 to 0.20, while M in the CQ unit is correspondingly adjusted ₄₅ 、M ₅₆ 、M ₆₇ And keeping the other coupling coefficients unchanged in the topology, the corresponding frequency response polynomials and pole-zero characteristics are shown in tables 7 and 8. Likewise, the finite transmission zero Ω of the filter _ITz The position is unchanged, the sideband roll-off of the near sideband is unchanged, and two real number transmission zero j omega are added _CTz = { -0.5661,0.5183}, group delay in passband is compensated, but sideband roll-off of far sideband is slightly slowed down, in passbandIs unchanged rl=20, jΩ _RZ All are purely imaginary numbers. And when M ₄₇ In the process of=0 up to 0.20, the phase state is updated from uncompensated to overcompensated, and the method has universality. From this, it can be seen that, based on the linear topology, the complex zero Ω of the reduced GC response can be achieved by introducing the phase compensated CQ unit _ITz And real zero point omega _CTz Is independently introduced and tuned.

Table 7: transfer coefficient and reflection coefficient polynomials for a 10 th order filter with four pairs of finite zeros, rl=20 dB, epsilon= 0.9486

Table 8: zero characteristic of 10 th order filter with three pairs of transmission zeros

Based on the analysis, the linear phase filter has a finite transmission zero omega _ITz The position is represented by n ^S And n ^L Is determined by the resonator of (a) a real zero omega _CTz Is of frequency position n ^D The CQ structure in (c) and the complementary effect between the two. To this end, a plurality of zero points Ω _ITz And real zero point omega _CTz The position of (2) can be conveniently and freely regulated.

Table 9 shows the coupling coefficients of the CQ structures for the bounded GC functions and the reduced GC functions.

Table 9: coupling coefficients of CQ structures of bounded GC functions and reduced GC functions

Based on this, another embodiment of the present application proposes a design method of a high-temperature superconducting filter, which is applied to a radar and a communication system, and implementation details of the design method of the high-temperature superconducting filter of the present embodiment are specifically described below, which are provided for convenience of understanding only, and are not necessary for implementing the present embodiment.

The specific flow of the design method of the high-temperature superconducting filter of the embodiment may be as shown in fig. 8, which includes:

step 101, determining the type of a response function, a topological structure and the number of resonators of the high-temperature superconducting filter according to a preset performance index, wherein the type of the response function is a bounded GC response or a reduced GC response, the topological structure is a linear topological structure with a phase compensation characteristic, and the performance index comprises the expected center frequency, bandwidth, relative bandwidth, insertion loss, return loss, order, reflection, sideband suppression and group delay of the high-temperature superconducting filter.

Step 102, deducing the transmission coefficient and reflection coefficient multiple patterns and the number of zero poles corresponding to the response function based on the topological structure and the number of resonators, and combining the return loss to synthesize the transmission coefficient and the reflection coefficient multiple patterns to obtain expressions of the transmission coefficient and the reflection coefficient multiple patterns and the zero poles.

At a total order of n ^P ＝n ^S +n ^D +n ^L In the topology of (2), n can be ^S And n ^L Part is seen as a separate unit, n ^D The two parts are regarded as an independent unit, so that the transmission zero points of the two parts can be regulated and controlled independently, and the requirements on the performance of the superconducting filter under different characteristic environments can be met well.

The high-temperature superconductive filter has good advantages in narrow-band and high-sideband suppression scenes by virtue of the characteristic of extremely high quality factors. However, in the design of narrow-band or very narrow-band filtering, the group delay characteristics within the passband deteriorate. Therefore, in the design of a narrow-band high temperature superconducting filter, a linear topology with phase compensation characteristics is the most suitable choice, and a set of preset performance indexes (units shown in brackets) are now given, including:

1) Center frequency (MHz): 4000MHz.

2) Bandwidth, bandwidth (MHz): 80MHz.

3) Relative bandwidth, fractional Bandwidth (%): 2%.

4) Insertion loss, insertion loss (dB): less than or equal to 0.25dB.

5) Order, order of filter:10.

6) Reflection, reflection (dB): less than or equal to-10 dB.

7) Sideband suppression, suppression (dB): and greater than or equal to 50dB.

8) Group delay (ns, @70% of BW): less than or equal to 5ns.

9) Return loss, return loss (dB): 20dB.

Based on the relative bandwidth and sideband suppression in the set of preset performance metrics, the type of response function of the high temperature superconducting filter can be determined to be a reduced GC response. By analysis of the phase, M is taken ₄₇ Normalized coupling coefficient of a set of 10 th order filters of =0.15, where n ^P ＝10，n ^S ＝n ^L ＝2，The function characteristic polynomial at this time is obtained by fine tuning as follows:

P(s)＝[-0.075j，0.032j，-0.268j，0.120j，-0.200j，0.107j，0.046j]；

F(s)＝[1.000，-1.436，1.675，-2.943，0.558，-1.940，-0.242，-0.429，-0.112，-0.0160]；

E(s)＝[1.000，3.891，8.216，13.206-0.118j，15.374-0.153j，14.312-0.163j，10.155-0.128j，5.502，2.117，0.503，0.051]。

step 103, increasing the coupling coefficient value of the cross coupling in the CQ unit from 0, and adjusting the coupling coefficient of the adjacent coupling in the CQ unit by an iterative optimization method until the suppression requirement of group delay is met.

And 104, adjusting the coupling coefficients of other adjacent couplings in the topological structure based on each coupling coefficient in the CQ unit under the requirement of restraining the group delay until the requirement of the group delay is met, and obtaining a normalized coupling coefficient matrix corresponding to the topological structure at the moment.

In a specific implementation, by continuously adjusting each coupling coefficient in the topological structure, the coupling coefficient meeting the suppression requirement of group delay and the coupling coefficient meeting the sideband suppression requirement are found out, and a normalized coupling coefficient matrix corresponding to the topological structure is obtained.

And 105, inversely normalizing the normalized coupling coefficient matrix to obtain general parameters required for realizing the physical structure size of the filter, wherein the general parameters comprise the physical coupling coefficient for realizing the filter, the quality factor of an input port and the quality factor of an output port.

An important step in filter design is to inverse normalize the coupling coefficient matrix to obtain the general parameters needed to achieve the physical structural dimensions of the path filter.

The low-pass prototype may be mapped to a bandpass filter by a transformation given by:

where ω is the angular frequency at low pass, f is the frequency in the bandpass, CF is the center frequency, BW is the bandwidth, and the coupling coefficient (physical coupling coefficient) m under physical structure _ij (i.noteq.j, i.noteq.S, i.noteq.L, j.noteq.S, j.noteq.L) and normalized coupling term M _ij The relationship of (2) is as follows:

the corresponding relation between the quality factors of the input and output ports S and L and the normalized coupling coefficient term is as follows:

i＝2，3

j＝8，9

In which Q _S2 And Q _S3 Q is the quality factor of the input port _L8 And Q _L9 Is the quality factor of the output port.

Thus, the normalized coupling coefficients according to fig. 4b and table 9 above. The general parameters of the 10 th order linear phase filter obtained by calculation through inverse normalization conversion are shown in the following table 10, and the physical coupling coefficients are all positive numbers, namely, the coupling between all resonators adopts the same coupling characteristic. Thus, the filter does not need to achieve negative coupling compared to typical cross-coupling alternatives, which would solve the problems of spurious resonances and process tolerances well.

Table 10: general parameters of a 10 th order linear phase filter

m 12	m 34	m 45	m 56	m 67	m 78
						0.0297	0.0116	0.0101	0.0074	0.0101	0.0117
m 910	m 47	Q S2	Q S3	Q L8	Q L9
						0.0240	0.0030	28.609	71.661	70.291	59.552

And 106, determining the distance between the resonators, the first distance between the resonator connected with the source and the first loading tap structure and the second distance between the resonator connected with the load and the second loading tap structure according to the general parameters.

In designing the high temperature superconducting filter, a miniaturized resonator based on a folded half-wavelength microstrip line as shown in fig. 9 can be used, and the resonator is fabricated on a film substrate having a three-layer structure of yttrium barium copper oxide/magnesium oxide/yttrium barium copper oxide with thickness and dielectric constants of 0.5mm and 9.8, respectively. Wherein the line width of the resonator and the width of the slit are the same and are both 0.1 mm. In FIG. 9, d ₁ ＝0.55,d ₂ ＝0.88,d ₃ ＝0.81,d ₄ ＝0.81,d ₅ ＝0.90,d ₆ ＝0.81,d ₇ ＝0.82,d ₈ ＝0.87,d ₉ =0.58, t=0.45, w=0.80. The resonatorThe resonator has a loop structure, so that adjacent resonators are convenient to couple, and the resonator is a practical topological structure. The resonance frequency of the microstrip resonator depends on the length of the microstrip line, and thus the resonance frequency f of resonance can be controlled by adjusting the length of the resonator ₀ . The amplitude and phase characteristics of the coupling between the two resonators are shown in fig. 10a and 10b, in the case where the external coupling is an extremely weak coupling. Fig. 10a shows the characteristics of coupling between adjacent resonators, and fig. 10b shows the characteristics of cross coupling between non-adjacent resonators. As can be obtained by analysis of the phase change of the transfer function response, the coupling between the two resonators is an electrical coupling, and the coupling coefficient between the two resonators can be calculated by the following formula:

wherein f _p1 And f _p2 The resonant frequencies of the two resonators in the electromagnetic simulation are shown. Then, in FIG. 9, the distance d between resonators _i When (i=1, 2, … 9) becomes larger, the coupling coefficient between the two resonators becomes smaller, whereas the coupling coefficient becomes larger, and therefore the coupling coefficient between the two resonators can be adjusted by adjusting the distance between the resonators, thereby obtaining the general parameters in table 10.

Analysis of the cross coupling characteristics between the two resonators under the condition of weak coupling at the outer edge, as shown in fig. 10b, shows that the two resonators are electrically coupled, and the cross coupling strength between the two resonators can be controlled by adjusting the length of the cross coupling line in fig. 9. Wherein the distance between the cross-coupled line and the resonator is 0.08mm in order to obtain a greater degree of coupling.

The position of the feed port directly affects the unloaded quality factor Q of the resonator _e The value can be calculated using the following formula:

the input/output microstrip line adopts a 50 ohm structure with the line width of 0.46 mm. Notably, in a linear topology, the input ports implement Q simultaneously _S2 And Q _S3 Numerical extraction between the two will affect each other, and the length of the input/output port (named tap) will also interfere with the extraction of the two figures of merit, so that an effective method is to use indirect coupling with less mutual interference.

The specific parameter extraction steps are as follows:

1) The tap is first coupled to the resonator 2 (or 3) and Q is obtained by adjusting the length and distance of the tap _S2 (or Q) _S3 ) Is set to be a constant value.

2) The length of resonator 2 (or 3) is shortened and resonator 3 (or 2) is introduced.

3) Q is obtained by adjusting the distance of the tap from the resonator 3 (or 2) _S3 (or Q) _S2 ) Is set to be a constant value.

4) Restoring the length of the resonator 2 (or 3), shortening the length of the resonator 3 (or 2), and repeating the Q adjustment _S2 (or Q) _S3 )。

Obtaining the effective value Q by iterating step 3) and step 4) _S2 、Q _S3 The method is equally applicable to Q _L8 And Q _L9 Is an extraction of (2). By distinguishing the lengths of the two resonators, the influence of mutual interference between the two resonators is reduced. The length of the tap and the distance between the two resonators are adjusted by three parameters, so that the degree of freedom and feasibility are increased.

The physical dimensions of each part of the filter can be finally constructed by successively calculating the distances corresponding to the coupling coefficients between the two resonators, the distances between the taps and the resonators 2, 3 and 8, 9, and the coupling lengths of the cross-coupled lines. The final overall planar circuit layout, corresponding frequency response S parameters and group delay characteristics are shown as 11. The coupling characteristics between the resonators may all be either electrical or magnetic, which significantly reduces the difficulty of design. The S parameter response of the filter is well matched with the principle calculation, and the group delay can reach an expected value less than or equal to 5 ns.

And 107, determining the size of the filter circuit according to the calculation result of the step 106, and processing and packaging the filter circuit.

Finally, the filter is manufactured on a film substrate with a three-layer structure of double-sided yttrium barium copper oxide/magnesium oxide/yttrium barium copper oxide, and the size is 7.74mm multiplied by 9.66mm (0.258 lambda) _go ×0.322λ _go )，λ _go For a wavelength on the substrate with a center frequency of 4GHz, one side of the film is circuit etched by standard processes of photolithography and ion etching, and the other side is used for grounding. The filter is then packaged in a metal shielding box. The finally produced filter is shown in fig. 12, and fig. 13 shows the S-parameter characteristic measured at a temperature of 77K.

The actual measurement result shows that the manufactured high-temperature superconducting filter has 4 transmission zero points at 3.919GHz, 3.953GHz, 4.046GHz and 4.063GHz, and can well improve sideband rolling reduction. The insertion loss in the passband is only 0.15dB, and the return loss is better than 17.8dB in this frequency range. The center frequency is 4GHz, the relative bandwidth is 2%, the band edge roll-off rate is more than 40dB/GHz, and the group delay fluctuation in 70% bandwidth is less than 5ns. The desired index can be well achieved and the validity of the linear topology with phase compensation characteristics presented herein can be verified.

The above steps of the methods are divided, for clarity of description, and may be combined into one step or split into multiple steps when implemented, so long as they include the same logic relationship, and they are all within the protection scope of this patent; it is within the scope of this patent to add insignificant modifications to the algorithm or flow or introduce insignificant designs, but not to alter the core design of its algorithm and flow.

Another embodiment of the present application relates to a high temperature superconducting filter, wherein the high temperature superconducting filter designed according to the design method of the high temperature superconducting filter is manufactured on a three-layer structure film substrate of double-sided yttrium barium copper oxide/magnesium oxide/yttrium barium copper oxide, one side of the three-layer structure film substrate is subjected to circuit etching through photolithography and ion etching processes, the other side of the three-layer structure film substrate is used for grounding, and the high temperature superconducting filter is packaged in a metal shielding box.

In a specific implementation, the high-temperature superconductive filter comprises a plurality of resonators, wherein the resonators are miniaturized resonators of folded half-wavelength microstrip lines manufactured on a film substrate with a three-layer structure of yttrium barium copper oxide/magnesium oxide/yttrium barium copper oxide.

In a specific implementation, the resonant frequency of the resonator is controlled by adjusting the length of the resonator, and the coupling coefficient between the two resonators is adjusted by adjusting the distance between the two resonators.

It will be understood by those of ordinary skill in the art that the foregoing embodiments are specific embodiments in which the present application is implemented and that various changes in form and details may be made therein without departing from the spirit and scope of the present application.

Claims (9)

1. The linear topological structure with the phase compensation characteristic is characterized in that the linear topological structure is based on a standard linear topological structure comprising a source suspension branch, a linear coupling main circuit and a load suspension branch, four continuous resonators are selected on the linear coupling main circuit, and a first resonator and a fourth resonator in the four continuous resonators are arranged to be in cross coupling to form the linear topological structure with the phase compensation characteristic;

wherein the source hanging branch is provided with n ^S A resonator, the linear coupling main path is provided with n ^D A plurality of resonators, the load hanging branch is provided with n ^L Resonators, n ^D N is an integer greater than 4 ^S And n ^L Are integers not less than 0, and the four continuous resonators form a CQ structure.

2. The linear topology with phase compensation characteristics of claim 1, wherein M of said CQ structure _i,i+1 、M _i+1,i+2 、M _i+2,i+3 、M _i,i+3 Satisfy |M _i,i+1 ·M _i+1,i+2 ·M _i+2,i+3 ·M _i,i+3 |>0, i.eThe coupling coefficients of the four pairs of couplings of the CQ structure are not equal to 0;

wherein M is _|,|+1 M is the coupling coefficient between the first resonator and the second resonator in the CQ structure _i+1,i+2 M is the coupling coefficient between the second resonator and the third resonator in the CQ structure _i+2,i+3 M is the coupling coefficient between the third resonator and the fourth resonator in the CQ structure _i,i+3 The first resonator in the CQ structure is the ith resonator in the linear topology with phase compensation characteristic, i is larger than n, and the coupling coefficient between the first resonator and the fourth resonator in the CQ structure ^S And is smaller than n ^D +n ^S -an integer of 2.

3. The linear topology with phase compensation characteristics according to any of claims 1 to 2, characterized in that 0 n +. ^S ≤2，0≤n ^L 2 is less than or equal to 0 and n is less than or equal to ^S +n ^L ≤4。

4. The linear topology with phase compensation characteristics according to any of the claims 1 to 2, characterized in that the resonators in the source suspension branch and the resonators in the load suspension branch are used for determining the frequency position of a finite transmission zero, i.e. a complex zero, and the resonators in the CQ structure are used for determining the frequency position of a real zero.

5. A method of designing a high temperature superconducting filter, the method comprising:

step 1: determining the type, topological structure and the number of resonators of a response function of the high-temperature superconducting filter according to a preset performance index; wherein the type of the response function is a bounded GC response or a reduced GC response, the topology is a linear topology with phase compensation characteristics as claimed in any one of claims 1 to 4, and the performance index includes a center frequency, a bandwidth, a relative bandwidth, an insertion loss, a return loss, an order, reflection, sideband suppression, and group delay expected by the high-temperature superconducting filter;

step 2: deducing the transmission coefficient and reflection coefficient multiple patterns and the number of zero poles corresponding to the response function based on the topological structure and the number of resonators, and combining the return loss to synthesize the transmission coefficient and reflection coefficient multiple patterns to obtain expressions and zero pole characteristics of the transmission coefficient and reflection coefficient multiple patterns;

step 3: increasing the coupling coefficient value of cross coupling in the CQ unit from 0, and adjusting the coupling coefficient of adjacent coupling in the CQ unit by an iterative optimization method until the suppression requirement of the group delay is met;

Step 4: based on each coupling coefficient in the CQ unit under the requirement of restraining the group delay, adjusting the coupling coefficients of other adjacent couplings in the topological structure until the requirement of the group delay is met, and obtaining a normalized coupling coefficient matrix corresponding to the topological structure at the moment;

step 5: performing inverse normalization on the normalized coupling coefficient matrix to obtain general parameters required by realizing the physical structure size of the filter, wherein the general parameters comprise the physical coupling coefficient for realizing the filter, the quality factor of an input port and the quality factor of an output port;

step 6: determining a distance between the resonators, a first distance between the resonator connected with the source and the first loading tap structure, and a second distance between the resonator connected with the load and the second loading tap structure according to the general parameters;

step 7: and (3) determining the size of the filter circuit according to the calculation result in the step (6), and processing and packaging the filter circuit.

6. The method of claim 5, wherein the physical coupling coefficients of the filter are positive numbers, i.e., the coupling between all resonators uses the same coupling characteristics.

7. A high-temperature superconducting filter, characterized in that the high-temperature superconducting filter designed according to the design method of the high-temperature superconducting filter as claimed in any one of claims 5 to 8 is manufactured on a three-layer structure film substrate of double-sided yttrium barium copper oxide/magnesium oxide/yttrium barium copper oxide, one side of the three-layer structure film substrate is subjected to circuit etching by a standard process of photoetching and ion etching, the other side of the three-layer structure film substrate is used for grounding, and the high-temperature superconducting filter is packaged in a metal shielding box.

8. The high temperature superconductor filter of claim 7, wherein the high temperature superconductor filter comprises a plurality of resonators, the resonators being miniaturized resonators of folded half wavelength microstrip lines fabricated on a film substrate of a three layer structure of yttrium barium copper oxide/magnesium oxide/yttrium barium copper oxide.

9. The high temperature superconducting filter according to claim 8, wherein the resonant frequency of the resonator is controlled by adjusting the length of the resonator, and the coupling coefficient between the two resonators is adjusted by adjusting the distance between the two resonators.