CN117458878A - Cascaded direct current converter and control method thereof - Google Patents
Cascaded direct current converter and control method thereof Download PDFInfo
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- CN117458878A CN117458878A CN202311248906.0A CN202311248906A CN117458878A CN 117458878 A CN117458878 A CN 117458878A CN 202311248906 A CN202311248906 A CN 202311248906A CN 117458878 A CN117458878 A CN 117458878A
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- 230000010363 phase shift Effects 0.000 claims description 12
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Classifications
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
- H02M3/33573—Full-bridge at primary side of an isolation transformer
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0048—Circuits or arrangements for reducing losses
- H02M1/0054—Transistor switching losses
- H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
- H02M1/007—Plural converter units in cascade
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
- H02M1/083—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the ignition at the zero crossing of the voltage or the current
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
- H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
- H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
Abstract
The invention discloses a cascading DC converter suitable for ultra-wide input voltage range and ultra-wide output voltage range application, which comprises: the phase-shifting full-bridge circuit comprises a first capacitor, an inductor, a first bridge arm connected with the first capacitor in parallel, a second bridge arm, a third bridge arm, a second capacitor and an isolation transformer. The cascaded direct current converter and the control method thereof solve the problems that the prior converter needs to use a larger resonant inductor to realize soft switching and can not realize soft switching application of all switching tubes in an ultra-wide input voltage range and an ultra-wide output voltage range. The defect and resonance inductance loss problem caused by the traditional phase-shifting full-bridge circuit are solved, soft switching is realized by all switching tubes in an ultra-wide input voltage range and an ultra-wide output voltage range, the transmission efficiency of the power converter is greatly improved, meanwhile, the switching frequency of the converter can be further improved, and the product volume is reduced.
Description
Technical Field
The invention relates to the field of power supply manufacturing, in particular to a cascade direct current converter circuit suitable for a wide input and wide output voltage range and a control method thereof.
Background
With the development of power supply technology, a high-efficiency and high-power-density power supply conversion circuit has become a trend, a hard switching technology has a larger switching loss during high-frequency switching, the circuit conversion efficiency is reduced, and an electromagnetic interference problem is also serious.
As described in the patent of application number CN200710069828.2, the full-bridge phase-shifting zero-voltage switching converter realizes soft switching by using parasitic parameters of devices on the basis of retaining the advantages of the conventional PWM constant frequency control, and is widely used in large and medium power DC/DC conversion, but the conventional full-bridge phase-shifting zero-voltage switching converter has drawbacks, and mainly appears in:
1. as shown in the figure 1, before the switching tubes Q1 and Q2 of the leading arm are conducted, the phase-shifting full-bridge ZVS converter is easy to realize ZVS of the leading arm because the energy stored by the output inductor Lo of the secondary side of the transformer is large enough to complete the current conversion process of the capacitors C1 and C2 in the leading arm. But before the switching tubes Q3, Q4 of the lagging bridge arm are turned on, the energy required by the circulating process of the parallel capacitors C3, C4 is provided by the resonant inductance Lr. If the resonant inductance Lr is smaller, the stored energy is likely to not complete the current conversion process of the capacitor, and especially under the light load condition, ZVS of the switching tubes Q3 and Q4 of the lagging bridge arm is more difficult to realize; however, the larger resonance inductance Lr will cause serious loss of the duty ratio of the converter, and the output voltage cannot be ensured; in addition, during the period when the switching transistors Q1, Q3 or the switching transistors Q2, Q4 are simultaneously turned on, the resonant inductor Lr has a large circulation in the primary loop due to the fact that the current cannot be suddenly changed, and the circulation does not provide energy to the secondary loop, but generates loss, so that the efficiency of the converter is reduced. Meanwhile, resonance is generated between the resonance inductance Lr and parasitic junction capacitance of the secondary rectifying diode, so that voltage stress on the diode is high, the requirement on electric parameters of the diode is high, the reliability of the system is reduced, and if an absorption circuit is added to improve the stress, the loss of the whole machine is increased, and the efficiency of the converter is reduced; in addition, the improper matching of the dead time of the switching tube and the circuit parameters can also seriously affect the ZVS performance of the system. Therefore, in practical engineering application, the optimal design of the parameters of the phase-shifting full-bridge ZVS converter is a long-standing problem, and the unreasonable parameter design directly affects the overall performance of the system. The circuit is further improved, and the implementation range of the hysteresis bridge arm ZVS can be increased by adopting the saturation inductance to the resonance inductance Lr in fig. 1, but the saturation inductance loss is also larger, and the scheme cannot completely solve all the defects.
2. Ruan Xinbo Yan Angguang, published in the scientific Press of Soft switching technology of pulse Width modulation DC/DC full bridge converter, 1999, book number: the auxiliary circuit controllable full-bridge DC/DC converter proposed by ISBN7-03-007499-8, pages 113 and 114 is shown in fig. 2, and the auxiliary circuit is composed of an inductor Laux and switching tubes Q5 and Q6 and is used for helping a hysteresis bridge arm to realize ZVS. When the switching tube Q4 is ready to be turned off, the switching tube Q5 is turned on, the current of the inductance Laux is linearly increased, the inductance current charges the capacitor C4 and simultaneously discharges the capacitor C3 until the voltage at two ends of the switching tube Q3 is zero, the switching tube Q3 is turned on, and meanwhile the switching tubes Q4 and Q5 are turned off, so that zero voltage on of the switching tube Q3 of a lagging bridge arm is realized; similarly, when the switching tube Q3 is ready to be turned off, the switching tube Q6 is turned on, the current of the inductance Laux is increased linearly, the inductance current charges the capacitor C3 and simultaneously discharges the capacitor C4 until the voltage at two ends of the switching tube Q4 is zero, the switching tube Q4 is turned on, and meanwhile the switching tubes Q4 and Q5 are turned off, so that zero voltage on of the switching tube Q4 of a lagging bridge arm is realized; the scheme has the advantages that under the condition that the leakage inductance of the transformer is small without adding a resonance inductor, the problem of loss of the duty ratio and the problem of device stress generated by the resonance inductor and the junction capacitance oscillation of the output rectifying diode are both improved, the working time of an auxiliary circuit is short, the on-state loss of an auxiliary switching tube and the auxiliary inductor is small, but the defect is that the added auxiliary switching tubes Q5 and Q6 work in a hard switching state, the switching loss is large, and the circuit is limited to be applied under the conditions of high input voltage and high switching frequency.
3. Meanwhile, in order to realize power conversion in an ultra-wide input and output voltage range, a two-stage power supply topology scheme, such as a well-known BOOST circuit boost+full bridge, a BUCK circuit buck+full bridge circuit and the like, is adopted at present, and the scheme has the defects that: 1. the power tube has the defects of high difficulty in realizing soft switching at the same time or needing to additionally add a circuit to realize soft switching, inapplicability to high input voltage application occasion 2, large front-stage output filter capacitor, large bearing ripple of the front-stage output and rear-stage circuit working ripple, large capacitor bearing ripple, large capacitor size, large loss and the like.
Disclosure of Invention
The invention provides a cascading direct current converter and a control method thereof, which are used for solving the problems that the existing converter needs to use a larger resonant inductor to realize soft switching and can not realize soft switching application of all switching tubes in an ultra-wide input voltage range and an ultra-wide output voltage range.
In order to solve the above technical problem, the present invention provides a cascaded dc converter, which includes:
a first capacitor;
the first bridge arm is connected with the first capacitor in parallel;
the full-bridge circuit comprises a second bridge arm and a third bridge arm, wherein the second bridge arm, the third bridge arm and the first bridge arm respectively comprise two switching tubes, and the switching tubes comprise a body diode and a parasitic capacitor;
the isolation transformer, one end of the primary winding of the isolation transformer is connected with the bridge arm midpoint A of the second bridge arm, and the other end of the primary winding of the isolation transformer is connected with the bridge arm midpoint B of the third bridge arm;
one end of the inductor is connected with a bridge arm midpoint C of the first bridge arm, and the other end of the inductor is connected with a bridge arm midpoint A of the second bridge arm;
and the second capacitor is respectively connected with the second bridge arm and the third bridge arm in parallel.
Preferably, the second bridge arm of the full-bridge circuit has a switching tube Q3 and a switching tube Q4, and the driving signal of the switching tube Q3 is complementary to the driving signal of the switching tube Q4; the third bridge arm has a switching tube Q5 and a switching tube Q6, the driving signal of the switching tube Q5 and the driving signal of the switching tube Q6 are complementary, and the duty ratio of the driving signal of the switching tube Q3 is the same as the duty ratio of the driving signal of the switching tube Q6.
Preferably, the second bridge arm of the full-bridge circuit has a switching tube Q3 and a switching tube Q4, and the driving signal of the switching tube Q3 is complementary to the driving signal of the switching tube Q4; the third bridge arm is provided with a switching tube Q5 and a switching tube Q6, the driving signals of the switching tube Q5 and the switching tube Q6 are complementary, and the duty ratio of the driving signal of the switching tube Q3 is different from that of the driving signal of the switching tube Q6.
The invention also provides a control method of the cascading direct current converter, wherein the first bridge arm is provided with a switching tube Q1 and a switching tube Q2, the second bridge arm is provided with a switching tube Q3 and a switching tube Q4, and the third bridge arm is provided with a switching tube Q5 and a switching tube Q6; according to the current change of the inductor, the cascaded direct current converter is divided into four phases in one working period, wherein the four phases are an input voltage phase, an input output voltage phase, a free wheel phase and a clamping phase in sequence; in the input voltage stage, a switching tube Q1 and a switching tube Q4 are turned on, and a switching tube Q2 and a switching tube Q3 are turned off; in the stage of input and output voltage, the switching tube Q1 is switched on, the switching tube Q2 is switched off, the switching tube Q3 is switched on, and the switching tube Q4 is switched off; in the freewheel phase, the switching tube Q2 and the switching tube Q3 are turned on, and the switching tube Q1 and the switching tube Q4 are turned off; in the clamping stage, the switching tube Q2 and the switching tube Q4 are turned on, the switching tube Q1 and the switching tube Q3 are turned off, and the control method comprises the following steps:
the voltage of the second capacitor is adjusted by adjusting the duration of at least one of the input voltage phase, the input output voltage phase and the freewheel phase, so that the output voltage of the cascade direct current converter is adjusted.
Preferably, the voltage of the second capacitor is adjusted by adjusting the duration of at least one of the input voltage stage, the input output voltage stage and the freewheel stage, and the phase shift angle between the second bridge arm and the third bridge arm is also adjusted at the same time, so as to realize the adjustment of the output voltage of the cascaded direct current converter.
The invention further provides a control method of the cascade direct current converter, wherein the second bridge arm is a lagging bridge arm with a phase lagging relative to the third bridge arm, and the third bridge arm is a leading bridge arm with a phase leading relative to the second bridge arm;
the output voltage of the cascade direct current converter is regulated by regulating the phase shift angle between the leading bridge arm and the lagging bridge arm.
Compared with the prior art, the invention has the following beneficial effects:
1. the cascade direct current converter can enable the switching tube to fully work in a zero-voltage on state without resonance inductance, so that the problem that the duty ratio of the converter is seriously lost due to the arrangement of resonance inductance is avoided, and the cascade direct current converter is applicable to the application scenes of ultra-wide high input voltage, ultra-wide output voltage and high switching frequency conversion;
2. through the cascade direct current converter, the output voltage can have various regulation modes: 1) Adjusting the voltage of the second capacitor by adjusting the duration of at least one of an input voltage phase, an input output voltage phase and a freewheel phase in the duty cycle of the DC converter; 2) Adjusting working phases of the second bridge arm and the third bridge arm; 3) And adjusting the voltage of the second capacitor and simultaneously adjusting the phase shift angle between the second bridge arm and the third bridge arm.
Drawings
FIG. 1 is a schematic diagram of a prior art scheme transfer full bridge circuit;
FIG. 2 is a schematic diagram of a prior art auxiliary circuit controllable ZVS phase-shifted full bridge;
FIG. 3 is a schematic diagram of a cascaded DC converter of the present invention;
FIG. 4 is a timing diagram illustrating operation of a cascaded DC converter according to the present invention;
fig. 5 is a waveform diagram of inductor current of the cascaded dc converter of the present invention.
Detailed Description
The invention and its advantageous effects will be described in further detail below with reference to the detailed description and the accompanying drawings, but the detailed description of the invention is not limited thereto.
Embodiments of the invention are described in further detail below with reference to the attached drawing figures:
referring to fig. 3, the present embodiment discloses a cascaded dc converter (hereinafter referred to as a converter) applicable to a wide input and wide output voltage range ZVS, which is composed of an input capacitor Cin (first capacitor), a first bridge arm, an inductor L1, a capacitor Cm (second capacitor), a second bridge arm, a third bridge arm, a transformer T1 (including an excitation inductor Lm), and an output rectifying filter circuit, wherein when two switching tubes in each set of bridge arms are turned on, dead time is inserted between the two switching tubes to prevent common. The first bridge arm comprises a switching tube Q1 and a switching tube Q2, the second bridge arm comprises a switching tube Q3 and a switching tube Q4, the third bridge arm comprises a switching tube Q5 and a switching tube Q6, and the switching tube Q3 in the second bridge arm, the switching tube Q4 in the second bridge arm and the switching tube Q5 in the third bridge arm and the switching tube Q6 form a full-bridge circuit.
In this embodiment, the driving signal of the switching tube Q3 and the driving signal of the switching tube Q4 are complementary and have a dead time therebetween; the driving signal of the switching tube Q5 and the driving signal of the switching tube Q6 are complementary and have a dead time, and the duty ratio of the driving signal of the switching tube Q3 is the same as the duty ratio of the driving signal of the switching tube Q4, and the duty ratio of the driving signal of the switching tube Q5 is the same as the duty ratio of the driving signal of the switching tube Q6. In other embodiments, the duty cycle of the driving signal of the switching transistor Q3 may be different from the duty cycle of the driving signal of the switching transistor Q4, and the duty cycle of the driving signal of the switching transistor Q5 may be different from the duty cycle of the driving signal of the switching transistor Q6.
The output rectifying and filtering circuit is positioned on the secondary side of the transformer T1, is a full-wave rectifying topology with a center tap and comprises: two transformer secondary windings, a rectifier diode D7, a rectifier diode D8, an output filter inductance Lo and an output filter capacitance Co, wherein the rectifier diodes D7, D8 may be replaced by switching tubes to reduce conduction losses, a technique commonly referred to as synchronous rectification. In other embodiments, the output rectifying and filtering circuit may employ other topologies than conventional center-tapped full-wave rectifying topologies, such as current-multiplying rectification, full-bridge rectification, and the like.
The working principle of the inverter of the present invention will be described with reference to fig. 4. Wherein Vgs1 to Vgs6 are driving signal waveforms of the switching transistors Q1 to Q6; i L1 For the waveform of the inductance L1 current, I P A primary side current waveform of the transformer T1; u (U) AB Is a voltage waveform between a midpoint A of the second bridge arm and a midpoint B of the third bridge arm; i D7 Current waveform of rectifier diode D7, I D8 The current waveform of the rectifier diode D8.
The specific working process is briefly described as follows:
the converter of the invention comprises the following processes in one switching cycle:
time t0 to t 1: the switching tube Q1, the switching tube Q4 and the switching tube Q5 are conducted, the switching tube Q2, the switching tube Q3 and the switching tube Q6 are turned off, the inductor L1 stores energy through Vin, the switching tube Q1, the inductor L1 and the switching tube Q4 form a loop, the primary winding Np of the transformer T1 is reversely excited by the exciting inductor Lm through the capacitor Cm, the switching tube Q5 and the switching tube Q4 form a loop, the rectifier diode D8 is conducted, and the rectifier diode D7 is cut off to output energy for the output filter inductor Lo.
time t1 to time t 2: the switching tube Q2, the switching tube Q3 and the switching tube Q6 are continuously turned off, the switching tube Q1 and the switching tube Q4 are continuously turned on, the inductor L1 is continuously used for storing energy, the switching tube Q5 is turned from on to off, the exciting voltage of the primary winding of the transformer is turned off, the rectifying diode D8 is continuously turned on due to the fact that the current of the output filter inductor Lo cannot be suddenly changed, the current direction of the primary winding Np of the transformer T1 is unchanged, secondary current is reflected to the current according to the turn ratio of the transformer T1, the current charges the capacitor C5, the capacitor C6 is discharged until the diode D6 is turned on, the switching tube Q6 is turned on, zero voltage is realized, the diode D8 is continuously turned on until the voltage applied to the exciting inductor Lm is approximately 0, and the rectifying diode D7 is turned on and the current Ip is 0.
time t2 to time t 3: the switching tube Q1, the switching tube Q4, the switching tube Q6 is conducted, the switching tube Q2 is conducted, the switching tube Q3 and the switching tube Q5 are turned off, the switching tube Q1 and the switching tube Q4 are continuously conducted, the inductor L1 is used for continuously storing energy, the switching tube Q4 and the switching tube Q6 are conducted, the primary winding Np of the transformer T1 is short-circuited, the secondary transformer winding is also short-circuited at the moment, and the output filter inductor Lo is conducted through the rectifier diodes D7 and D8 in a follow current mode to provide energy for the output filter capacitor Co.
time t3 to time t 4: the switching tube Q1 and the switching tube Q6 are conducted, the switching tube Q2, the switching tube Q3 and the switching tube Q5 are turned off, the switching tube Q4 is turned off from on, the inductor L1 charges the capacitor C4, the capacitor C3 discharges until the voltage at the point A rises to be higher than the voltage of the capacitor Cm, the body diode of the switching tube Q3 is conducted, the switching tube Q3 is conducted at the moment, the switching tube Q3 is conducted to achieve zero voltage on, meanwhile, the primary winding Np of the transformer T1 enables the exciting inductor Lm to be excited positively along with the rising of the voltage of the capacitor C4, the rectifying diode D8 is turned off, the rectifying diode D7 is conducted continuously, and the conversion of the secondary diode is completed.
time t4 to time t 5: the switching tube Q1, the switching tube Q3 and the switching tube Q6 are conducted, the switching tube Q2, the switching tube Q4 and the switching tube Q5 are turned off, the inductor L1 current charges the capacitor Cm through the switching tube Q3, meanwhile, the switching tube Q6 is conducted, the inductor L1 current synchronously transmits energy to the secondary output through the rectifier diode D7 through the transformer T1.
time t5 to t 6: the switching tube Q3 and the switching tube Q6 are conducted, the switching tube Q2, the switching tube Q4 and the switching tube Q5 are turned off, the switching tube Q1 is turned off from on, the inductor L1 current freewheels to charge the capacitor C1, the capacitor C2 discharges until the voltage of the capacitor C2 is approximately zero, the diode D2 is turned on, and the switching tube Q2 achieves zero voltage on until the moment t 7.
time t6 to time t 7: the switching tube Q2, the switching tube Q3 and the switching tube Q6 are conducted, the switching tube Q1, the switching tube Q4 and the switching tube Q5 are turned off, the inductor L1 continuously charges the capacitor Cm through the switching tube Q3 in a follow current mode, and meanwhile energy is output to the secondary through the transformer T1 and the switching tube Q6.
time t7 to time t 8: the switching tube Q2 and the switching tube Q3 are conducted, the switching tube Q1, the switching tube Q4 and the switching tube Q5 are turned off, the switching tube Q6 is turned off from on, at the moment, the current of the inductor L1 and the voltage of the capacitor Cm are discharged to the capacitor C5 through the transformer T1 together through the switching tube Q3, and the capacitor C6 is charged until the diode D5 is conducted and the switching tube Q5 is turned on to realize zero-voltage on.
time t8 to t 9: the switching tube Q2, the switching tube Q3 and the switching tube Q5 are conducted, the switching tube Q1, the switching tube Q4 and the switching tube Q6 are turned off, after the current of the inductor L1 is freewheeled to zero through the switching tube Q3, the inductor L1 forms a loop through the capacitor Cm, the switching tube Q3 and the switching tube Q2, the inductor L1 is reversely excited to the maximum, the exciting inductor Lm current Ip is gradually reduced to 0, the primary winding of the transformer T1 is clamped through the switching tube Q3 and the switching tube Q5, and the inductor L1 is added to the primary winding U of the transformer AB The voltage is 0 volts.
time t9 to t 10: the switching tube Q2 and the switching tube Q5 are switched on, the switching tube Q1, the switching tube Q4 and the switching tube Q6 are switched off, the switching tube Q3 is switched from on to off, and the inductance L1 current I L1 Charging capacitor C3, discharging capacitor C4, switching tube Q4 is turned on at zero voltage and negative voltage U AB Exciting the exciting inductance Lm of the transformer T1 to secondarilyThe stage transfers energy, rectifying diode D7 is off and rectifying diode D8 is on.
time t10 to time t 11: the switching tube Q2, the switching tube Q4 and the switching tube Q5 are turned on, the switching tube Q1, the switching tube Q3 and the switching tube Q6 are turned off, at the moment, the switching tube Q2 and the switching tube Q4 are turned on, and the inductive current I L1 Clamp is kept unchanged, inductor current I L1 The direction is negative (point A to point C), and meanwhile, the switching tube Q5 and the switching tube Q4 are conducted and continuously add negative voltage U AB Exciting the exciting inductance Lm of the transformer T1 to transfer energy to the secondary, the rectifying diode D8 is turned on, and the rectifying diode D7 is turned off.
t11 to the next period t0 time: at this time, the switching tube Q2 is turned from on to off, at this time, the inductor L1 charges the capacitor C2, and discharges the capacitor C1 until the diode D1 is turned on, at this time, the switching tube Q1 is turned on to realize zero-voltage on. The working state enters the next period to continue the circulation work.
In the working process, the two switching tubes of each of the second bridge arm and the third bridge arm are conducted in a 180-degree complementary mode, the driving signals of the corresponding switching tubes of the two bridge arms are different by one phase, namely the phase shifting angle, and the output voltage is regulated by regulating the phase shifting angle. Here, the driving signals of the switching tube Q5 and the switching tube Q6 respectively advance the driving signals of the switching tube Q3 and the switching tube Q4, so that the third bridge arm formed by the switching tube Q5 and the switching tube Q6 is called an advanced bridge arm, and the second bridge arm formed by the switching tube Q3 and the switching tube Q4 is called a retarded bridge arm. In FIG. 4, [ t6-t9 ]]The phase difference corresponding to the time period is the phase shift angle delta, and the magnitude of the phase shift angle delta is(Ts is one switching period of the full-bridge circuit formed by the second bridge arm and the third bridge arm). The smaller the phase shift angle delta, the higher the output voltage; conversely, the larger the phase shift angle δ, the lower the output voltage. According to the invention, the output voltage of the cascade direct current converter can be regulated by regulating the phase shift angle delta of the third bridge arm (leading bridge arm) and the second bridge arm (lagging bridge arm).
In another embodiment, the voltage of the capacitor Cm may be adjusted to achieve the adjustment of the output voltage of the cascaded dc converter, and the adjustment process of the voltage of the capacitor Cm is described below with reference to fig. 3 and 5.
According to the current change of the inductor, the converter is divided into four phases in one working period, wherein the four phases are an input voltage phase T1, an input output voltage phase T2, a free wheel phase T3 and a clamping phase T4 in sequence. The voltage of the capacitor Cm is adjusted by adjusting the duration of at least one of the input voltage stage T1, the input output voltage stage T2 and the freewheel stage T3, so that the output voltage of the cascaded direct current converter is adjusted.
In the input voltage phase T1, the inductor current I L1 With slopeAscending; in the input/output voltage stage T2, the inductor current I L1 With slope +.>Ascending; in the freewheel phase T3, the inductor current I L1 With slope +.>Falling to a negative value, in the clamping phase T4, the inductor current I L1 Remain unchanged. In fig. 5, the hatched area represents the output energy transfer time, and at least one of the time length ratios (the minimum time in the T4 stage can be adjusted to 0) of T1, T2, T3 is adjusted by adjusting the switching time sequences (refer to fig. 4) of the switching transistors Q1, Q2, Q3, Q4, so that the output energy is continuously adjusted, and the voltage level of the capacitor Cm is further adjusted.
The following is a list of descriptions of the current phase of the corresponding inductor L1 for the switching time sequence state of a complete duty cycle:
| working phase | Q1 | Q2 | Q3 | Q4 |
| T1 (input voltage stage) | Conduction | Shut off | Shut off | Conduction |
| T2 (input output voltage stage) | Conduction | Shut off | Conduction | Shut off |
| T3 (freewheel phase) | Shut off | Conduction | Conduction | Shut off |
| T4 (clamping stage) | Shut off | Conduction | Shut off | Conduction |
In yet another embodiment, the voltage of the capacitor Cm may be adjusted, and the phase shift angle δ of the second bridge arm and the third bridge arm may be adjusted to adjust the output voltage.
Compared with the prior art, the invention has the following beneficial effects:
1. according to the analysis of the working principle, all the switching tubes can work in a zero-voltage switching mode under all working conditions without additional auxiliary devices and limiting conditions, the converter is suitable for high-voltage input, the switching frequency is high, the power density of the product is greatly improved, and meanwhile, the switching loss and the EMI noise are reduced.
2. From the above analysis, the inductor current I L1 During the phase of T2 (input-output voltage phase), the inductor L1 and the capacitor Cm transmit energy to output through the transformer, so that the intermediate bus filter capacitor Cm of the converter has smaller filter pressure, the same output voltage ripple can be realized under the same working condition, the capacitor with smaller volume can be selected, meanwhile, the charge-discharge current of the capacitor Cm is reduced through the switching tubes Q3 and Q5, the loss is lower, and the converter efficiency is higher.
3. The output voltage has a number of regulation modes: 1. the switching time sequence of the switches Q1, Q2, Q3 and Q4 is regulated, and the working time of each stage of the inductor L1 is regulated, so that the voltage of the capacitor Cm is regulated, and the output voltage is regulated; 2. the output voltage is regulated by regulating the working phases of the second bridge arm and the third bridge arm; 3. the voltage of the regulating capacitor Cm and the phase shift angle delta of the second bridge arm and the third bridge arm are regulated together to regulate the output voltage, so that the ultra-wide input and output voltage range conversion is realized.
4. Reference is made to the principle of phase-shifting full bridge operation: under the high-frequency condition, the influence of leakage inductance on the whole system is larger, the leakage inductance is mainly reflected in that the loss of the system is increased, the back electromotive force is generated by the leakage inductance at the moment of switching off, the overvoltage breakdown of a switching tube is easy to cause, the leakage inductance forms an oscillation loop together with a distributed capacitor (especially a parasitic capacitor of a rectifier diode) in a circuit and a distributed capacitor of a transformer coil, the oscillation is generated, the voltage stress of a secondary rectifier diode is large, electromagnetic energy is radiated outwards, the absorption circuit is restrained to reduce the efficiency of a converter, the problem of loss of a duty ratio is solved, and the like.
It should be emphasized that the examples described herein are illustrative rather than limiting, and that this invention includes but is not limited to the examples described in the detailed description, and that various modifications, which do not depart from the gist of the invention, are intended to be within the scope of the invention as well as other embodiments which are derived from the technical solution of the invention by a person skilled in the art.
Claims (6)
1. A cascaded dc converter, characterized by: it comprises the following steps:
a first capacitor;
the first bridge arm is connected with the first capacitor in parallel;
the full-bridge circuit comprises a second bridge arm and a third bridge arm, wherein the second bridge arm, the third bridge arm and the first bridge arm respectively comprise two switching tubes, and the switching tubes comprise a body diode and a parasitic capacitor;
one end of a primary winding of the isolation transformer is connected with a bridge arm midpoint A of the second bridge arm, and the other end of the primary winding of the isolation transformer is connected with a bridge arm midpoint B of the third bridge arm;
one end of the inductor is connected with a bridge arm midpoint C of the first bridge arm, and the other end of the inductor is connected with a bridge arm midpoint A of the second bridge arm;
and the second capacitor is respectively connected with the second bridge arm and the third bridge arm in parallel.
2. The cascaded dc converter of claim 1, wherein: the second bridge arm of the full-bridge circuit is provided with a switching tube Q3 and a switching tube Q4, and the driving signal of the switching tube Q3 is complementary with the driving signal of the switching tube Q4; the third bridge arm has a switching tube Q5 and a switching tube Q6, the driving signal of the switching tube Q5 is complementary to the driving signal of the switching tube Q6, the duty ratio of the driving signal of the switching tube Q3 is the same as the duty ratio of the driving signal of the switching tube Q4, and the duty ratio of the driving signal of the switching tube Q5 is the same as the duty ratio of the driving signal of the switching tube Q6.
3. The cascaded dc converter of claim 1, wherein: the second bridge arm of the full-bridge circuit is provided with a switching tube Q3 and a switching tube Q4, and the driving signal of the switching tube Q3 is complementary with the driving signal of the switching tube Q4; the third bridge arm is provided with a switching tube Q5 and a switching tube Q6, the driving signals of the switching tube Q5 and the driving signals of the switching tube Q6 are complementary, the duty ratio of the driving signals of the switching tube Q3 is different from that of the driving signals of the switching tube Q4, and the duty ratio of the driving signals of the switching tube Q5 is different from that of the driving signals of the switching tube Q6.
4. A control method of the cascaded dc converter of claim 1, wherein: the first bridge arm is provided with a switching tube Q1 and a switching tube Q2, the second bridge arm is provided with a switching tube Q3 and a switching tube Q4, and the third bridge arm is provided with a switching tube Q5 and a switching tube Q6; according to the current change of the inductor, the cascaded direct current converter is divided into four phases in a working period, wherein the four phases are an input voltage phase, an input and output voltage phase, a free wheel phase and a clamping phase in sequence; in the input voltage stage, a switching tube Q1 and a switching tube Q4 are turned on, and a switching tube Q2 and a switching tube Q3 are turned off; in the input and output voltage stage, the switching tube Q1 is switched on, the switching tube Q2 is switched off, the switching tube Q3 is switched on, and the switching tube Q4 is switched off; in the freewheel phase, the switching tube Q2 and the switching tube Q3 are turned on, and the switching tube Q1 and the switching tube Q4 are turned off; in the clamping stage, the switching tube Q2 and the switching tube Q4 are turned on, the switching tube Q1 and the switching tube Q3 are turned off, and the control method comprises the following steps:
and adjusting the voltage of the second capacitor by adjusting the duration of at least one of the input voltage stage, the input and output voltage stage and the free wheel stage, so as to further realize the adjustment of the output voltage of the cascaded direct current converter.
5. A control method of the cascaded dc converter as set forth in claim 4, characterized in that: the voltage of the second capacitor is regulated by regulating the duration of at least one of the input voltage stage, the input and output voltage stage and the free wheel stage, and the phase shift angle between the second bridge arm and the third bridge arm is regulated at the same time, so that the regulation of the output voltage of the cascaded direct current converter is realized.
6. A control method of the cascaded dc converter of claim 1, wherein: the second bridge arm is a lagging bridge arm with a phase lagging relative to the third bridge arm, and the third bridge arm is a leading bridge arm with a phase leading relative to the second bridge arm;
and adjusting the output voltage of the cascaded direct current converter by adjusting the phase shift angle between the leading bridge arm and the lagging bridge arm of the full-bridge circuit.
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| CN202311248906.0A CN117458878A (en) | 2023-09-26 | 2023-09-26 | Cascaded direct current converter and control method thereof |
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| CN202311248906.0A CN117458878A (en) | 2023-09-26 | 2023-09-26 | Cascaded direct current converter and control method thereof |
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