CN116667347A - Harmonic current control method for active power filter - Google Patents
Harmonic current control method for active power filter Download PDFInfo
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Classifications
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J3/00—Circuit arrangements for ac mains or ac distribution networks
- H02J3/01—Arrangements for reducing harmonics or ripples
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/12—Arrangements for reducing harmonics from ac input or output
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/483—Converters with outputs that each can have more than two voltages levels
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
- H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
- H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
- Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
- Y02E40/20—Active power filtering [APF]
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- Supply And Distribution Of Alternating Current (AREA)
Abstract
The invention relates to a harmonic current control method of an active power filter, which comprises the following steps: 1) Acquiring a current instantaneous value of a nonlinear load, sending the instantaneous value into a current inner loop module, simultaneously acquiring a public coupling point voltage, and sending the voltage into a voltage outer loop module; 2) Performing current inner loop control; 3) Performing voltage outer loop control; 4) And 5) carrying out midpoint balance strategy control, namely adding the modulated wave of the harmonic current, the modulated wave of the direct current bus voltage-stabilizing ring and the modulated wave of the direct current bus voltage-stabilizing ring to obtain a total modulated wave, inputting the total modulated wave into an SPWM link, and controlling the states of all switching tubes of the inverter. Compared with the prior art, the invention has the advantages of high flexibility, no need of harmonic detection link, high supplementing precision and the like.
Description
Technical Field
The invention relates to the technical field of power quality and power electronic control, in particular to a harmonic current control method of an active power filter.
Background
Aiming at the problem of restraining the power grid harmonic waves, in 1969, B.M.bird et al firstly put forward the idea of an active power filter (active power filter, APF), namely a method for enabling the total harmonic waves and reactive currents of the power grid to be zero by injecting compensation currents with the same magnitude and opposite directions as the original harmonic waves and reactive currents into the power grid. The highest output current of a single APF complete machine of the current NC AH active filter can reach 600A, and the harmonic compensation effect reaches the advanced level at home and abroad.
The harmonic current detection technology commonly used by APF is mainly divided into two major categories, namely frequency domain and time domain. The most widely applied in the time domain is i based on instantaneous reactive power theory p -i q The method has the advantages of insensitivity to power grid voltage distortion and quick dynamic response. The literature improves the system so that the system can independently extract specific times of harmonic waves and can be applied to a single-phase system. The most widely used frequency domain methods are FFT, DFT and SDFT. The FFT and DFT methods calculate after collecting harmonic current data of a fundamental wave period, have inherent delay of the fundamental wave period, and have consistent results of the two algorithms, but the FFT algorithm can achieve the purpose of simplifying operation after calculating all subharmonic results, and the harmonic current detection usually only needs to calculate limited subharmonic, so the FFT calculated amount is larger than the DFT, and the two methods are suitable for occasions with low requirements on most dynamic performances; the SDFT algorithm can have a dynamic response smaller than the fundamental period when extracting for a specific subharmonic, but the algorithm has error accumulation and needs to operate after each sampling. For the Fourier transform method in the frequency domain, the problems of fence and spectrum leakage exist, and the influence of the method can be reduced by a window function method. In addition, the Kalman filter method, the wavelet transform method and the like can be applied to harmonic current detection, but are limited by the problems of complex algorithm, slow dynamic response and excessive calculation amount, and the methods are generally applied to off-line detectionIn the above, the method is not suitable for the situation with high real-time requirements such as APF. Because the controlled harmonic current comprises a plurality of frequency sine waves, in order to enable the compensation current output by the APF to quickly and accurately track the given value after harmonic current detection, the scholars propose various high-performance harmonic current control technologies, such as traditional PI, PR, RC, sliding mode control, predictive control, compound control combining the control methods and the like. But is limited by the contradiction between the computational power and algorithm complexity of the controller, and the actual engineering products still mainly adopt the control or optimization improvement scheme based on PI+PR/RC at present. Therefore, in future studies of harmonic current control, attention is paid to the feasibility of implementation of the proposed control method in addition to further improvement of dynamic and static performance thereof. Generally, an LCL filter is adopted for the high-capacity APF, but a resonance peak exists near the resonance frequency, which results in a reduction of system stability, and in addition, reactive compensation capacitance, harmonic load, grid impedance and coupling between a plurality of APFs affect frequency domain characteristics of the system.
In summary, the existing harmonic current control method of the active power filter has low system stability, requires a complex harmonic detection link during calculation and filtering, and has large calculated amount.
Disclosure of Invention
The invention aims to overcome the defects of the prior art and provides a harmonic current control method of an active power filter.
The aim of the invention can be achieved by the following technical scheme:
a harmonic current control method of active power filter is used for voltage source inverter, the inverter is connected with power grid through active power filter, the active power filter is LCL filter, which comprises filter capacitor, first filter inductance and second filter inductance, the power grid side of filter is connected with second filter inductance, the inverter side is connected with first filter inductance, the power grid is connected with nonlinear load,
the inverter is controlled based on a filter control unit and an SPWM link, wherein the control unit comprises a voltage outer ring module, a midpoint balance strategy module and a current inner ring module comprising an active damping control module;
the control method comprises the following steps:
1) Acquiring the current instantaneous value of the nonlinear load, sending the instantaneous value into a current inner loop module, simultaneously acquiring the voltage of the public coupling point, sending the voltage into a voltage outer loop module,
2) The inner loop control of the current is carried out, and the specific control process in the inner loop module of the current is as follows: sampling the current of the first filter inductor, making a difference between the current instantaneous value of the nonlinear load and the current of the first filter inductor, inputting the difference into an active damping control module, and taking the output of the active damping control module as a modulation wave of harmonic current;
3) The voltage outer ring control is carried out, and the specific control process in the voltage outer ring module is as follows: subtracting the DC capacitor voltage from the DC capacitor voltage set value to obtain an error value, inputting the error value into a first proportional integral controller, and multiplying the output of the first proportional integral controller by the voltage of the public coupling point to serve as a modulation wave of a DC bus voltage stabilizing ring;
4) And performing midpoint balance strategy control, wherein the specific control process of the midpoint balance strategy module is as follows: making a difference between half of the DC capacitor voltage and the capacitor voltage value under the DC bus, and obtaining a modulated wave of the DC bus equalizing ring through a second proportional-integral controller;
5) Adding the modulated wave of the harmonic current, the modulated wave of the DC bus voltage-stabilizing ring and the modulated wave of the DC bus voltage-stabilizing ring to obtain a total modulated wave, inputting the total modulated wave into an SPWM link, and controlling the states of all switching tubes of the inverter.
Further, the active damping control module comprises cascade links of a band-pass filter and a quasi-proportional resonance controller, which are respectively arranged for each subharmonic, the cascade links of each subharmonic are mutually connected in parallel, the product of the feedback parameter and the current of the first filter inductor is calculated, the sum of the outputs of each sublink is differed from the product, and the difference is the output of the active damping control module.
Further, the transfer function of the bandpass filter of the ith harmonic is:
wherein i represents the harmonic order of control, ζ BPFi Represents the damping ratio, ω, of the band-pass filter i Is the center frequency, omega of the band-pass filter i And is the resonant frequency of the quasi-proportional resonant controller.
Further, the transfer function of the quasi-proportional resonant controller of the ith harmonic is:
wherein ,ξi1 With xi i2 The first damping ratio and the second damping ratio, K, of the quasi-proportional resonant controller are respectively Pi Is a parameter of a quasi-proportional resonant controller, omega i Is the center frequency, omega of the band-pass filter i At the same time, the resonant frequency of the quasi-proportional resonant controller, s represents the differential operator.
Further, in order to simplify the controller structure, the first damping ratio of the quasi-proportional resonant controller is set to be equal to the damping ratio of the band-pass filter, so that the transfer function of the cascade link of the band-pass filter of the ith harmonic and the quasi-proportional resonant controller is as follows:
wherein ,transfer function of quasi-proportional resonant controller for ith harmonic, +.>The transfer function, K, of the bandpass filter being the ith harmonic Pi Is the parameter of the quasi-proportional resonance controller, xi BPFi Represents the damping ratio, ω, of the band-pass filter i Is the center frequency, omega of the band-pass filter i At the same time is the resonance frequency of the quasi-proportional resonance controller, ζ i2 Is the second damping ratio of the quasi-proportional resonant controller.
Further, when the active damping control module is digitally implemented, a ZOH method, a FOH method or a bilinear transformation method based on frequency correction is adopted.
Further, a frequency-modified bilinear transform method is used, and the formula used for mapping the s-plane to the z-plane is:
wherein ,ωi Is the center frequency, omega of the band-pass filter i At the same time, the resonant frequency of the quasi-proportional resonant controller, T s Is the sampling period.
Furthermore, each cascade link is provided with a switch, the switch state is determined based on the required compensation frequency, and before each cascade link starts to work, the input and output of the band-pass filter and the quasi-proportional resonance controller as well as intermediate variables are set to zero.
Further, the dc capacitor of the inverter is precharged by diode rectification, and when the dc capacitor voltage reaches a set threshold value, 1) to 5) are started to be executed.
Further, the voltage source type inverter is an I-type or T-type three-level converter or a multi-level converter, adopts IGBT, MOSFET, SIC or GaN power modules, works under three-phase three-wire and three-phase four-wire systems, and is connected with a power grid side N line at the middle point of the voltage at the direct current side when the voltage source type inverter works under the three-phase four-wire system.
Compared with the prior art, the invention has the following beneficial effects:
(1) The invention has simple and reliable structure, solves the defect that the traditional centralized proportional resonance controller can not carry out frequency division control on each subharmonic in a loop and needs to set the required compensation frequency in a harmonic detection link, does not need a special and complex harmonic detection link, directly completes harmonic detection separation and control in a control loop, and has small calculated amount.
(2) The compensation precision is far higher than that of the traditional PI control method, and the proportion and repetition control method.
(3) The method can be realized by simply improving the original equipment, has low improvement cost and does not need to add an additional sensor.
Drawings
FIG. 1 is a flow chart of the present invention;
FIG. 2 is a block diagram of the current loop control of the present invention, wherein FIG. 2 (a) is a detailed block diagram of the filter and FIG. 2 (b) is a detailed block diagram of the active damping control module;
FIG. 3 is a block diagram of the overall system architecture of the present invention, wherein FIG. 3 (a) is a hardware block diagram and FIG. 3 (b) is a software block diagram;
fig. 4 is a diagram showing the output waveform of the harmonic APF and the residual harmonic content at the network side when the harmonic APF is compensated 5,7,11,13 according to the present invention, wherein fig. 4 (a) is an output current waveform of the NPC type inverter, and fig. 4 (b) is an FFT analysis chart of the network side current waveform;
fig. 5 is a dynamic response waveform of the band pass filter bandwidth according to the present invention, wherein fig. 5 (a) is a dynamic response waveform of the APF output current when the BPF damping coefficient is 0.001, and fig. 5 (b) is a dynamic response waveform of the APF output current when the BPF damping coefficient is 0.00001;
FIG. 6 is a bode diagram of the current loop input and output of the present invention;
fig. 7 is a diagram of two current sampling methods that can be used in the present invention, wherein fig. 7 (a) is a diagram of a first current sampling method and fig. 7 (b) is a diagram of a second current sampling method.
Detailed Description
The invention will now be described in detail with reference to the drawings and specific examples. The present embodiment is implemented on the premise of the technical scheme of the present invention, and a detailed implementation manner and a specific operation process are given, but the protection scope of the present invention is not limited to the following examples.
The invention provides a method for controlling harmonic currents of an active power filter (active power filter, APF) based on a band-pass filter (BPF) and a quasi-proportional resonance (QPR) controller. The method is used for a voltage source type inverter (voltage source inverter, VSI), the inverter is connected with a power grid through an active power filter, the active power filter is an LCL filter and comprises a filter capacitor, a first filter inductor and a second filter inductor, the power grid side of the filter is connected with the second filter inductor, the inverter side is connected with the first filter inductor, the power grid is connected with a nonlinear load, the inverter is controlled based on a filter control unit and an SPWM link, and the control unit comprises a voltage outer ring module, a midpoint balance strategy module and a current inner ring module comprising an active damping control module.
According to the invention, a harmonic current control loop carries out frequency division filtering and frequency division control according to harmonic load current detected by a current transformer, and finally, the output of each controller is overlapped to be used as the input of an SPWM modulator to control the switching tube of a voltage source type inverter to work; the current control loop realizes the frequency division control of the harmonic load while carrying out active damping, and a conventional harmonic detection link of instantaneous reactive power, FFT and DFT is not needed; the LCL filter is used to reduce the ripple magnitude of the output current. The invention solves the defects that the traditional centralized proportional resonance controller cannot carry out frequency division control on each subharmonic in a loop and needs to set the required compensation frequency in a harmonic detection link, and the steady-state precision of the method is far higher than that of a common proportional+repeated control parallel mode, and the method can realize the treatment of harmonic current in a power grid without the harmonic detection link.
A flow chart of the method of the present invention is shown in fig. 1. Fig. 3 (a) is a hardware block diagram, and fig. 3 (b) is a software block diagram. The method comprises the following steps:
1) Acquiring the current instantaneous value of the nonlinear load, sending the instantaneous value into a current inner loop module, simultaneously acquiring the voltage of the public coupling point, sending the voltage into a voltage outer loop module,
2) The inner loop control of the current is carried out, and the specific control process in the inner loop module of the current is as follows: sampling the current of the first filter inductor, making a difference between the current instantaneous value of the nonlinear load and the current of the first filter inductor, inputting the difference into an active damping control module, and taking the output of the active damping control module as a modulation wave of harmonic current;
3) The voltage outer ring control is carried out, and the specific control process in the voltage outer ring module is as follows: subtracting the DC capacitor voltage from the DC capacitor voltage set value to obtain an error value, inputting the error value into a first proportional integral controller, and multiplying the output of the first proportional integral controller by the voltage of the public coupling point to serve as a modulation wave of a DC bus voltage stabilizing ring;
4) And performing midpoint balance strategy control, wherein the specific control process of the midpoint balance strategy module is as follows: making a difference between half of the DC capacitor voltage and the capacitor voltage value under the DC bus, and obtaining a modulated wave of the DC bus equalizing ring through a second proportional-integral controller;
5) Adding the modulated wave of the harmonic current, the modulated wave of the DC bus voltage-stabilizing ring and the modulated wave of the DC bus voltage-stabilizing ring to obtain a total modulated wave, inputting the total modulated wave into an SPWM link, and controlling the states of all switching tubes of the inverter.
The active damping control module comprises cascade links of a band-pass filter and a quasi-proportional resonance controller, which are respectively arranged for each subharmonic, wherein the cascade links of each subharmonic are mutually connected in parallel, the product of a feedback parameter and the current of the first filter inductor is calculated, the sum of the outputs of each sublink is differed from the product, and the difference is the output of the active damping control module.
VSI is a main power unit of the system, having good output voltage waveforms and highly reliable voltage regulation performance. Moreover, the VSI has the advantages of low output impedance, high instantaneous load bearing capacity, no great fluctuation of output current caused by load change and the like. In the invention, the VSI can adopt a plurality of different structures, including a six-switch three-phase full-bridge converter, an I-type and T-type three-level converter, a three-phase four-switch converter, a multi-level converter and the like, so as to meet the requirements of different application occasions.
LCL filters are a type of filter commonly used in power electronics systems and are structured with two inductors and a capacitor. One of the inductors is connected in series with the output of the inverter and the other inductor is connected in parallel with the capacitor. Compared with a common L-shaped filter, the LCL filter has better high-frequency harmonic filtering effect and can effectively reduce the ripple wave size of output current. The filter is connected to the output port of VSI, and the inductor current at the inverter side is used as the control target of the harmonic current control loop and as the active damping link to strengthen the stability of the system. The LCL filter design for the APF of the present invention is significantly different from that for a typical grid-tie inverter. The filter cut-off frequency of the APF is 5-10 times that of a common grid-connected inverter, and the APF output current range is far larger than that of the common grid-connected inverter.
The BPF link of the invention has the function of realizing the extraction of the harmonic signal to be compensated by carrying out band-pass filtering on the difference between the harmonic current and the current signal output by the LCL filter. The QPR controller performs high gain control on the output of the BPF link to reduce the steady state error of the system. In addition, the inverter side current proportion feedback also plays a role in changing poles of the LCL filter so as to improve the stability of the whole system, and if the link is not provided, the system cannot work stably. The input signals of the link are the difference between the harmonic current and the inverter side inductor current signal which is independently used as the active damping link, and the output of the inverter side inductor current signal is used as the input signal of the modulator for controlling and modulating the switching tube of the VSI.
Fig. 2 (a) is a detailed block diagram of a filter, and fig. 2 (b) is a detailed block diagram of an active damping control module. The active damping control module comprises a cascading link of total inverter side current active damping and BPF and QPR respectively set for other subharmonics. The inverter side current proportion feedback is used for changing poles of the LCL filter, so that the stability of the system is improved (the system cannot stably work without the link). The BPF and QPR links of each harmonic wave comprise a switch, and before each starting, the input and output of the BPF and QPR links and intermediate variables are set to zero. The BPF link is used for carrying out band-pass filtering on the difference between the harmonic current signal and the output current of the LCL filter, the bandwidth range is the harmonic signal to be compensated, and the QPR controller carries out high-gain control on the BPF output to realize the smallest steady-state error as possible. Transfer function equations (1) and (2) describe specific forms of BPF and QPR, where i represents the harmonic that is controlledTimes, xi BPFi Represents the damping ratio, ω, of the band-pass filter i Is the center frequency of the BPF, ζ i1 With xi i2 Is the damping ratio of the QPR controller. To simplify the controller structure, ζ in equation (1) and equation (2) can be used BPFi ＝ξ i1 The simplified controller structure is shown in the formula (3), so that the calculation amount and the complexity of parameter setting in the controller can be reduced. If digital implementation in a DSP or FPGA is desired, the common forward differential, backward differential and bilinear transformation methods are not well suited in BPFs and QPR of 11 times and above because the current frequency to be compensated is high. For BPF and QPR at high frequencies, possible methods include a ZOH method, a FOH method, and a bilinear transformation method based on frequency correction, wherein the formula of the frequency correction bilinear transformation method is shown in formula (4). The control algorithm of the invention is based on an abc coordinate system instead of a dq coordinate system, and is therefore suitable for use in three-phase three-wire system conditions, as well as in single-phase or three-phase four-wire system conditions with zero sequence current.
Before executing the steps 1) to 5), diode rectification is firstly carried out, and the VSI direct current side current is precharged, at the moment, the harmonic current signal is 0, and the APF system direct current side voltage is ready after reaching a set value. Then, a command harmonic current signal can be added and subjected to high inertia low pass filtering to slow down the speed of current change and prevent the VSI output from flowing excessively. And then, a switch needing to compensate the frequency is opened, and the input and output of the corresponding BPF and the corresponding QPR and the data of the intermediate variable are cleared, so that overcurrent faults are prevented. And then starting to execute 1) to 5), and entering a compensation state and normal operation. If the system needs to be closed, the harmonic current signal is directly set to zero. In addition, the direct-current side capacitor is slowly discharged through a large resistor connected in parallel to consume energy therein.
The control method is generally applied to parallel APF, and when the control method is applied to series APF, the harmonic current command signal is required to be modified into a harmonic voltage signal of the PCC point, and an isolation transformer is connected to the output of the LCL filter to compensate the harmonic voltage of the PCC point.
FIGS. 2 (a) and (b) show a specific form of implementation of harmonic current control of an active power filter based on bandpass filtering and quasi-proportional resonance, comprising an inverter-side current feedback loop K p And each band-pass filtering and proportional resonance link. The design idea of each parameter is as follows:
firstly, adopting matetica software to calculate cut-off frequency f for which gain when QPR controller and LCL filter are separately cascaded c To determine the parameter K of the QPR controller pi 、ξ i1 With xi i2 。
Damping ratio xi of the selected band-pass filter BPFi The closed loop transfer function shown in FIG. 2 is then plotted at different K p A lower pole, selecting a proper K p After parameter calculation is completed, the harmonic wave of each time is calculated in simulation software such as PLECS/Simulink and the like.
Table 1 shows the specific magnitude of the harmonic load current and the parameters of the LCL filter, and the steady-state and dynamic results of the simulation of the controller parameters designed above are shown in fig. 4 and 5. Fig. 4 (a) is an output current waveform of the NPC inverter, and fig. 4 (b) is an FFT analysis chart of a network-side current waveform; FIG. 5 (a) shows the dynamic response waveform of the APF output current when the BPF damping coefficient is 0.001, and FIG. 5 (b) shows the dynamic response waveform of the APF output current when the BPF damping coefficient is 0.00001
TABLE 1 specific magnitude of harmonic load current and parameters of LCL filter
The stability and precision of the system are far higher than those of common proportion and repeated control, and no harmonic detection link is required to be additionally arranged. The dynamic response and steady state response of the system are mainly determined by the BPF and QPR links of each subharmonic, and the current feedback link K of the inverter side p The stability of the system is determined.
In addition to the harmonic current loop parameter design, the closed loop transfer function of the system current, namely the bode diagram of formula (6), is shown in fig. 6, and it can be known that the control algorithm designed by the present patent basically realizes 0dB and 0 phase shift gain for the harmonic frequency band to be controlled, which is also the reason that the steady state accuracy of the present algorithm is higher than that of the general harmonic control algorithm.
Then introducing the hardware implementation method of the method, two common current measurement modes are shown in fig. 7, and two sets of current sensors (current transformer, CT) CT1 and CT2 are needed for the current inner loop to measure the load side current i load Source side current i s And APF output current i F The relationship between these three values satisfies the formula (8). The invention usesIn the manner of load-side current feedback control, if the nonlinear load-side current is not well measured, the nonlinear load-side current is calculated indirectly using equation (8).
i s ＝i F -i load Formula (8)
The CT can be a transformer structure or a Hall sensor structure, the sampled result is scaled to a true value and then is sent to the current inner loop, and the current inner loop and the calculated result of the voltage outer loop and the midpoint balance strategy are acted on the input of the transfer function of the inner loop together.
The foregoing describes in detail preferred embodiments of the present invention. It should be understood that numerous modifications and variations can be made in accordance with the concepts of the invention by one of ordinary skill in the art without undue burden. Therefore, all technical solutions which can be obtained by logic analysis, reasoning or limited experiments based on the prior art by the person skilled in the art according to the inventive concept shall be within the scope of protection defined by the claims.
Claims (10)
1. A harmonic current control method of an active power filter is characterized in that the method is used for a voltage source type inverter, the inverter is connected with a power grid through the active power filter, the active power filter is an LCL filter and comprises a filter capacitor, a first filter inductor and a second filter inductor, the power grid side of the filter is connected with the second filter inductor, the inverter side is connected with the first filter inductor, the power grid is connected with a nonlinear load,
the inverter is controlled based on a filter control unit and an SPWM link, wherein the control unit comprises a voltage outer ring module, a midpoint balance strategy module and a current inner ring module comprising an active damping control module;
the control method comprises the following steps:
1) Acquiring a current instantaneous value of a nonlinear load, sending the instantaneous value into a current inner loop module, simultaneously acquiring a public coupling point voltage, and sending the voltage into a voltage outer loop module;
2) The inner loop control of the current is carried out, and the specific control process in the inner loop module of the current is as follows: sampling the current of the first filter inductor, making a difference between the current instantaneous value of the nonlinear load and the current of the first filter inductor, inputting the difference into an active damping control module, and taking the output of the active damping control module as a modulation wave of harmonic current;
3) The voltage outer ring control is carried out, and the specific control process in the voltage outer ring module is as follows: subtracting the DC capacitor voltage from the DC capacitor voltage set value to obtain an error value, inputting the error value into a first proportional integral controller, and multiplying the output of the first proportional integral controller by the voltage of the public coupling point to serve as a modulation wave of a DC bus voltage stabilizing ring;
4) And performing midpoint balance strategy control, wherein the specific control process of the midpoint balance strategy module is as follows: making a difference between half of the DC capacitor voltage and the capacitor voltage value under the DC bus, and obtaining a modulated wave of the DC bus equalizing ring through a second proportional-integral controller;
5) Adding the modulated wave of the harmonic current, the modulated wave of the DC bus voltage-stabilizing ring and the modulated wave of the DC bus voltage-stabilizing ring to obtain a total modulated wave, inputting the total modulated wave into an SPWM link, and controlling the states of all switching tubes of the inverter.
2. The method for controlling harmonic current of an active power filter according to claim 1, wherein the active damping control module comprises cascade links of a band-pass filter and a quasi-proportional resonance controller, which are respectively provided for each subharmonic, the cascade links of each subharmonic are mutually parallel-connected, a product of a feedback parameter and a current of the first filter inductor is calculated, and a sum of outputs of each sublink is differed from the product, the difference being an output of the active damping control module.
3. The method of claim 2, wherein the transfer function of the bandpass filter of the ith harmonic is:
wherein i represents the harmonic order of control, ζ BPFi Represents the damping ratio, ω, of the band-pass filter i Is the center frequency, omega of the band-pass filter i And is the resonant frequency of the quasi-proportional resonant controller.
4. A method of harmonic current control of an active power filter as claimed in claim 3, wherein the transfer function of the quasi-proportional resonant controller of the ith harmonic is:
wherein ,ξi1 With xi i2 The first damping ratio and the second damping ratio, K, of the quasi-proportional resonant controller are respectively pi Is a parameter of a quasi-proportional resonant controller, omega i Is the center frequency, omega of the band-pass filter i At the same time, the resonant frequency of the quasi-proportional resonant controller, s represents the differential operator.
5. The method of claim 4, wherein, to simplify the controller structure, the first damping ratio of the quasi-proportional resonant controller is set to be equal to the damping ratio of the bandpass filter, so that the transfer function of the cascade connection of the bandpass filter of the ith harmonic and the quasi-proportional resonant controller is:
wherein ,transfer function of quasi-proportional resonant controller for ith harmonic, +.>The transfer function, K, of the bandpass filter being the ith harmonic pi Is the parameter of the quasi-proportional resonance controller, xi BPFi Represents the damping ratio, ω, of the band-pass filter i Is the center frequency, omega of the band-pass filter i At the same time is the resonance frequency of the quasi-proportional resonance controller, ζ i2 Is the second damping ratio of the quasi-proportional resonant controller.
6. The method according to claim 4, wherein the active damping control module is digitally implemented by a ZOH method, a FOH method, or a bilinear transformation method based on frequency correction.
7. The method of claim 6, wherein the frequency-modified bilinear transform is used, and the s-plane mapping to the z-plane uses the formula:
wherein ,ωi Is the center frequency, omega of the band-pass filter i At the same time, the resonant frequency of the quasi-proportional resonant controller, T s Is the sampling period.
8. The method of claim 2, wherein each cascade is provided with a switch, the switching state is determined based on the required compensation frequency, and the input and output of the bandpass filter and the quasi-proportional resonant controller and the intermediate variable are set to zero before each cascade starts to operate.
9. The method according to claim 1, wherein the dc capacitor of the inverter is precharged by diode rectification, and 1) to 5) are started when the dc capacitor voltage reaches a set threshold value.
10. The method for controlling harmonic current of an active power filter according to claim 1, wherein the voltage source inverter is an I-type or T-type three-level converter or a multi-level converter, and adopts IGBT, MOSFET, SIC or GaN power modules, and operates under three-phase three-wire and three-phase four-wire systems, and the voltage midpoint on the dc side is connected with the N-wire on the grid side when operating under the three-phase four-wire system.
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| CN117833248B (en) * | 2024-03-06 | 2024-05-10 | 电子科技大学 | Model-free predictive control method for T-shaped three-level parallel active power filter |
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