CN116470289A - Microstrip reflective array antenna and design method thereof - Google Patents
Microstrip reflective array antenna and design method thereof Download PDFInfo
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- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
- H01Q9/04—Resonant antennas
- H01Q9/0407—Substantially flat resonant element parallel to ground plane, e.g. patch antenna
- H01Q9/045—Substantially flat resonant element parallel to ground plane, e.g. patch antenna with particular feeding means
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- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q1/00—Details of, or arrangements associated with, antennas
- H01Q1/36—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith
- H01Q1/38—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith formed by a conductive layer on an insulating support
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- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q1/00—Details of, or arrangements associated with, antennas
- H01Q1/48—Earthing means; Earth screens; Counterpoises
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- H—ELECTRICITY
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- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q1/00—Details of, or arrangements associated with, antennas
- H01Q1/52—Means for reducing coupling between antennas; Means for reducing coupling between an antenna and another structure
- H01Q1/521—Means for reducing coupling between antennas; Means for reducing coupling between an antenna and another structure reducing the coupling between adjacent antennas
- H01Q1/523—Means for reducing coupling between antennas; Means for reducing coupling between an antenna and another structure reducing the coupling between adjacent antennas between antennas of an array
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- H—ELECTRICITY
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- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q15/00—Devices for reflection, refraction, diffraction or polarisation of waves radiated from an antenna, e.g. quasi-optical devices
- H01Q15/14—Reflecting surfaces; Equivalent structures
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- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q21/00—Antenna arrays or systems
- H01Q21/0006—Particular feeding systems
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- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q21/00—Antenna arrays or systems
- H01Q21/0006—Particular feeding systems
- H01Q21/0075—Stripline fed arrays
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- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01Q—ANTENNAS, i.e. RADIO AERIALS
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- H01Q21/00—Antenna arrays or systems
- H01Q21/24—Combinations of antenna units polarised in different directions for transmitting or receiving circularly and elliptically polarised waves or waves linearly polarised in any direction
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- H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
- H01Q3/26—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture
- H01Q3/30—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array
- H01Q3/32—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array by mechanical means
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- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
- H01Q9/04—Resonant antennas
- H01Q9/0407—Substantially flat resonant element parallel to ground plane, e.g. patch antenna
- H01Q9/0442—Substantially flat resonant element parallel to ground plane, e.g. patch antenna with particular tuning means
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Abstract
The invention discloses a microstrip reflective array antenna and a design method thereof, wherein the microstrip reflective array antenna comprises a feed source and a planar reflective array antenna which is arranged opposite to the feed source through a bracket, and the planar reflective array antenna is a unit structure which is used for realizing 15% of standing wave bandwidth, meeting the working characteristics of dual polarization, stably working at an oblique angle of 45 degrees and having caliber efficiency of not less than 30% and is coupled to a back microstrip line through a slot for phase shifting. According to the microstrip reflective array antenna and the design method thereof, through design based on simulation analysis, the obtained sample has excellent caliber efficiency.
Description
Technical Field
The invention relates to the technical field of antennas, in particular to a microstrip reflective array antenna and a design method thereof.
Background
The microstrip reflective array antenna is formed by two parts of a feed source antenna and a super surface, integrates the advantages of a reflective surface antenna and an array antenna, is widely applied to satellite communication systems and radar systems, and expands the working frequency range from a microwave range to a terahertz frequency range.
In the design of the microstrip reflective array antenna, the research on broadband, oblique angle stability and dual polarization work of a super surface unit, the influence of a feed source on the gain and caliber efficiency of the planar reflective array antenna and the like are lacking.
Disclosure of Invention
In order to solve the problems, the invention provides a microstrip reflective array antenna and a design method thereof, and the obtained sample has excellent caliber efficiency through design based on simulation analysis.
In order to achieve the above purpose, the invention provides a microstrip reflective array antenna, which comprises a feed source and a planar reflective array antenna arranged opposite to the feed source through a bracket, wherein the planar reflective array antenna is a unit structure for realizing 15% of standing wave bandwidth, meeting the working characteristics of dual polarization, and enabling a single-layer patch with stable working under an oblique angle of 45 degrees and caliber efficiency of not less than 30% to be coupled to a back microstrip line through a slot for phase shifting.
Preferably, the unit structure comprises a first dielectric layer, a first radiation patch layer, a second dielectric layer, a first gap coupling layer, a third dielectric layer, a strip line feeder layer, a fourth dielectric layer, a floor layer, a fifth dielectric layer and a microstrip phase shift layer which are sequentially bonded through a PP layer from top to bottom;
The first dielectric layer is used for enabling the feed source to be better matched with the unit structure under the incidence of oblique angles, expanding the bandwidth and protecting the first radiation patch layer;
the first radiation patch layer is used for receiving electromagnetic waves sent by the feed source and radiating the electromagnetic waves outwards after phase shifting;
the second dielectric layer is used for expanding the bandwidth of the gap coupling;
a third dielectric layer for coupling the dual-line polarization to the stripline;
a fourth dielectric layer for isolating the stripline feed line layer from the floor layer;
and the fifth dielectric layer is used for achieving the phase shifting effect by loading microstrip lines with different lengths.
Preferably, the first dielectric layer has a thickness h 1 Tacnoic RF-35A2 of (3) having a dielectric constant of 3.5 and a loss tangent of 0.0018;
a single-layer square patch with a side length of l is attached to the first radiation patch layer with a side length of p;
the second dielectric layer adopts a thickness h 2 F4B-M220 having a dielectric constant of 2.2 and a loss tangent of 0.0007;
two first I-shaped slits which are orthogonally arranged are formed in the first slit coupling layer and are used for coupling the double-line polarization and the strip line feeder layer, the length of the middle section of each first I-shaped slit is g_l, and the length of the rectangles at the two ends of each first I-shaped slit is b_l and the width of each rectangle is b_w;
The thickness of the third dielectric layer is h 3 Tacnoic RF-35A2;
two strip lines which are orthogonally arranged are arranged on the strip line feeder layer, and the length of the strip line is c_l, and the width of the strip line is c_w;
the thickness of the fourth dielectric layer is h 4 F4B-M220 of (F);
the thickness of the fifth dielectric layer is h 5 ；
The microstrip phase shift layer is provided with two microstrip delay lines which are orthogonally arranged, wherein the length of the microstrip delay line is m_l, and the width of the microstrip delay line is m_w;
the PP layer is a Rojess 4450 prepreg with a dielectric constant close to that of the first radiation patch layer and that of the microstrip phase shift layer, and the dielectric constant of the prepreg is 3.52.
Preferably, the first slot coupling layer, the third dielectric layer, the strip line feeder layer, the fourth dielectric layer, the floor layer, the fifth dielectric layer and the microstrip phase shift layer are respectively provided with a first metallization through hole correspondingly for reducing coupling among array elements;
the position of the strip line feeder layer surrounding the first metallized through holes and the position of the microstrip phase shift layer surrounding the first metallized through holes are provided with a plurality of second metallized through holes which are arranged in a matrix and are used for conducting and inhibiting the generation of cavity modes.
Preferably, the switching structure composed of the fourth dielectric layer, the floor layer and the fifth dielectric layer is used for separating the strip line feeder layer and the microstrip phase shift layer;
The switching structure is in the range of 9-18 GHz, the modulus value of the transmission coefficient is not less than-0.5 dB, and the modulus value of the reflection coefficient of the switching structure is less than-15 dB; in the frequency range of 12.5-14.5GHz of the working frequency band, the modulus of the transmission coefficient is larger than-0.3 dB, and the modulus of the reflection coefficient is smaller than-15 dB.
Preferably, the feed source is a microstrip quaternary array antenna with a working frequency band of 12.5-14.5GHz, and the microstrip quaternary array antenna comprises a second radiation patch layer, a sixth dielectric layer, a seventh dielectric layer, a second gap coupling layer, an eighth dielectric layer and a feed layer which are sequentially arranged from top to bottom;
a plurality of p side lengths arranged in a rectangular array are attached on the second radiation patch layer with the side length of a l Is a single layer square patch;
the second I-shaped slits arranged in an array are formed in the second slit coupling layer and are used for increasing the coupling between the second radiation patch layer and the feed layer, so that the bandwidth is increased;
the length of the middle section of the second I-shaped gap is g l G width of w The length of the rectangle at the two ends of the second I-shaped gap is b l Width b w ；
The feed layer adopts a T-shaped power divider, and the T-shaped power divider comprises an input section and an output section which are respectively connected with the second gap coupling layer and the SMA head, and a lambda/4 impedance conversion section connected between the input section and the output section; microstrip impedance of the input section and the output section is 50Ω, and microstrip line width is 1.7mm; the width of the microstrip line of the lambda/4 impedance transformation section is 0.9mm;
The thickness of the sixth dielectric layer is h p F4B-M220 of (F);
the eighth dielectric layer has a thickness h m Tacnoic RF-35A2;
the seventh dielectric layer has a thickness h air An air layer or a PMI foam layer.
Preferably, the feed source is a horn antenna, the horn antenna comprises a rectangular waveguide and a circular gradient which are sequentially connected, an SMA head of 50 omega is connected to the rectangular waveguide, and a metal disc with an open terminal is arranged at the top of a probe of the SMA head of 50 omega;
the overall length of the feedhorns is 137mm;
the length and width of the rectangular waveguide are 22.1mm and 13.2mm respectively;
the maximum caliber of the circular gradual change is 35mm;
the inner and outer radii of the 50Ω SMA head were 1.27mm and 4.1mm, respectively, with polytetrafluoroethylene filled between the SMA head and the metal disc.
A design method of a microstrip reflective array antenna comprises the following steps:
s1, designing a unit structure to achieve 15% of standing wave bandwidth, meet the working characteristics of dual polarization, stably work at an oblique angle of 45 degrees and have caliber efficiency not less than 30%;
s2, designing a feed source, selecting a microstrip quaternary array antenna or a horn antenna as the feed source, wherein the size of the feed source is 120 multiplied by 120mm 2 The focal diameter ratio is 0.8, and the 3dB wave beam width is between 40 and 50 degrees in the working frequency band;
S3, estimating antenna caliber efficiency: according to the unit structure designed in the step S1 and the feed source directional diagram designed in the step S2, preliminary estimation is made on the aperture efficiency of the whole planar reflection array antenna;
s4, designing an array form;
s5, firstly, simulating an array in CST MICROWAVE STUDIO simulation software, analyzing and optimizing results, then processing experimental samples, and testing in a microwave darkroom to verify the feasibility of design.
Preferably, the step S4 specifically includes the following steps:
s41, dispersing phases of 0-360 degrees into 8 phase intervals by utilizing 3-bit phase quantization, and connecting the phase intervals by utilizing 8 microstrip delay lines with different lengths to form reflecting units, wherein each unit represents the phase of one phase interval:
the phase interval corresponding to the 0 degree unit is 315-360 degrees; the phase interval corresponding to the 45 DEG unit is 270-315 DEG; the phase interval corresponding to the 90 DEG unit is 215-270 DEG; the phase interval corresponding to the 135-degree unit is 180-215 degrees; the phase interval corresponding to the 180 DEG unit is 135-180 DEG; the phase interval corresponding to the 215 DEG unit is 90-135 DEG; the phase interval corresponding to the 270-degree unit is 45-90 degrees; the phase interval corresponding to the 315 DEG unit is 0-45 DEG;
s42, designing an array form according to the compensation phase required by the beam deflection angle of the array antenna:
Designing an array of 12×12 array elements, i.e. array plane of 120×120mm 2 The focal diameter ratio is 0.8, the feed source selects a space side feed mode, and the coordinates of the feed source phase center are selected as (20, 60, 84) by taking the lower left corner of the array as the origin of coordinates; then according to the phase compensation formula, calculating the phase distribution of the beam 0-degree emergent array surface, and then designing the form of a planar reflection array according to the phase distribution;
preferably, the step S5 specifically includes the following steps:
s51, respectively carrying out space side feed by a microstrip quaternary array antenna and a horn antenna, and carrying out simulation after the feed source is obliquely incident with the antenna array surface at different angles;
s52, solving by using a CST time domain solver to obtain a simulation result;
s53, processing the horn antenna through a 3D printing technology, and integrally installing and fixing the horn antenna, the plane reflection array and the bracket; the microstrip quaternary array antenna is processed through a PCB process, and is integrally installed and fixed with the planar reflection array and the bracket;
s54, experimental tests are carried out to load patterns of different feed source plane reflection arrays at frequencies of 12.5GHz and 14.5GHz, so that test results are obtained;
s55, comparing the test result with the simulation result, and verifying feasibility.
The invention has the following beneficial effects:
1. The unit structure of broadband, oblique angle stability and dual polarization work is provided, the influence of different types of feeds on the gain and caliber efficiency of the planar reflective array antenna is analyzed and compared, and finally the functionality and rationality of the design are verified through experiments, so that the prepared sample has excellent caliber efficiency;
2. the phase quantization concept is utilized to quantize the phase of 0-360 degrees into 8 phase states, and 8 corresponding reflecting units are correspondingly designed. By utilizing the principle of phase compensation, 8 reflecting units are formed into a plane reflecting array with different beam deflection capacities. The horn feed source and the microstrip quaternary array feed source are processed through a 3D printing technology and a PCB technology, meanwhile, differences brought by different feed sources to the planar reflective array feed are compared, and finally, the working capacities of the planar reflective array in the directions of 0 DEG, 15 DEG, 30 DEG and 45 DEG beams are designed by utilizing the quaternary array feed.
The technical scheme of the invention is further described in detail through the drawings and the embodiments.
Drawings
FIG. 1 is a flow chart of a method for designing a microstrip reflective array antenna according to the present invention;
fig. 2 is a schematic structural diagram of a planar reflective array antenna of the microstrip reflective array antenna according to the present invention;
Fig. 3 is a schematic diagram of a switching structure of a planar reflective array antenna of the microstrip reflective array antenna according to the present invention;
FIG. 4 is a diagram showing the simulation transmission and reflection coefficient results of the switching structure between strip line and microstrip in accordance with the embodiment of the present invention;
FIG. 5 is a graph showing reflectance distribution of a feed port in different modes according to an embodiment of the present invention;
FIG. 6 is a graph showing the mode number of the feed port coupling coefficients in different modes according to an embodiment of the present invention;
FIG. 7 is a graph showing the distribution of the transmission coefficients of the feed ports in different modes according to the embodiment of the present invention;
FIG. 8 is a graph showing the mode value distribution of the polarization coupling coefficient in different modes according to the embodiment of the present invention;
fig. 9 is a schematic diagram of a microstrip quadrifilar array antenna structure of a planar reflective array antenna of a microstrip reflective array antenna according to the present invention;
FIG. 10 is a diagram of reflection coefficient and different frequency points simulated by a microstrip quaternary array antenna according to an embodiment of the present invention;
FIG. 11 is a schematic diagram of a horn antenna of a microstrip reflective array antenna according to the present invention;
FIG. 12 is a graph of simulated reflection coefficients and different frequency points for a horn antenna according to an embodiment of the present invention;
FIG. 13 is a graph of microstrip quad test results according to an embodiment of the present invention;
FIG. 14 is a graph of horn antenna test results according to an embodiment of the present invention;
FIG. 15 is a state diagram of 8 reflective units according to an embodiment of the invention;
FIG. 16 is a graph of S-parameter and phase difference results for different cell states according to an embodiment of the present invention;
FIG. 17 is a plot of the phase profile of a 0℃exit array in accordance with an embodiment of the present invention;
FIG. 18 is an array element layout diagram of an array according to an embodiment of the present invention;
FIG. 19 is a graph of gain versus frequency for different feed plane reflective arrays according to an embodiment of the invention;
FIG. 20 is a 15, 30, and 45 exit array face phase distribution and array layout diagram of an embodiment of the present invention;
FIG. 21 is a graph of different frequencies and different angular deflections of a planar reflective array according to an embodiment of the invention;
FIG. 22 is a schematic diagram of sample processing according to an embodiment of the present invention;
FIG. 23 is a schematic diagram of a quaternary feed source rack construction according to an embodiment of the present invention;
FIG. 24 is a schematic view of a horn feed source mount configuration according to an embodiment of the present invention;
FIG. 25 is a diagram comparing experimental and simulated patterns of different feeds at different frequencies in accordance with an embodiment of the present invention.
Detailed Description
The present invention will be further described with reference to the accompanying drawings, and it should be noted that, while the present embodiment provides a detailed implementation and a specific operation process on the premise of the present technical solution, the protection scope of the present invention is not limited to the present embodiment.
For broadband planar reflective arrays in linear polarization, common forms are mainly square single and double ring structures, square patches and multi-layer patch structures. The embodiment provides a unit structure that a single-layer patch is coupled to a back microstrip line through a slot to carry out phase shifting, the unit structure is simple, and a modulation phase and a receiving patch are positioned on different layers, so that the possibility of coupling is reduced, and most importantly, the reflection phase within 360 degrees can be easily achieved according to the length of a microstrip delay line which is designed.
The microstrip reflection array antenna comprises a feed source and a planar reflection array antenna which is arranged opposite to the feed source through a bracket, wherein the planar reflection antenna is a unit structure which is used for realizing 15% of standing wave bandwidth, meeting the working characteristics of dual polarization, and enabling a single-layer patch with caliber efficiency of not less than 30% to stably work at an oblique angle of 45 degrees to be coupled to a back microstrip line through a gap for phase shifting.
Specifically, the unit structure comprises a first dielectric layer, a first radiation patch layer, a second dielectric layer, a first gap coupling layer, a third dielectric layer, a strip line feeder layer, a fourth dielectric layer, a floor layer, a fifth dielectric layer and a microstrip phase shift layer which are sequentially bonded from top to bottom through a PP layer; the first dielectric layer is used for enabling the feed source to be better matched with the unit structure under the incidence of oblique angles, expanding the bandwidth and protecting the first radiation patch layer; the first radiation patch layer is used for receiving electromagnetic waves sent by the feed source and radiating the electromagnetic waves outwards after phase shifting; the second dielectric layer is used for expanding the bandwidth of the gap coupling; a third dielectric layer for coupling the dual-line polarization to the stripline; a fourth dielectric layer for isolating the stripline feed line layer from the floor layer; and the fifth dielectric layer is used for achieving the phase shifting effect by loading microstrip lines with different lengths.
Preferably, the first mediumThe thickness of the mass layer is h 1 Tacnoic RF-35A2 of (3) having a dielectric constant of 3.5 and a loss tangent of 0.0018; a single-layer square patch with a side length of l is attached to the first radiation patch layer with a side length of p; the second dielectric layer adopts a thickness h 2 F4B-M220 having a dielectric constant of 2.2 and a loss tangent of 0.0007; two first I-shaped slits which are orthogonally arranged are formed in the first slit coupling layer and are used for coupling the double-line polarization and the strip line feeder layer, the length of the middle section of each first I-shaped slit is g_l, and the length of the rectangles at the two ends of each first I-shaped slit is b_l and the width of each rectangle is b_w; the thickness of the third dielectric layer is h 3 Tacnoic RF-35A2; two strip lines which are orthogonally arranged are arranged on the strip line feeder layer, and the length of the strip line is c_l, and the width of the strip line is c_w; the thickness of the fourth dielectric layer is h 4 F4B-M220 of (F); the thickness of the fifth dielectric layer is h 5 The method comprises the steps of carrying out a first treatment on the surface of the The microstrip phase shift layer is provided with two microstrip delay lines which are orthogonally arranged, wherein the length of the microstrip delay line is m_l, and the width of the microstrip delay line is m_w; the PP layer is a Rojess 4450 prepreg with a dielectric constant close to that of the first radiation patch layer and that of the microstrip phase shift layer, and the dielectric constant of the prepreg is 3.52.
In this embodiment, a super-surface unit for slot radiation coupling feed operating in Ku band is designed based on multilayer PCB technology, and the unit will be used for broadband planar array antennas and planar folded array antennas.
Table 1 is a structural dimension table unit of Ku band array unit: mm (mm)
Preferably, the first slot coupling layer, the third dielectric layer, the strip line feeder layer, the fourth dielectric layer, the floor layer, the fifth dielectric layer and the microstrip phase shift layer are respectively provided with a first metallization through hole correspondingly for reducing coupling among array elements;
the position of the strip line feeder layer surrounding the first metallized through holes and the position of the microstrip phase shift layer surrounding the first metallized through holes are provided with a plurality of second metallized through holes which are arranged in a matrix and are used for conducting and inhibiting the generation of cavity modes. Because four isolated second metallization through holes penetrate through the whole layer, the PCB is convenient to press in the process of pressing, the processing difficulty is reduced, but simultaneously, the lengths of the plurality of through holes are equivalent to those of parallel connection with an inductive load, the performance of the switching structure is directly influenced, and the influence can be reduced through simulation optimization. Meanwhile, in order to simulate the processing test more closely, bonding pads are added at corresponding positions of the through holes for simulation.
Preferably, the switching structure composed of the fourth dielectric layer, the floor layer and the fifth dielectric layer is used for separating the strip line feeder layer and the microstrip phase shift layer;
the switching structure is in the range of 9-18 GHz, the modulus value of the transmission coefficient is not less than-0.5 dB, and the modulus value of the reflection coefficient of the switching structure is less than-15 dB; in the frequency range of 12.5-14.5GHz of the working frequency band, the modulus of the transmission coefficient is larger than-0.3 dB, and the modulus of the reflection coefficient is smaller than-15 dB.
The switching structure can be used for ensuring the effective transmission of electromagnetic wave energy between the strip line and the microstrip phase shift. Therefore, performance simulation of the switching structure is necessary before designing the unit structure.
Preferably, as the resonance of the microstrip antenna is similar to a resonance circuit with a high Q value, the bandwidth is narrow, and the bandwidth can only reach about 5-10% of the relative bandwidth, and the bandwidth is difficult to reach by using a common microstrip antenna for the working frequency band of 12.5-14.5 GHz; therefore, the feed source in the embodiment is a microstrip quaternary array antenna with the working frequency band of 12.5-14.5GHz, and the microstrip quaternary array antenna comprises a second radiation patch layer, a sixth dielectric layer, a seventh dielectric layer, a second gap coupling layer, an eighth dielectric layer and a feed layer which are sequentially arranged from top to bottom; a plurality of p side lengths arranged in a rectangular array are attached on the second radiation patch layer with the side length of a l Is a single layer square patch; the second H-shaped slits arranged in an array are arranged on the second slit coupling layer for enlargingCoupling between the second radiating patch layer and the feed layer, thereby increasing bandwidth; the length of the middle section of the second I-shaped gap is g l G width of w The length of the rectangle at the two ends of the second I-shaped gap is b l Width b w The method comprises the steps of carrying out a first treatment on the surface of the The feed layer adopts a T-shaped power divider, and the T-shaped power divider comprises an input section and an output section which are respectively connected with the second gap coupling layer and the SMA head, and a lambda/4 impedance conversion section connected between the input section and the output section; microstrip impedance of the input section and the output section is 50Ω, and microstrip line width is 1.7mm; the width of the microstrip line of the lambda/4 impedance transformation section is 0.9mm; the thickness of the sixth dielectric layer is h p F4B-M220 of (F); the eighth dielectric layer has a thickness h m Tacnoic RF-35A2; the seventh dielectric layer has a thickness h air An air layer or a PMI foam layer. In this embodiment, 7 through holes with the same size are uniformly and correspondingly formed in each layer, the diameter of each through hole is 4mm, and the two layers of dielectric plates and the PMI foam layer are fixed by nylon screws.
Table 2 is a structural parameter table unit of the microstrip quaternary array antenna: mm (mm)
| a | p l | g l |
| 30 | 6.5 | 3.2 |
| g w | b l | b w |
| 0.9 | 1.8 | 0.7 |
| h p | h m | h air |
| 0.5 | 0.8 | 2 |
The simulation is carried out on the designed microstrip quaternary array, and in order to enable the simulation result to be more approximate to the real sample test result, a 50 omega SMA head is designed at a microstrip feed port during the simulation. As shown in FIG. 10 (a), the reflection coefficient of the quaternary array and the standing wave thereof are smaller than-15 dB at the port in the range of the working frequency band, the size of the standing wave is within 1.3, and the surface electromagnetic wave can be effectively radiated in the working frequency band. The simulation result which needs to be concerned is the shape of the pattern in the working frequency band and the 3dB wave beam width, and 5 working frequency point analyses of 12.5GHz, 13GHz, 13.5GHz, 14GHz and 14.5GHz are selected. As shown in fig. 10 (b, c, d, E, f), first, the 3dB beam widths of the E plane and the H plane at the 5 frequency points are basically between 45 ° and 55 °, the E plane and the H plane have better symmetry and consistent radiation directions, and the sidelobe levels are all less than-20 dB.
Preferably, the feed source is a horn antenna, the horn antenna comprises a rectangular waveguide and a circular gradient which are sequentially connected, an SMA head of 50 omega is connected to the rectangular waveguide, and a metal disc with an open terminal is arranged at the top of a probe of the SMA head of 50 omega; the overall length of the feedhorns is 137mm; the length and width of the rectangular waveguide are 22.1mm and 13.2mm respectively; the maximum caliber of the circular gradual change is 35mm; the inner and outer radii of the 50Ω SMA head were 1.27mm and 4.1mm, respectively, with polytetrafluoroethylene filled between the SMA head and the metal disc.
As shown in FIG. 12, the simulation result of the horn antenna shows that the reflection coefficient is smaller than-15 dB in the working frequency band, the gain of the horn antenna at the lowest frequency of 12.5GHz is 12.1dBi, the 3dB beam width of the E plane is 47 degrees, and the 3dB beam width of the H plane is 47 degrees. At the highest frequency of 14.5GHz, the gain is 13.5dBi, the 3dB wave beam widths of the E face and the H face are 38 degrees and 40 degrees respectively, and the E face and the H face directional diagrams under different frequency points show perfect symmetry. The reasonable beam width can enable the planar reflective array to obtain higher caliber efficiency, and the horn antenna of the embodiment considers the size of the 3dB beam width in design, so that the gains of the horn at different frequency points are not required to be too large, and the horn antenna can meet the requirement that the horn has better radiation characteristics and directivity within the 3dB beam width. Therefore, the horn antenna meets the working requirement of a planar reflective array feed source at 12.5-14.5 GHz.
Compared with the two feeds, from simulation results, the horn antenna has stable standing waves and perfect symmetry of E-plane and H-plane directional patterns, the horn antenna has no stronger back lobes, the energy radiation direction is concentrated, and the microstrip quaternary array has the advantages of lower profile and capability of greatly reducing the overall height of the planar reflective array antenna.
The embodiment further compares experimental test results of two different feeds:
simultaneously, respectively processing a microstrip quaternary array and a horn antenna.
As shown in FIG. 13, the reflection coefficient S11 measured by the experiment is greatly different from that of the simulation, although three resonance points are reserved, the resonance frequency of the resonance points is shifted to high frequency and the resonance interval is increased, and the reason for the phenomenon is that the PMI foam with the dielectric constant of 1.15 is selected for an air layer to replace a nylon rivet with a larger dielectric constant to bring about a certain effect, but the S11 is below-10 dB in the working frequency range. For the directional diagram, the 3dB wave beam width is between 40 and 50 degrees, and the requirements of the planar reflective array are met.
The processing of horn antenna adopts 3D printing technique, and the part of printing is horn antenna's main part, and the metal material of printing is aluminum alloy, directly adopts the mode of purchasing to feed port SMA joint, later stage with print the main part and weld with the SMA joint. The processed samples are shown in fig. 14, and the reflectance and E-plane pattern were also tested. The reflection coefficient curves of the experiment and the simulation can be seen to be basically consistent, but the reflection coefficient curves are slightly different at 13.5GHz, because the 3D printed metal surface has certain unevenness, the resonance points are inconsistent, but the reflection coefficient of the loudspeaker is smaller than-15 dB at the working frequency band of the planar reflection array, and the loudspeaker has better radiation characteristics. The directional diagrams of the antenna at 12.5GHz and 14.5GHz are measured in the microwave darkroom, and the 3dB wave beam width is between 40 and 50 degrees in the working frequency band, so that the requirements of a planar reflective array feed source are met.
A design method of a microstrip reflective array antenna comprises the following steps:
s1, designing a unit structure to achieve 15% of standing wave bandwidth, meet the working characteristics of dual polarization, stably work at an oblique angle of 45 degrees and have caliber efficiency not less than 30%;
s2, designing a feed source, selecting a microstrip quaternary array antenna or a horn antenna as the feed source, wherein the size of the feed source is 120 multiplied by 120mm 2 The focal diameter ratio is 0.8, and the 3dB wave beam width is between 40 and 50 degrees in the working frequency band;
s3, estimating antenna caliber efficiency: according to the unit structure designed in the step S1 and the feed source directional diagram designed in the step S2, preliminary estimation is made on the aperture efficiency of the whole planar reflection array antenna;
the process of calculating the aperture efficiency in this embodiment takes into consideration a plurality of factors of the antenna, including the radiation pattern of the antenna, the aperture of the antenna, the beam width and the gain (1) firstly, we need to calculate the theoretical gain G of the antenna t ：G t ＝4πA e λ, wherein A e Is the effective aperture area of the antenna, lambda is the wavelength of the signal; (2) And obtaining the actual gain G of the antenna, wherein the obtaining method comprises the following steps: actual measurements or from the antenna specifications provided by the antenna manufacturer; (3) calculating caliber efficiency:
In this embodiment, if the directional diagram of the feed source is narrower, the gain of the array element at the caliber edge is much smaller than that received by the array element at the caliber center, so that the caliber efficiency is reduced, and the caliber efficiency can be improved by increasing the proper focal diameter ratio. For the array element pattern, if the pattern is too narrow, the gain received by the edge array element is smaller than that of the center array element, so the pattern of the array element is generally designed into a wider form.
S4, designing an array form;
preferably, the step S4 specifically includes the following steps:
s41, larger arrays will bring about higher gain and stronger directivity, smaller arrays will have significant cross-polarization and larger side lobes. Thus, the proper array size is designed according to the requirements; therefore, in this embodiment, the 3-bit phase quantization is used to disperse the phase of 0-360 ° into 8 phase intervals, and the reflection units are formed by connecting 8 microstrip delay lines with different lengths, each unit represents the phase of a phase interval:
as shown in fig. 15, the phase interval corresponding to the 0 ° unit is 315-360 °; the phase interval corresponding to the 45 DEG unit is 270-315 DEG; the phase interval corresponding to the 90 DEG unit is 215-270 DEG; the phase interval corresponding to the 135-degree unit is 180-215 degrees; the phase interval corresponding to the 180 DEG unit is 135-180 DEG; the phase interval corresponding to the 215 DEG unit is 90-135 DEG; the phase interval corresponding to the 270-degree unit is 45-90 degrees; the phase interval corresponding to the 315 DEG unit is 0-45 DEG;
Table 3 is a table of 8 reflective element size units: mm (mm)
| Cell state | l 1 | l 2 | l 3 | l 4 | l 5 | l 6 | l 7 |
| 0° | 1.7 | 1.1 | 4.9 | 2 | 2.3 | 6.4 | 3.8 |
| 45° | 1.5 | 1.5 | 4.4 | 2 | 2.1 | 6.3 | 3.6 |
| 90° | 6.5 | 4.4 | 3.2 | \ | \ | \ | \ |
| 135° | 5.8 | 4.4 | 2.5 | \ | \ | \ | \ |
| 180° | 1.7 | 2.6 | 6.8 | \ | \ | \ | \ |
| 225° | 3 | 1 | 1.3 | 4.9 | 1.5 | \ | \ |
| 270° | 3 | 4.8 | \ | \ | \ | \ | \ |
| 315° | 1.7 | 6 | 6.4 | 3 | \ | \ | \ |
All unit simulations are based on the periodic boundary conditions of the CST simulation software, and since the planar reflective array is formed by arranging 8 types of reflective units, the coupling of other array elements must be considered in the design of the array elements. According to Floquet theory, the periodic boundary condition can simulate an infinite array environment, the coupling of other array elements can be considered in the simulation of the units, and the S parameter of each unit and the phase difference between the units can be obtained through the simulation of the designed excitation ports and the boundary condition. It should be noted here that the time-harmonic factor of CST simulation software is e -iωt The 0 ° cell is therefore instead a microstrip delay line with a longer loading length. The Floquet mode was selected as excitation, and simulation was performed on 8 units, respectively.
As shown in FIG. 16, the cross polarization conversion rate of 8 states is within-0.7 dB in the working frequency band, and the co-polarization is less than-10 dB, so that the cross polarization isolation performance is good. The phase difference is designed by taking the reflection phase of the 315 DEG reflecting unit as a reference and taking the highest frequency 14.5GHz of the bandwidth, and the phase difference of 8 reflecting units at 14.5GHz is 0 DEG, 45 DEG, 90 DEG, 135 DEG, 180 DEG, 225 DEG, 270 DEG and 315 DEG respectively, so that the design requirement of the array is met. It can be seen that the resonance points of the reflection units of 0 ° and 45 ° are more than those of other reflection units and the insertion loss is larger, because the microstrip phase shift of the reflection units of 0 ° and 45 ° is more complex than that of other reflection units, and the microstrip lines inevitably cause coupling between lines and certain reflection of microstrip corners when the microstrip lines are routed due to the limitation of the back space of the units, and multiple reflections form fabry-perot-like resonance, so that more resonance points are brought, and the insertion loss is reduced to the minimum by optimizing the routing mode of the microstrip lines.
S42, designing an array form according to the compensation phase required by the beam deflection angle of the array antenna:
as shown in FIG. 17, an array of 12×12 array elements, i.e., array plane of 120×120mm in size, is designed 2 The focal diameter ratio is 0.8, the feed source selects a space side feed mode, and the coordinates of the feed source phase center are selected as (20, 60, 84) by taking the lower left corner of the array as the origin of coordinates; the reflection phase of the planar reflective array element is then compensated for the spatial phase delay phi from the feed phase center to the element according to the formula for phase compensation pd ＝-k 0 R i Wherein R is i K is the distance from the feed phase center to the ith cell 0 For wave number in vacuum), the phase distribution of the beam 0-degree exit array surface is calculated, and then the form of the planar reflection array is designed according to the phase distribution (one key step of designing the planar reflection array antenna is to select a proper phase tuning method (loading a phase delay line), so that the planar reflection array element of the planar reflection array can realize the required phase tuning range. Once the phase tuning method is selected, the unit characteristics can be determined, and finally, the radiation characteristics of the antenna are realized by using the unit characteristic arrangement);
s5, firstly, simulating an array in CST MICROWAVE STUDIO simulation software, analyzing and optimizing results, then processing experimental samples, and testing in a microwave darkroom to verify the feasibility of design.
In this embodiment, simulation is performed in CST MICROWAVE STUDIO simulation software, and periodic boundary conditions are used for the unit structure to simulate an infinite periodic structure. For a dual-polarized unit, attention is paid to the reflection coefficient of a port under different incident modes, and the reflection coefficient of the port reflects the quality of matching between the unit and a feed source. As shown in FIG. 5, under normal incidence of TE and TM modes, the reflection coefficient of the port is less than-15 dB in the range of 12-15 GHz. With the increase of the incident angle, the reflection coefficients of TE and TM mode ports are also increased gradually, when the incident angle is increased by 45 degrees, the reflection coefficient of the ports is increased obviously, but is smaller than-10 dB in the range of 12-15 GHz, and the requirements for units working at 12.5-14.5 GHz are met.
Second, the coupling coefficient between the different modes of the feed ports is one of the indicators of concern, and its low coupling coefficient ensures that the unit does not receive electromagnetic energy of the reverse polarization at single polarization incidence. As shown in fig. 6, the coupling coefficient of the ports is less than-25 dB in the range of 12-15 GHz at normal incidence of TE mode. For the TM mode, as the incident angle increases, the coupling coefficient of the TM mode port increases gradually to-22 dB, and the requirements of units working at 12.5-14.5 GHz can be met.
The most focused unit design is the transmission coefficient from the feed source to the unit, and the value of the transmission coefficient reflects the intensity of the energy of the feed source received by the unit. Under TE and TM feeding modes, the energy of the feed source is received through the patch and then coupled to the strip line through the slot and finally transmitted to the back microstrip phase shifting layer through the through hole, a matching Port is attached to the terminal of the microstrip line, as shown in FIG. 7, the received energy is better than-1 dB in the range of 12-15 GHz, and the stable transmission coefficient is maintained when the incident angle is increased to 45 degrees. For the working frequency band of the array of 12.5-14.5 GHz, the transmission coefficient is smaller than-0.7 dB, and the unit meets the requirement of array design.
Finally, under different mode incidence, the coupling coefficient between the microstrip line receiving polarizations is as shown in fig. 8, and the simulation result shows that under normal incidence, the polarization coupling coefficient at high frequency is larger, but smaller than-25 dB in the operating frequency band of the unit. With the increase of the incident angle, the polarization coupling coefficient of the receiving port is gradually reduced under the incident of two modes, and the good polarization isolation performance is shown.
Preferably, the step S5 specifically includes the following steps:
s51, respectively carrying out space side feed by a microstrip quaternary array antenna and a horn antenna, and carrying out simulation after the feed source is obliquely incident with the antenna array surface at different angles;
After the array arrangement is completed, space side feeding is respectively carried out by the horn antenna and the micro-strip quaternary array, and incidence is carried out by the included angle between the feed source and the antenna array surface being 25 degrees. In simulation, far-field patterns of 12.5GHz, 13.5GHz and 14.5GHz are observed respectively, and as shown in a far-field pattern pair such as that shown in FIG. 19, the planar reflective array has strong beam concentration in the 0-degree direction, and the gains of the horn and the quaternary array feed source are 16.7 and 18.1dBi respectively at a frequency point of 12.5 GHz; under the 13.5GHz frequency point, the gains of the horn and the quaternary array feed source are 19.6 dBi and 20.6dBi respectively; at the frequency point of 14.5GHz, the gains of the horn and the quaternary array feed source are 22.2 dBi and 22.0dBi respectively. The gain of the horn serving as the feed source planar reflective array is obviously smaller than that of the quaternary array serving as the feed source planar reflective array at low frequency, and the gains of the two feed sources are approximately the same at high frequency. But the side lobe of the horn feed source planar reflection array can be seen to be superior to that of the quaternary array serving as the feed source. The caliber efficiency of the antenna under the three different frequency points is calculated by utilizing a formula of the caliber efficiency of the antenna, and the caliber efficiencies of the horn serving as a feed source at 12.5GHz, 13.5GHz and 14.5GHz are respectively 14.88%, 24.89% and 39.26%. Caliber efficiencies of the quaternary array serving as a feed source at 12.5GHz, 13.5GHz and 14.5GHz are 20.55%, 31.33% and 37.49%, respectively. The reason for the low-frequency aperture efficiency is that the phase delay is designed according to the highest frequency of 14.5GHz when the reflection unit is designed, and dispersion correction of the whole frequency band is not achieved, so that the low-frequency gain is smaller than the theoretical gain, and the low-frequency aperture efficiency is lower.
As shown in fig. 20, in order to verify the working condition of the oblique angle of the plane reflection array, the plane reflection arrays of 15 °, 30 ° and 45 ° are designed, and the forward emergent quaternary array is superior to the horn feed source, so that the plane reflection arrays obliquely emergent are simulated by using the quaternary array as the feed source.
S52, solving by using a CST time domain solver to obtain a simulation result;
as shown in fig. 21, the array is in the XOY plane, and the pattern view is all of the plane phi=0, i.e., the E-plane pattern. As a result, the beam is emitted in the designed direction, and the gain of the planar reflective array is reduced as the deflection angle of the beam is increased. At the lowest frequency of 12.5GHz of the working frequency band, the maximum gain of the oblique emergent beam is 17.2dBi, the minimum gain is 15.5dBi, and at the highest frequency of 14.5GHz of the working frequency band, the maximum gain of the oblique emergent beam is 22.6dBi, and the minimum gain is 21.3dBi.
S53, processing the horn antenna through a 3D printing technology, and integrally installing and fixing the horn antenna, the plane reflection array and the bracket; the microstrip quaternary array antenna is processed through a PCB process, and is integrally installed and fixed with the planar reflection array and the bracket;
the plane reflection array real object test of the Ku wave band selects the outgoing sample for processing. The fixed joint of the planar reflective array antenna and the bracket of the space feed source must be designed during the transmission and processing. As shown by the design of array elements, the whole array is composed of 5 layers of PCB boards, prepregs are needed to be used for laminating and fixing each layer of boards during processing, rogowski 4450 is selected as the prepreg, the dielectric constant of the prepreg is 3.52, the laminating thickness is 0.1mm, and the phenomenon of air bubbles can occur when the prepreg with the thickness of 0.1mm is laminated due to the fact that the thickness of the dielectric board of the phase-shifting layer is 0.127mm, so that the prepreg with the thickness of 0.2mm is laminated with the phase-shifting layer during sample processing, and the occurrence of air bubbles is prevented. In addition, during simulation, the prepreg needs to be added into the simulation design together so as to achieve the consistency of simulation and experiment. The schematic diagram of the designed sample is shown in fig. 22, when the design is carried out, the dielectric plate and the grounding plate are outwards extended by 2.7mm in the upper, lower and right directions of the planar reflection array, the left direction of the planar reflection array is outwards extended by 8.7mm, and meanwhile, five through holes with the diameter of 4mm and the interval of 15mm are formed in the center of the left extension, so that the planar reflection array and the bracket are fixed. The processing of the planar reflection array antenna bracket adopts a 3D printing technology, the used material is white resin, the dielectric constant of the used resin is low, the strength is enough to support the weight of the planar reflection array and the feed source, the emergent influence on the beam of the planar reflection array is small, and strong reflection and absorption can not be caused.
In order to verify the difference of two feeds to a planar reflecting array, a microstrip quaternary array and a feed source bracket of a loudspeaker are designed at the same time, the processed planar reflecting array and the feed source are fixedly assembled by nylon rivets, the structural form is shown in fig. 23 and 24, meanwhile, the directional diagram test is carried out on a university of Shanghai microwave darkroom, and the whole size of the darkroom is 10 multiplied by 6 multiplied by 4mm 3 The environment of the pattern test is satisfied. The size of the beam gain of the planar reflective array is received by using a horn antenna with standard gain, and the loading of the planar reflective array with different feeds is tested at 12 respectively.Pattern at 5GHz and 14.5GHz frequencies.
S54, experimental tests are carried out to load patterns of different feed source plane reflection arrays at frequencies of 12.5GHz and 14.5GHz, so that test results are obtained;
s55, comparing the test result with the simulation result, and verifying feasibility.
The test result is shown in fig. 25, and the test result is compared with the simulation result, and at 12.5GHz and 14.5GHz, the deflection direction of the wave beam is deflected to the designed direction, the maximum gain of the main lobe is 21.2dBi, the side lobe level is 0dBi, and meanwhile, the gain of the main lobe measured by the experiment is slightly lower than the simulation result, which is caused by a certain error in the experiment and a certain absorption of the bracket. The caliber efficiency is calculated to be 16.3% and 38.5% at 12.5GHz and 14.5GHz respectively.
Therefore, the microstrip reflective array antenna and the design method thereof are adopted, and the obtained sample has excellent caliber efficiency through the design based on simulation analysis.
Finally, it should be noted that: the above embodiments are only for illustrating the technical solution of the present invention and not for limiting it, and although the present invention has been described in detail with reference to the preferred embodiments, it will be understood by those skilled in the art that: the technical scheme of the invention can be modified or replaced by the same, and the modified technical scheme cannot deviate from the spirit and scope of the technical scheme of the invention.
Claims (10)
1. The utility model provides a microstrip reflection array antenna, includes feed and through the relative plane reflection array antenna who sets up of support and feed, its characterized in that: the planar reflection antenna is a unit structure for realizing 15% of standing wave bandwidth, meeting the working characteristics of dual polarization, and enabling a single-layer patch with caliber efficiency of not less than 30% to stably work at an oblique angle of 45 degrees to be coupled to a back microstrip line through a slot for phase shifting.
2. A microstrip reflective array antenna according to claim 1, wherein: the unit structure comprises a first dielectric layer, a first radiation patch layer, a second dielectric layer, a first gap coupling layer, a third dielectric layer, a strip line feeder layer, a fourth dielectric layer, a floor layer, a fifth dielectric layer and a microstrip phase shift layer which are sequentially bonded from top to bottom through a PP layer;
The first dielectric layer is used for enabling the feed source to be better matched with the unit structure under the incidence of oblique angles, expanding the bandwidth and protecting the first radiation patch layer;
the first radiation patch layer is used for receiving electromagnetic waves sent by the feed source and radiating the electromagnetic waves outwards after phase shifting;
the second dielectric layer is used for expanding the bandwidth of the gap coupling;
a third dielectric layer for coupling the dual-line polarization to the stripline;
a fourth dielectric layer for isolating the stripline feed line layer from the floor layer;
and the fifth dielectric layer is used for achieving the phase shifting effect by loading microstrip lines with different lengths.
3. A microstrip reflective array antenna according to claim 2, wherein: the thickness of the first dielectric layer is h 1 TacnoicRF-35A2 of (3) having a dielectric constant of 3.5 and a loss tangent of 0.0018;
a single-layer square patch with a side length of l is attached to the first radiation patch layer with a side length of p;
the second dielectric layer adopts a thickness h 2 F4B-M220 having a dielectric constant of 2.2 and a loss tangent of 0.0007;
two first I-shaped slits which are orthogonally arranged are formed in the first slit coupling layer and are used for coupling the double-line polarization and the strip line feeder layer, the length of the middle section of each first I-shaped slit is g_l, and the length of the rectangles at the two ends of each first I-shaped slit is b_l and the width of each rectangle is b_w;
The thickness of the third dielectric layer is h 3 TacnoicRF-35A2;
two strip lines which are orthogonally arranged are arranged on the strip line feeder layer, and the length of the strip line is c_l, and the width of the strip line is c_w;
the thickness of the fourth dielectric layer is h 4 F4B-M220 of (F);
the thickness of the fifth dielectric layer is h 5 ；
The microstrip phase shift layer is provided with two microstrip delay lines which are orthogonally arranged, wherein the length of the microstrip delay line is m_l, and the width of the microstrip delay line is m_w;
the PP layer is a Rojess 4450 prepreg with a dielectric constant close to that of the first radiation patch layer and that of the microstrip phase shift layer, and the dielectric constant of the prepreg is 3.52.
4. A microstrip reflective array antenna according to claim 3, wherein: the first slot coupling layer, the third dielectric layer, the strip line feeder layer, the fourth dielectric layer, the floor layer, the fifth dielectric layer and the microstrip phase shift layer are respectively provided with a first metallization through hole correspondingly for reducing coupling among array elements;
the position of the strip line feeder layer surrounding the first metallized through holes and the position of the microstrip phase shift layer surrounding the first metallized through holes are provided with a plurality of second metallized through holes which are arranged in a matrix and are used for conducting and inhibiting the generation of cavity modes.
5. A microstrip reflective array antenna according to claim 2, wherein: the switching structure consisting of the fourth dielectric layer, the floor layer and the fifth dielectric layer is used for separating the strip line feeder layer and the microstrip phase shift layer;
The switching structure is in the range of 9-18 GHz, the modulus value of the transmission coefficient is not less than-0.5 dB, and the modulus value of the reflection coefficient of the switching structure is less than-15 dB; in the frequency range of 12.5-14.5GHz of the working frequency band, the modulus of the transmission coefficient is larger than-0.3 dB, and the modulus of the reflection coefficient is smaller than-15 dB.
6. A microstrip reflective array antenna according to claim 1, wherein: the feed source is a microstrip quaternary array antenna with a working frequency band of 12.5-14.5GHz, and the microstrip quaternary array antenna comprises a second radiation patch layer, a sixth dielectric layer, a seventh dielectric layer, a second gap coupling layer, an eighth dielectric layer and a feed layer which are sequentially arranged from top to bottom;
a plurality of p side lengths arranged in a rectangular array are attached on the second radiation patch layer with the side length of a l Is a single layer square patch;
the second I-shaped slits arranged in an array are formed in the second slit coupling layer and are used for increasing the coupling between the second radiation patch layer and the feed layer, so that the bandwidth is increased;
the length of the middle section of the second I-shaped gap is g l G width of w The length of the rectangle at the two ends of the second I-shaped gap is b l Width b w ；
The feed layer adopts a T-shaped power divider, and the T-shaped power divider comprises an input section and an output section which are respectively connected with the second gap coupling layer and the SMA head, and a lambda 4 impedance transformation section connected between the input section and the output section; microstrip impedance of the input section and the output section is 50Ω, and microstrip line width is 1.7mm; the width of the microstrip line of the lambda 4 impedance transformation section is 0.9mm;
The thickness of the sixth dielectric layer is h p F4B-M220 of (F);
the eighth dielectric layer has a thickness h m TacnoicRF-35A2;
the seventh dielectric layer has a thickness h air An air layer or a PMI foam layer.
7. A microstrip reflective array antenna according to claim 1, wherein: the feed source is a horn antenna, the horn antenna comprises a rectangular waveguide and a circular gradient which are sequentially connected, an SMA head of 50Ω is connected to the rectangular waveguide, and a metal disc with an open terminal is arranged at the top of a probe of the SMA head of 50Ω;
the overall length of the feedhorns is 137mm;
the length and width of the rectangular waveguide are 22.1mm and 13.2mm respectively;
the maximum caliber of the circular gradual change is 35mm;
the inner and outer radii of the 50Ω SMA head were 1.27mm and 4.1mm, respectively, with polytetrafluoroethylene filled between the SMA head and the metal disc.
8. A design method of a microstrip reflective array antenna is characterized in that: the method comprises the following steps:
s1, designing a unit structure to achieve 15% of standing wave bandwidth, meet the working characteristics of dual polarization, stably work at an oblique angle of 45 degrees and have caliber efficiency not less than 30%;
s2, designing a feed source, selecting a microstrip quaternary array antenna or a horn antenna as the feed source, wherein the size of the feed source is 120 multiplied by 120mm 2 The focal diameter ratio is 0.8, and the 3dB wave beam width is between 40 and 50 degrees in the working frequency band;
s3, estimating antenna caliber efficiency: according to the unit structure designed in the step S1 and the feed source directional diagram designed in the step S2, preliminary estimation is made on the aperture efficiency of the whole planar reflection array antenna;
s4, designing an array form;
s5, firstly, simulating an array in CSTIMICROWAVESTUDIO simulation software, analyzing and optimizing results, then processing experimental samples, and testing in a microwave darkroom to verify the feasibility of design.
9. The method for designing a microstrip reflective array antenna according to claim 8, wherein: the step S4 specifically comprises the following steps:
s41, dispersing phases of 0-360 degrees into 8 phase intervals by utilizing 3-bit phase quantization, and connecting the phase intervals by utilizing 8 microstrip delay lines with different lengths to form reflecting units, wherein each unit represents the phase of one phase interval:
the phase interval corresponding to the 0 degree unit is 315-360 degrees; the phase interval corresponding to the 45 DEG unit is 270-315 DEG; the phase interval corresponding to the 90 DEG unit is 215-270 DEG; the phase interval corresponding to the 135-degree unit is 180-215 degrees; the phase interval corresponding to the 180 DEG unit is 135-180 DEG; the phase interval corresponding to the 215 DEG unit is 90-135 DEG; the phase interval corresponding to the 270-degree unit is 45-90 degrees; the phase interval corresponding to the 315 DEG unit is 0-45 DEG;
S42, designing an array form according to the compensation phase required by the beam deflection angle of the array antenna:
designing an array of 12×12 array elements, i.e. array plane of 120×120mm 2 The focal diameter ratio is 0.8, the feed source selects a space side feed mode, and the coordinates of the feed source phase center are selected as (20, 60, 84) by taking the lower left corner of the array as the origin of coordinates; and then according to a phase compensation formula, calculating the phase distribution of the beam 0-degree emergent array surface, and then designing a plane reflection array form according to the phase distribution.
10. The method for designing a microstrip reflective array antenna according to claim 8, wherein: the step S5 specifically comprises the following steps:
s51, respectively carrying out space side feed by a microstrip quaternary array antenna and a horn antenna, and carrying out simulation after the feed source is obliquely incident with the antenna array surface at different angles;
s52, solving by using a CST time domain solver to obtain a simulation result;
s53, processing the horn antenna through a 3D printing technology, and integrally installing and fixing the horn antenna, the plane reflection array and the bracket; the microstrip quaternary array antenna is processed through a PCB process, and is integrally installed and fixed with the planar reflection array and the bracket;
s54, experimental tests are carried out to load patterns of different feed source plane reflection arrays at frequencies of 12.5GHz and 14.5GHz, so that test results are obtained;
S55, comparing the test result with the simulation result, and verifying feasibility.
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| CN117039433B (en) * | 2023-10-07 | 2023-12-29 | 成都辰星迅联科技有限公司 | Thin film phased array antenna and phased array antenna array based on lead patch |
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