CN116317753A - Weak magnetic control method and system for rectangular peak parity type motor - Google Patents

Weak magnetic control method and system for rectangular peak parity type motor Download PDF

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Publication number
CN116317753A
CN116317753A CN202310286604.6A CN202310286604A CN116317753A CN 116317753 A CN116317753 A CN 116317753A CN 202310286604 A CN202310286604 A CN 202310286604A CN 116317753 A CN116317753 A CN 116317753A
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motor
current
axis
value
torque
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CN116317753B (en
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赵文良
朱格非
吴昊
王秀和
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Shandong University
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Shandong University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
    • H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop

Abstract

The invention provides a weak magnetic control method and a weak magnetic control system for a rectangular peak parity type motor, which are used for determining an electromagnetic torque given value according to a measured rotor position angle and a given target rotating speed of the rectangular peak parity type motor; constructing an auxiliary function according to an electromagnetic torque equation of the torque peak homotopic motor, introducing the auxiliary function by utilizing Lagrangian multipliers under the condition of meeting MTPA to obtain an objective function, solving an extreme point of the objective function, and obtaining an alternating-axis current given value and a direct-axis current given value which meet the MTPA condition by giving the electromagnetic torque given value; judging whether the flux weakening control needs to be started according to the difference value between the limit bearing voltage of the inverter and the stator voltage, and correcting the orthogonal and direct-axis current given value when the flux weakening control is started; and generating a space vector pulse width modulation signal according to the given value of the alternating current and the direct current and the obtained actual value of the alternating current and the direct current, and driving the moment peak co-located motor to operate without demagnetizing risk.

Description

Weak magnetic control method and system for rectangular peak parity type motor
Technical Field
The invention belongs to the technical field of motor drive, and particularly relates to a torque peak parity type motor flux weakening control method and system.
Background
The statements in this section merely provide background information related to the present disclosure and may not necessarily constitute prior art.
In recent years, a motor with a peak-to-peak parity characteristic is paid attention to, and the motor utilizes an asymmetric rotor structure to enable a maximum value of permanent magnet torque and a maximum value of reluctance torque to be overlapped at the same current phase angle so as to achieve the purpose of increasing output torque. However, the topology structure of the novel motor can lead the main flux linkage vector of the rotor to lead the magnetic pole center line by 45 degrees of electric angle, so that the rotor is oriented according to the coordinate orientation rule of the alternating-direct axis (dq) of the rotor of the salient pole type permanent magnet motor (the d axis is oriented at the magnetic pole center line of the rotor, and the q axis leads the d axis by 90 degrees of electric angle), the phenomenon that the d axis and the main flux linkage vector of the rotor are mutually different by 45 degrees can occur, and the phenomenon cannot occur in the traditional salient pole type permanent magnet motor. Therefore, the mathematical model equation of the rectangular peak parity type motor is different from that of the conventional salient pole motor, and a different control system needs to be matched.
On the other hand, in recent years, the field weakening control system designed for the traditional salient pole permanent magnet motor is mature. In order to widen the speed regulation range of the permanent magnet motor, weak magnetic current is usually introduced when the motor operates in a high-speed section, so that counter electromotive force generated by the motor is reduced. However, when the permanent magnet performs field weakening control, if the field weakening current is too large, a problem of demagnetization may occur, and therefore, the field weakening control system of the motor, which does not cause the problem of demagnetization of the permanent magnet, has more reliable performance, which requires that the permanent magnet be in a field-assisted state all the time. However, research shows that the traditional non-salient pole type or salient pole type motor cannot achieve both permanent magnet flux and maximum torque current ratio (MTPA) control and permanent magnet flux and maximum torque voltage ratio (MTPV) control, and the occurrence of the torque peak parity type motor provides conditions for solving the problem. However, since the rectangular peak parity type motor has a mathematical model equation different from that of the conventional salient pole permanent magnet motor, the weak magnetic control system of the conventional motor cannot be applied to the rectangular peak parity type motor.
Disclosure of Invention
In order to overcome the defects in the prior art, the invention provides a field weakening control method for a rectangular peak parity type motor, which can consider the special topological structure of the motor, broaden the speed regulation range of the motor, and simultaneously provide a field weakening control system and method which do not need to consider the problem of demagnetization of a permanent magnet.
The invention provides a weak magnetic control method of a rectangular peak parity type motor, which comprises the following steps:
determining an electromagnetic torque given value according to the measured rotor position angle of the torque peak parity type motor and a given target rotating speed;
constructing an auxiliary function according to an electromagnetic torque equation of the torque peak homotopic motor, introducing the auxiliary function by utilizing Lagrangian multipliers under the condition of meeting MTPA to obtain an objective function, solving an extreme point of the objective function, and obtaining an alternating-axis current given value and a direct-axis current given value which meet the MTPA condition by giving the electromagnetic torque given value;
judging whether the flux weakening control needs to be started according to the difference value between the limit bearing voltage of the inverter and the stator voltage, and correcting the orthogonal and direct-axis current given value when the flux weakening control is started;
and generating a space vector pulse width modulation signal according to the given value of the alternating current and the direct current and the obtained actual value of the alternating current and the direct current, and driving the moment peak parity type motor to operate.
A second aspect of the present invention provides a torque peak parity type motor flux weakening control system, comprising: the system comprises an angular velocity processing module, a rotating speed PI controller, an MTPA module, a flux weakening correction module, a three-phase static coordinate system-two-phase rotating coordinate system converter, a d-axis/q-axis current PI controller, a two-phase rotating coordinate system-two-phase static coordinate system converter, an SVPWM module and an inverter bridge;
the rotating speed PI controller receives the set target rotating speed value and the rotating speed value of the motor rotor transmitted by the angular speed processing module as negative feedback values to calculate and form an electromagnetic torque given value;
the MTPA module calculates and obtains a given value of motor AC/DC shaft current under the MTPA condition according to an electromagnetic torque given value transmitted by the rotating speed PI controller;
the weak magnetic correction module judges whether weak magnetic is required to be started according to the difference value of the limit bearing voltage value of the inverter and the stator voltage, if so, signals are sent to the MTPA module, and the given current values of the alternating axis and the direct axis output by the MTPA module are corrected;
the three-phase stationary coordinate system-two-phase rotating coordinate system converter converts three-phase current values input by the motor into two-phase rotating coordinate system by utilizing an electric angle to obtain an alternating-direct-axis actual current value;
the d-axis/q-axis current PI controller receives the actual values of the alternating-axis current and the direct-axis current and the given value of the dq-axis current of the motor transmitted by the MTPA module, and calculates the given value of the alternating-axis voltage and the direct-axis voltage;
the two-phase rotating coordinate system-two-phase static coordinate system converter converts the given value of the AC/DC axis voltage into the two-phase static coordinate system by utilizing an electric angle to obtain the AC/DC axis voltage value;
the SVPWM module generates a three-phase PWM signal according to the alternating-current and direct-current voltage value;
and the inverter bridge receives PWM signals of the SVPWM module to generate three-phase voltage values to drive the motor to operate.
The one or more of the above technical solutions have the following beneficial effects:
according to the field weakening control system, when negative torque is output, motor current is always in a magnetism assisting state, so that the problem of demagnetization of a permanent magnet is not needed to be considered.
The maximum value of the torque-to-current ratio or the torque-to-voltage ratio is obtained through equation extremum in the MTPA module and the current correction module, so that the module can realize maximum torque-to-current ratio and maximum torque-to-voltage ratio control.
The control system disclosed by the disclosure is not only suitable for a peak-moment co-located motor, but also can be used for motors designed by the idea that the maximum value of the permanent magnet torque of the motor and the maximum value of the reluctance torque are overlapped at the same or similar current phase angles by utilizing an asymmetric rotor structure.
The current setting module of the present disclosure can be applied to model predictive control systems, and sensorless control system designs based on model predictive control systems and flux weakening control systems as described above.
Additional aspects of the invention will be set forth in part in the description which follows and, in part, will be obvious from the description, or may be learned by practice of the invention.
Drawings
The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the invention.
FIG. 1 is a control schematic diagram of a weak magnetic control system of a rectangular peak parity type motor in a permanent magnet auxiliary magnetic state according to an embodiment of the present invention;
FIG. 2 (a) is a topology structure diagram of each pole of a rectangular peak parity type motor according to the present invention;
FIG. 2 (b) shows the dq axis orientation position and the rotor main flux linkage position of each pole of the rectangular peak homotopy motor of the present invention;
fig. 3 (a) is a relationship of torque components in the motor according to the current phase angle in the first embodiment of the present invention;
FIG. 3 (b) is a graph showing the torque components as a function of current phase angle for a conventional salient pole permanent magnet motor;
FIG. 4 is a space-time unified vector diagram of each electric quantity of the motor in the first embodiment of the invention;
FIG. 5 is a graph of MTPA and constant torque direction in accordance with an embodiment of the present invention;
FIG. 6 is a schematic view of MTPV direction in accordance with a first embodiment of the present invention;
FIG. 7 is a schematic diagram of the field weakening control d, q-axis current set point in the first embodiment of the invention;
FIG. 8 is a simulated rotational speed diagram in accordance with a first embodiment of the present invention;
FIG. 9 is a simulated output torque map in accordance with a first embodiment of the invention;
FIG. 10 is a simulated d, q-axis actual current scatter plot in accordance with an embodiment of the present invention;
FIG. 11 is a control schematic diagram of a motor with a rectangular peak parity type motor according to a second embodiment of the present invention;
fig. 12 (a) is a relationship of torque in the motor according to the current phase angle in the second embodiment of the present invention;
FIG. 12 (b) is a graph showing torque as a function of current phase angle for a conventional salient pole permanent magnet motor;
FIG. 13 is a space vector diagram of each electrical quantity of a motor in a second embodiment of the present invention;
FIG. 14 is a graph of MTPA and constant torque direction in a second embodiment of the present invention;
FIG. 15 is a schematic view of MTPV direction in a second embodiment of the present invention;
FIG. 16 is a schematic diagram of the field weakening control d, q axis current setpoint in embodiment two of the invention;
FIG. 17 is a simulated rotational speed diagram in a second embodiment of the invention;
FIG. 18 is a simulated output torque map in a second embodiment of the invention;
FIG. 19 is a simulated d, q-axis actual current scatter plot in embodiment two of the invention;
FIG. 20 is a diagram showing a comparison between a field weakening control system and a general motor control system according to the second embodiment of the present invention;
FIG. 21 is a simulated rotational speed map for deceleration by output negative torque in a second embodiment of the invention;
FIG. 22 is a simulated output torque map for deceleration by output negative torque in a second embodiment of the invention;
fig. 23 is a simulated d, q-axis actual current scatter plot by output negative torque deceleration in embodiment two of the present invention.
Detailed Description
It should be noted that the following detailed description is exemplary and is intended to provide further explanation of the invention. Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this invention belongs.
It is noted that the terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of exemplary embodiments according to the present invention.
Embodiments of the invention and features of the embodiments may be combined with each other without conflict.
Example 1
The torque peak parity type motor is a novel motor, and by using an asymmetric rotor structure, the maximum value of the permanent magnet torque of the motor and the maximum value of the reluctance torque are overlapped at the same or similar current phase angle, so that the aim of outputting larger torque when the motor is electrified with sine current with the same amplitude is fulfilled. In addition, the motor is misaligned with the rotor main flux linkage vector position per pole center line, resulting in a torque equation that is different from that of a conventional salient pole permanent magnet motor.
As shown in fig. 1 to 10, this embodiment proposes a method for controlling field weakening of a motor of the peak-to-peak parity type, which includes: determining an electromagnetic torque given value according to the measured rotor position angle of the torque peak parity type motor and a given target rotating speed;
constructing an auxiliary function according to an electromagnetic torque equation of the torque peak homotopic motor, introducing the auxiliary function by utilizing Lagrangian multipliers under the condition of meeting MTPA to obtain an objective function, solving an extreme point of the objective function, and obtaining an alternating-axis current given value and a direct-axis current given value which meet the MTPA condition by giving the electromagnetic torque given value;
judging whether the flux weakening control needs to be started according to the difference value between the limit bearing voltage of the inverter and the stator voltage, and correcting the orthogonal and direct-axis current given value when the flux weakening control is started;
and generating a space vector pulse width modulation signal according to the given value of the alternating current and the direct current and the obtained actual value of the alternating current and the direct current, and driving the moment peak parity type motor to operate.
It should be noted that, the control system provided in this embodiment is not only applicable to the peak-to-moment motor used in this embodiment, but also applicable to any motor having a characteristic in which the maximum value of the permanent magnetic torque of the motor and the maximum value of the reluctance torque are overlapped at the same or similar current phase angles.
This embodiment is described by taking a rectangular peak parity type motor as an example, and as shown in fig. 1, includes: the motor comprises a moment-peak co-located motor 1, a direct current power supply 2, an inverter bridge 3, an ABC-dq converter 4, a q-axis current PI controller 5, a d-axis current PI controller 6, a dq-alpha beta converter 7, an SVPWM module 8, a photoelectric encoder 9, an angular velocity processing module 10, a rotating speed PI controller 11, an MTPA module 12, a current correction module 13, a weak magnetism judging module 14 and a current limiting module 15.
Wherein the photoelectric encoder 9 is mounted on the rotor shaft of the rectangular peak parity type motor, and measures the mechanical rotor position angle θ of the rectangular peak parity type motor m And respectively sent to the angular velocity processing modules.
An angular velocity processing module 10 for measuring a rotor position angle θ based on the photoelectric encoder 9 m Differential calculation to obtain rotation speed omega r 。
A rotational speed PI controller for calculating rotational speed omega according to the angular speed processing module r And a given target rotational speed ω r * And calculating to obtain the electromagnetic torque set value.
The MTPA module 12 is used for obtaining the given value i of the d and q axis currents of the motor under the MTPA condition according to the given value of the electromagnetic torque and by relying on an auxiliary function d * 、i q * 。
The weak magnetic judgment module 14 receives the voltage U according to the limit of the inverter smax With stator voltage U s And judging whether the flux weakening is carried out or not, namely, switching on the current correction module.
A current correction module 13 for receiving the voltage U according to the inverter limit smax With stator voltage U s And (3) calculating the correction value of the d and q axis currents of the motor.
A current limiting module 15 for limiting the given value i of the d and q axis currents of the motor d * 、i q * The value corrected by the current correction module is limited within the maximum value of the motor current.
ABC-dq converter 4 for utilizing electric angle θ e Transforming the three-phase current value input by the motor obtained by the current transformer into a dq coordinate system to obtain the actual current value i of the d axis and the q axis d And i q 。
A d-axis current PI controller 6 for controlling the motor according to the given value i of the d-axis current d * Actual value i of d-axis current d Calculating to obtain d-axis voltage given value u d * 。
A q-axis current PI controller 5 for controlling the motor according to the given value i of the q-axis current q * Q-axis current actual value i q Calculating to obtain the q-axis voltage given value u q * 。
dq-alpha beta converter 7 for utilizing electric angle theta e By giving the voltage a given value u d * 、u q * Transforming from d-q coordinate system to alpha-beta coordinate system to obtain u α And u β 。
SVPWM module 8 for giving u based on voltage α And u β And obtaining a three-phase PWM signal and sending the three-phase PWM signal to an inverter bridge module.
And the inverter bridge module is connected with the direct-current voltage source and the rectangular peak parity-type motor and is used for generating a three-phase voltage value according to the three-phase PWM signals and driving the motor to operate.
The embodiment provides a weak magnetic control method of a rectangular peak parity type motor, which comprises the following steps:
determining an electromagnetic torque given value according to the measured rotor position angle of the torque peak parity type motor and a given target rotating speed;
constructing an auxiliary function according to an electromagnetic torque equation of the torque peak homotopic motor, introducing the auxiliary function by utilizing Lagrangian multipliers under the condition of meeting MTPA to obtain an objective function, solving an extreme point of the objective function, and obtaining an alternating-axis current given value and a direct-axis current given value which meet the MTPA condition by giving the electromagnetic torque given value;
the field weakening control judges whether to start according to the actual current value, corrects the given AC-DC axis current when starting, and then generates a space vector pulse width modulation signal according to the given AC-DC axis current value and the obtained AC-DC axis current actual value to drive the moment peak parity-type motor to operate.
The topological structure of each pole of the rectangular peak parity type motor is shown in fig. 2 (a), and each pole of the motor is provided with four built-in permanent magnets which are embedded in a rotor groove; the dq axis orientation position on each pole is shown in fig. 2 (b), the direction and position of the rotor main flux linkage vector being at the angular bisector of the dq axis.
The special topological structure of the torque peak parity type motor enables the permanent magnet torque T PM And reluctance torque T re The maximum value is superimposed at the same current phase angle to obtain total torque T em As shown in fig. 3 (a), the total torque maximum is raised compared to the conventional salient pole permanent magnet motor (as shown in fig. 3 (b)), so that the motor has a greater torque density.
From the distribution of magnetic lines, a space vector diagram of the motor shown in fig. 4 is obtained, wherein i s I is a stator current space vector d 、i q I respectively s Is an alternating-direct axis component, ψ PM For the permanent magnet flux linkage generated by the permanent magnet, the permanent magnet flux linkage offset angle is 45 degrees, and psi 0 Is i s The resulting flux linkage, ψ s Is psi 0 And psi is equal to PM Is a composite flux linkage of (c).
The electromagnetic torque equation for the torque peak parity type motor is:
Figure BDA0004140075690000061
wherein p is the pole pair number of the motor, L d Is d-axis inductance, L q For q-axis inductance, T e Is electromagnetic torque.
As indicated by the dashed line in fig. 5, the constant torque direction of the torque peak motor, i.e., the tangent line of the electromagnetic torque equation, can be expressed as:
Figure BDA0004140075690000062
wherein, the liquid crystal display device comprises a liquid crystal display device,
Figure BDA0004140075690000063
in order to obtain the partial derivative,
Figure BDA0004140075690000064
in the MTPA module of the present embodiment, the relation between the current and the torque is obtained by determining that the current is equal to or greater than a predetermined valueT e Lower part(s)
Figure BDA0004140075690000065
To find i under this condition q And i d By using Lagrange extremum theorem, and introducing an auxiliary function by using Lagrange multiplier lambda to obtain:
Figure BDA0004140075690000071
according to Lagrangian extremum theorem, i is required q And i d The relation of (2) is the extreme point of the function F, namely:
Figure BDA0004140075690000072
discarding root multiplication and obtaining:
Figure BDA0004140075690000073
in order to ensure that the permanent magnet of the weak magnetic control system is always in a magnetic assisting state, the following constraint conditions are added:
i d +i q ＞0 (6)
to visually represent the above-described derivation, an MTPA graph as shown in FIG. 5 is plotted at i d -i q The constant torque curve in the plane is shown by the dotted lines, and the problem is to solve the point closest to the origin on each dotted line, and connect the points to obtain the MTPA curve, that is, the root solved by the lagrangian equation (3).
Wherein the MTPA curve contains only a portion of the booster region.
In the field weakening judgment module of the embodiment, in order to realize automatic field weakening, the limit bearing voltage U of the inverter is judged in real time smax And stator voltage U s Is realized by the difference DeltaU, and the specific process is as follows:
Figure BDA0004140075690000074
wherein U is dc Is the DC bus voltage.
Stator voltage U of rectangular peak parity type motor s The following relationship is satisfied:
Figure BDA0004140075690000075
wherein, the stator resistance is ignored, and the permanent magnet flux linkage psi of the rectangular peak parity type motor is used PM Leading the d-axis by 45 degrees, and stabilizing the d-axis voltage u of the motor d And q-axis voltage u q Can be expressed as:
Figure BDA0004140075690000081
therefore there are
Figure BDA0004140075690000082
Wherein omega r Is the electrical angular velocity of the motor.
Then deltau is:
ΔU＝U smax -U s (11)
when DeltaU is less than 0, the flux weakening is performed, namely the current correction module is turned on.
In the current correction module of the present embodiment, it is necessary to establish a motor constant torque direction and a motor voltage decreasing direction.
The motor voltage decreasing direction is established by a gradient decreasing method, and the specific process is as follows: the method comprises the following steps:
to determine the voltage drop direction, a cost function S is designed:
Figure BDA0004140075690000083
wherein, the stator resistance is ignored, and the permanent magnet flux linkage psi of the rectangular peak parity type motor is used PM Leading the d-axis by 45 degrees, and stabilizing the d-axis voltage u of the motor d And q-axis voltage u q Can be expressed as:
Figure BDA0004140075690000084
the voltage decreasing direction may be represented by a gradient descent method,
Figure BDA0004140075690000085
the gradient representing S satisfies:
Figure BDA0004140075690000086
the voltage decreasing direction is defined by (U) d ，U q ) And (3) representing.
In the current correction module, the required motor MTPV equation needs to be obtained firstly, and the deduction process is as follows:
neglecting stator resistance, permanent magnet flux linkage ψ of motor with parity type moment peaks PM The lead d-axis 45 °, the steady state torque peak on-peak motor d-axis and q-axis voltages can be expressed as:
Figure BDA0004140075690000091
at this time, since the both ends of the inverter cannot withstand the voltage without limitation, there are the following voltage equation limitations:
Figure BDA0004140075690000092
namely:
Figure BDA0004140075690000093
wherein U is max Voltage is applied to the inverter limit.
The voltage limiting equation is an ellipse with its ellipse center being
Figure BDA0004140075690000094
The long axis peak is
Figure BDA0004140075690000095
Its minor axis apex is->
Figure BDA0004140075690000096
The MTPV direction of the motor is the connection line between the motor voltage limit ellipse and the motor torque hyperbola tangent point, and the deduction process is as follows:
the motor MTPV equation can be expressed by the following formula:
Figure BDA0004140075690000097
wherein, electromagnetic torque T of torque peak parity type motor e The equation is:
Figure BDA0004140075690000098
stator voltage U of rectangular peak parity type motor s The equation is:
Figure BDA0004140075690000099
there is a case where the number of the group,
Figure BDA0004140075690000101
namely:
Figure BDA0004140075690000102
the MTPV direction is tangential to formula (22), i.e.:
Figure BDA0004140075690000103
wherein, the liquid crystal display device comprises a liquid crystal display device,
Figure BDA0004140075690000104
Figure BDA0004140075690000105
in order to ensure that the permanent magnet of the weak magnetic control system is always in a magnetic assisting state, the following constraint conditions are added in the present disclosure:
i d +i q ＞0 (24)
to intuitively represent the above derivation, an MTPV graph as shown in FIG. 6 is plotted at i d -i q The constant torque curve in the plane is shown by the dotted lines, and the problem is in fact to solve the problem of the different rotational speed (omega) from the motor at each dotted line r ) The MTPV curve is obtained by connecting the tangent points of the lower motor voltage limit circle.
The MTPV equation takes only part of the assist region, where the torque output is negative and the MTPV curve contains only part of the assist region.
In the current correction module, the field weakening region is determined by the following process:
and (3) making:
Figure BDA0004140075690000111
wherein θ represents an angle between the constant torque direction and the voltage drop direction,
Figure BDA0004140075690000112
Figure BDA0004140075690000113
when cos θ >0, the current correction module operates in the field weakening region 1 (FW 1), the correction current is given in the constant torque direction, and when cos θ <0, the current correction module operates in the field weakening region 2 (FW 2), the correction current is given in the MTPV direction.
The d-axis and q-axis weak current correction values given by the current correction module are as follows:
Figure BDA0004140075690000114
wherein alpha and beta are correction coefficients.
After the weak magnetism is turned on, the given value of the motor current is the sum of the given value of the current of the MTPA module and the weak magnetism current correction value given by the current correction module.
In the embodiment, when the motor operates at-1700 rpm and below, the motor does not turn on the flux weakening, and the d and q axis given currents of the motor are only given by the MTPA module; when the running speed of the motor exceeds-1700 rpm, the motor flux weakening module is turned on, and the d and q axes given current of the motor is a constant torque section in FIG. 7, namely a flux weakening area 1; when the motor is accelerated above-1900 rpm, the d and q axes of the motor are given current in the MTPV segment in FIG. 7, namely the field weakening region 2.
FIGS. 8, 9, and 10 show simulated waveforms for the present embodiment, U in the present embodiment dc 200V was taken.
As shown in FIG. 8, the motor is given a rotating speed of-1700 rpm at 0s and runs in idle, the motor is increased to-1700 rpm according to the rotating speed of-2 Nm constantly output by the MTPA module, the motor does not turn on weak magnetism, and the d and q axes given current of the motor is in the MTPA section; when the motor rotates at-1900 rpm and runs in no-load, the motor is started to perform weak magnetism, the motor constantly outputs electromagnetic torque of-2 Nm, and d and q axes of the motor are in a constant torque section; and 1.5s, setting-2900 rpm, enabling the motor to run in idle mode, enabling the motor to turn on weak magnetism, outputting maximum torque according to unit voltage in the MTPV direction, and setting d and q axes of the motor to be in the MTPV section. All of the d-axis currents and q-axis currents described above are shown as scatter in fig. 10.
The invention limits the given current of d and q axes, which aims to prevent the permanent magnet of the motor from demagnetizing when the motor always works in the permanent magnet auxiliary state.
Example two
As shown in fig. 11-23, the present embodiment provides a flux weakening control method for a rectangular peak parity type motor, in which, as shown in fig. 11, a photoelectric encoder 9 is mounted on a rotor shaft of the rectangular peak parity type motor, and a rotor position angle θ of the rectangular peak parity type motor is measured m And performing angular velocity processing to obtain the actual rotation speed omega r Optionally, differentiating the rotor position angle to obtain the actual rotation speed of the rectangular peak homotopic motor. The rotational speed PI controller 11 calculates the rotational speed ω according to the angular velocity processing module 10 r And a given target rotational speed ω r * And calculating to obtain the electromagnetic torque set value.
The MTPA module 12 constructs an auxiliary function according to an electromagnetic torque equation of the torque peak co-located motor, introduces the auxiliary function by using lagrangian multipliers to obtain an objective function under the condition of meeting the MTPA, solves an extreme point of the objective function, and further obtains an alternating-axis current given value and a direct-axis current given value which meet the MTPA condition by giving an electromagnetic torque given value.
Specifically, as shown in fig. 2 (a), each pole of the rectangular peak parity type motor has four built-in permanent magnets which are respectively embedded in the rotor grooves; the dq-axis (i.e., the ac-dc axis) orientation position on each pole is shown in fig. 2 (b), which also marks the direction and position of the main flux linkage vector of the rotor, i.e., at the angular bisector of the dq-axis.
The special topological structure of the motor enables the permanent magnet torque T PM And reluctance torque T re The maximum value is superimposed at the same current phase angle to obtain total torque T em As shown in fig. 12 (a), the total torque maximum is raised compared to the conventional salient pole permanent magnet motor (as shown in fig. 12 (b)), so that the motor has a greater torque density.
From the distribution of magnetic lines, a motor space vector diagram is obtained as shown in FIG. 13, where i s I is a stator current space vector d 、i q Respectively the quadrature and direct currents (i.e.) s The alternating and direct axis components) ψ PM For the permanent magnet flux linkage generated by the permanent magnet, the permanent magnet flux linkage offset angle is 45 degrees, and psi 0 Is i s The resulting flux linkage, ψ s Is psi 0 And psi is equal to PM Is a composite flux linkage of (c).
The electromagnetic torque equation for the torque peak parity type motor is:
Figure BDA0004140075690000121
wherein p is the pole pair number of the motor, T e Is electromagnetic torque, L d Is a direct axis inductance L q Is the quadrature axis inductance.
The constant torque direction of the torque peak parity type motor, i.e. the tangent line of the electromagnetic torque equation, can be expressed as:
Figure BDA0004140075690000122
i.e. in the tangential direction of the dashed line in fig. 14.
The relation between current and torque in the MTPA (maximum torque to current ratio) module is required, i.e. at a certain T e Lower part(s)
Figure BDA0004140075690000123
To find i under this condition q And i d By using the Lagrange extremum theorem and introducing an auxiliary function by using Lagrange multiplier lambda, the objective function F is obtained as follows:
Figure BDA0004140075690000131
according to Lagrangian extremum theorem, i is required q And i d The relation of (a) is that of the above objectThe extreme points of the scalar function F are thus:
Figure BDA0004140075690000132
eliminating root and solving to obtain
Figure BDA0004140075690000133
In order to ensure the weak magnetic performance, the following constraint conditions are added:
i d +i q ≤0 (33)
to intuitively represent the above derivation, an MTPA graph as shown in FIG. 14 is plotted at i d -i q The constant torque curve in the plane is shown by the dotted lines, the problem is to solve the point closest to the origin on each dotted line, and connect the points to obtain the MTPA curve, that is, the root solved by the lagrangian equation, so as to obtain the given values of the alternating and direct axis currents meeting the MTPA condition.
As an alternative embodiment, in order to realize automatic field weakening, the limit bearing voltage U of the inverter is judged in real time smax And stator voltage U s Is realized by the difference DeltaU, and the specific process is as follows:
inverter limit bearing voltage U smax The following relationship is satisfied:
Figure BDA0004140075690000134
wherein U is dc Is the DC bus voltage.
Determining stator voltage based on actual values of ac and dc currents of the rectangular peak homomorphic motor, in particular implementation, measuring three-phase current values of rectangular peak homomorphic motor by current transformer, ABC-dq converter 4 uses electric angle θ e Transforming the three-phase current values obtained by the current transformer into dq coordinate system to obtain d-axis and q-axis actual current values i d And i q . And determining the stator voltage according to the actual current values of the d axis and the q axis, and determining the stator voltage U of the rectangular peak parity type motor s The following relationship is satisfied:
Figure BDA0004140075690000135
wherein, the stator resistance is ignored, and the permanent magnet flux linkage psi of the rectangular peak parity type motor is used PM Leading the d-axis by 45 degrees, and stabilizing the d-axis voltage u of the motor d And q-axis voltage u q Can be expressed as:
Figure BDA0004140075690000141
therefore there are
Figure BDA0004140075690000142
Define Δu as: Δu=u smax -U s And judging whether to perform flux weakening control according to the difference value between the stator voltage and the DC bus voltage value.
Specifically, when Δu <0, the flux weakening is performed, that is, the current correction module is turned on.
Further, by a gradient descent method, a motor voltage descending direction is established, and the specific process is as follows:
to determine the voltage drop direction, a cost function S is designed:
Figure BDA0004140075690000143
wherein, the stator resistance is ignored, and the permanent magnet flux linkage psi of the rectangular peak parity type motor is used PM Leading the d-axis by 45 degrees, and stabilizing the d-axis voltage u of the motor d And q-axis voltage u q Can be expressed as:
Figure BDA0004140075690000144
the voltage decreasing direction may be represented by a gradient decreasing method, v S representing the gradient of S, satisfying:
Figure BDA0004140075690000145
the voltage decreasing direction is defined by (U) d ，U q ) And (3) representing.
The MTPV equation of the motor is required to be obtained, the motor voltage limit ellipse is required to be obtained, and the deduction process is as follows:
neglecting stator resistance, permanent magnet flux linkage ψ of motor with parity type moment peaks PM The lead d-axis 45 °, the steady state torque peak on-peak motor d-axis and q-axis voltages can be expressed as:
Figure BDA0004140075690000151
at this time, since the both ends of the inverter cannot withstand the voltage without limitation, there are the following voltage equation limitations:
Figure BDA0004140075690000152
i.e.
Figure BDA0004140075690000153
The voltage limiting equation is an ellipse with its ellipse center being
Figure BDA0004140075690000154
The long axis peak is
Figure BDA0004140075690000155
Its minor axis apex is->
Figure BDA0004140075690000156
The MTPV equation of the motor is the connection between the motor voltage limit ellipse and the tangent point of the motor torque curve, and the deduction process is as follows: the motor MTPV equation can be represented by
Figure BDA0004140075690000157
Wherein, electromagnetic torque T of torque peak parity type motor e The equation is:
Figure BDA0004140075690000158
stator voltage U of rectangular peak parity type motor s The equation is:
Figure BDA0004140075690000159
there is a case where the number of the group,
Figure BDA00041400756900001510
i.e.
Figure BDA0004140075690000161
The MTPV direction being tangential to the above, i.e
Figure BDA0004140075690000162
In order to ensure the weak magnetic performance, the following constraint conditions are added:
i d +i q ≤0 (50)
to intuitively represent the above derivation, an MTPV graph as shown in FIG. 15 is plotted at i d -i q The constant torque curve in the plane is shown by the dotted line, the above problem is in fact that each is solvedThe dotted line is different from the motor in rotation speed (omega r ) The MTPV curve is obtained by connecting the tangent points of the lower motor voltage limit circle.
The weak magnetic area of the current correction module is determined by the following process:
order the
Figure BDA0004140075690000163
Here, θ represents an angle between the constant torque direction and the voltage drop direction;
wherein, the liquid crystal display device comprises a liquid crystal display device,
Figure BDA0004140075690000164
Figure BDA0004140075690000171
Figure BDA0004140075690000172
when cos theta >0, the current correction module operates in a weak magnetic area 1 (FW 1) and gives a correction current along the constant torque direction, and when cos theta <0, the current correction module operates in a weak magnetic area 2 (FW 2) and gives a correction current along the MTPV direction;
the d-axis weak current correction value and the q-axis weak current correction value given by the current correction module are as follows:
Figure BDA0004140075690000173
here, α, β are correction coefficients.
After the flux weakening is turned on, the given value of the motor current is the sum of the given value of the current of the MTPA module and the alternating-axis and direct-axis flux weakening correction current given by the current correction module.
Optionally, the d and q axis current setpoint of the motor is also limited to be within the maximum current of the motor by the current limiting module 15 before the ac and dc axis current setpoint is corrected.
Further, byThe current transformer measures three-phase current values of the rectangular peak parity type motor 1, and the ABC-dq converter 4 utilizes the electric angle theta e Transforming the three-phase current value into dq coordinate system to obtain d-axis and q-axis actual current value i d And i q The method comprises the steps of carrying out a first treatment on the surface of the The d-axis current PI controller 6 is used for controlling the d-axis current of the motor according to the given value i of the d-axis current of the motor d * Actual value i of d-axis current d Calculating to obtain d-axis voltage given value u d * The method comprises the steps of carrying out a first treatment on the surface of the The q-axis current PI controller 5 controls the motor to drive the motor in accordance with the given value i of the q-axis current of the motor q * Q-axis current actual value i q Calculating to obtain the q-axis voltage given value u q * The method comprises the steps of carrying out a first treatment on the surface of the The dq-alpha beta converter 7 uses the electrical angle theta e Transforming the given value of the AC-DC axis voltage from d-q coordinate system to alpha-beta coordinate system to obtain u α And u β The method comprises the steps of carrying out a first treatment on the surface of the Ac-dc axis voltage given value u α And u β The space vector pulse width modulation signal (i.e. three-phase PWM signal) is generated by a signal generating module 8 (i.e. SVPWM module) and is sent to an inverter bridge 3, and the inverter bridge 3 is connected with a direct-current voltage source 2 and a rectangular peak parity type motor 1 and is used for generating three-phase voltage values according to the three-phase PWM signal to drive the motor to operate.
The given values of the d and q axes currents of the weak magnetic control in the embodiment are shown in fig. 16, when the motor operates at 700rpm and below, the motor does not turn on weak magnetic, and the given currents of the d and q axes of the motor are only given by the MTPA module; when the motor runs to more than 700rpm, the motor flux weakening module is turned on, and the d and q axes given current of the motor is a constant torque section in fig. 16, namely a flux weakening area 1; when the motor is accelerated to above 1100rpm, the d and q axes given current of the motor is limited by the current limiting module, the d and q axes given current of the motor is the limit current circle section in fig. 16, and when the motor is accelerated to above 3600rpm, the d and q axes given current of the motor is the MTPV section in fig. 16, namely the weak magnetic area 2.
FIGS. 17, 18, and 19 show simulated waveforms for the present embodiment, U in the present embodiment dc 100V was taken.
As shown in fig. 17, the motor is given a rotation speed of 700rpm at 0s, the motor constantly outputs a torque of 5Nm according to the MTPA module, the rotation speed is increased to 700rpm, and the d and q axes given currents of the motor are in the MTPA section; then carrying out 3Nm load at 0.5s, wherein the motor constantly outputs electromagnetic torque of 3Nm according to the MTPA module, and the d and q axes given current of the motor is in the MTPA section; when 1s, setting 1100rpm and enabling the motor to run in idle load, switching on the flux weakening of the motor, constantly outputting electromagnetic torque of 4Nm by the motor, and setting d and q axes of the motor to be in a constant torque section; 1.5s, setting 3600rpm and enabling the motor to run in idle load, switching on weak magnetism of the motor, wherein the output torque of the motor is continuously reduced due to the limitation of a current limit circle, and the d-axis and q-axis given currents of the motor are positioned in a limit current circle section; and 3s, setting 9000rpm, enabling the motor to run in a no-load mode, enabling the motor to turn on weak magnetism, outputting maximum torque according to unit voltage in the MTPV direction, and setting d and q axes of the motor to be in an MTPV section. All of the d-axis currents and q-axis currents described above are shown as scattered points in fig. 19.
Simulation results show that the weak magnetic control system designed by the embodiment has good performance, can realize the no-difference tracking of the rotating speed, and the d and q axes actual current of the motor is identical with the expected given value.
FIG. 20 is a graph showing the comparison of the motor speed ranges for the flux weakening control system provided in this embodiment with the U in this embodiment dc 100V was taken. As shown in fig. 20, when the motor outputs a load of 2Nm, the speed regulation range of the motor can be widened from 1660rpm to 3070rpm by adopting the field weakening control system, which represents good performance of the method provided by the embodiment in specific application.
Similarly, the method is also suitable for the working condition of outputting negative torque, and fig. 21, 22 and 23 show simulation waveforms of the embodiment of outputting negative torque to perform motor deceleration, and similarly, U in the embodiment dc 100V was taken.
As shown in fig. 21, the motor was given 6000rpm at 0s, and the motor was rapidly raised to the given rpm; setting 2700rpm at 3s, switching on weak magnetism of the motor, and setting d and q axes of the motor to be in MTPV section; when the rotating speed is set to 1800rpm, the motor is switched on for weak magnetism, the negative torque output by the motor is increased along with the rising of the rotating speed due to the limitation of a current limit circle, and the d-axis and q-axis given currents of the motor are positioned in a limit current circle section; when the rotating speed of 1150rpm is set at 6s, the motor turns on weak magnetism and constantly outputs negative torque of-2 Nm, and the d and q axes of the motor are in a constant torque section; and 7s, the motor is not switched on with weak magnetism at a given rotating speed of 0rpm, negative torque of-2 Nm is constantly output, and d and q axes of the motor are in MTPA section at given current. All of the d-axis currents and q-axis currents described above are shown as scattered points in fig. 23. Simulation results show that the flux weakening control system disclosed by the disclosure is also suitable for the working condition of outputting negative torque for deceleration.
According to the above solving process, a control system block diagram is designed as shown in fig. 11.
When the control system works, firstly, the angle signal of the motor is collected by the photoelectric encoder 9 and used for angular velocity processing, and the angular velocity signal omega is obtained r With a given target rotational speed omega r * And taking difference to form a negative feedback channel, calculating a difference signal by a rotating speed PI controller 11 to obtain a motor electromagnetic torque given value, giving a current given value under the MTPA condition by an MTPA module, combining a weak magnetic judgment module 14, entering a current correction module 13 to obtain motor correction current, adding the given current of the MTPA module and the given current of the current correction module, and obtaining a motor d-axis and q-axis current given value by a current limiting module 15. The d-axis and q-axis current set value is differenced with the d-axis and q-axis current actual value obtained by ABC-dq coordinate transformation, and the voltage set value u is obtained by calculation through two current PI controllers 6 and 5 d * And u q * Then the voltage given u under the alpha-beta coordinate system is obtained through dq-alpha beta coordinate transformation α And u β After the signal generating module 8, an ABC three-phase PWM signal can be obtained and input to the inverter bridge 3, and then three-phase voltage values required by the driving motor can be generated.
Example two
The embodiment provides a torque peak parity type motor field weakening control system, adopts and includes:
a torque peak parity type motor flux weakening control system comprising: the system comprises an angular velocity processing module, a rotating speed PI controller, an MTPA module, a flux weakening correction module, a three-phase static coordinate system-two-phase rotating coordinate system converter, a d-axis/q-axis current PI controller, a two-phase rotating coordinate system-two-phase static coordinate system converter, an SVPWM module and an inverter bridge;
the rotating speed PI controller receives the set target rotating speed value and the rotating speed value of the motor rotor transmitted by the angular speed processing module as negative feedback values to calculate and form an electromagnetic torque given value;
the MTPA module calculates and obtains a given value of motor AC/DC shaft current under the MTPA condition according to an electromagnetic torque given value transmitted by the rotating speed PI controller;
the weak magnetic correction module judges whether weak magnetic is required to be started according to the difference value of the limit bearing voltage value of the inverter and the stator voltage, if so, signals are sent to the MTPA module, and the given current values of the alternating axis and the direct axis output by the MTPA module are corrected;
the three-phase stationary coordinate system-two-phase rotating coordinate system converter converts three-phase current values input by the motor into two-phase rotating coordinate system by utilizing an electric angle to obtain an alternating-direct-axis actual current value;
the d-axis/q-axis current PI controller receives the actual values of the alternating-axis current and the direct-axis current and the given value of the dq-axis current of the motor transmitted by the MTPA module, and calculates the given value of the alternating-axis voltage and the direct-axis voltage;
the two-phase rotating coordinate system-two-phase static coordinate system converter converts the given value of the AC/DC axis voltage into the two-phase static coordinate system by utilizing an electric angle to obtain the AC/DC axis voltage value;
the SVPWM module generates a three-phase PWM signal according to the alternating-current and direct-current voltage value;
and the inverter bridge receives PWM signals of the SVPWM module to generate three-phase voltage values to drive the motor to operate.
It will be appreciated by those skilled in the art that the modules or steps of the invention described above may be implemented by general-purpose computer means, alternatively they may be implemented by program code executable by computing means, whereby they may be stored in storage means for execution by computing means, or they may be made into individual integrated circuit modules separately, or a plurality of modules or steps in them may be made into a single integrated circuit module. The present invention is not limited to any specific combination of hardware and software.
While the foregoing description of the embodiments of the present invention has been presented in conjunction with the drawings, it should be understood that it is not intended to limit the scope of the invention, but rather, it is intended to cover all modifications or variations within the scope of the invention as defined by the claims of the present invention.

Claims (10)

1. A torque peak parity type motor flux weakening control method is characterized by comprising the following steps:
determining an electromagnetic torque given value according to the measured rotor position angle of the torque peak parity type motor and a given target rotating speed;
constructing an auxiliary function according to an electromagnetic torque equation of the torque peak homotopic motor, introducing the auxiliary function by utilizing Lagrangian multipliers under the condition of meeting MTPA to obtain an objective function, solving an extreme point of the objective function, and obtaining an alternating-axis current given value and a direct-axis current given value which meet the MTPA condition by giving the electromagnetic torque given value;
judging whether the flux weakening control needs to be started according to the difference value between the limit bearing voltage of the inverter and the stator voltage, and correcting the orthogonal and direct-axis current given value when the flux weakening control is started;
and generating a space vector pulse width modulation signal according to the given value of the alternating current and the direct current and the obtained actual value of the alternating current and the direct current, and driving the moment peak parity type motor to operate.
2. The method for flux weakening control of a motor of the peak-to-peak type according to claim 1, wherein an electromagnetic torque equation is determined based on a topology of each pole of the motor of the peak-to-peak type, the electromagnetic torque equation being expressed as:
Figure FDA0004140075680000011
wherein T is e Is electromagnetic torque, p is the pole pair number of the motor, and psi PM Permanent magnet produced for permanent magnetFlux linkage, i q For the quadrature current, i d Is a direct axis current, L d Is a direct axis inductance L q Is the quadrature axis inductance.
3. The method for flux weakening control of a motor according to claim 1, wherein the objective function F is expressed as:
Figure FDA0004140075680000012
wherein lambda is Lagrangian multiplier, p is motor pole pair number, L d Is d-axis inductance, L q For q-axis inductance, i d 、i q I respectively s Is an alternating-direct axis component, ψ PM Permanent magnet flux linkage generated for permanent magnet, T e Is electromagnetic torque.
4. The method for controlling the flux weakening of a motor with the same moment peak as defined in claim 3, wherein when the extreme point of the objective function is solved, in order to ensure that the permanent magnet is under the auxiliary magnetic strip, the constraint condition is added that:
i d +i q ＞0
wherein i is q For the quadrature current, i d Is a straight axis current;
or, in order to ensure the weak magnetic performance, constraint conditions are added as follows:
i d +i q ≤0
wherein i is q For the quadrature current, i d Is a straight axis current.
5. The method for controlling the flux weakening of a motor with the same torque peak as claimed in claim 1, wherein the decreasing direction of the motor voltage is determined by a gradient descent method, and the constant torque direction of the motor is further determined; and determining the MTPV direction of the motor through a connecting line of a motor voltage limit ellipse and a motor torque hyperbola tangent point.
6. The method for controlling flux weakening of a motor according to claim 5, wherein the current correction direction is determined based on cosθ, which is the cosine of the angle between the constant torque direction and the voltage drop direction, and when cosθ >0, the correction current is given along the corresponding constant torque direction; when cos θ <0, a correction current is given along the MTPV direction.
7. The method for controlling the flux weakening of a motor according to claim 1, wherein the correction value of the motor's ac and dc currents is calculated based on the difference between the limit bearing voltage value of the inverter and the stator voltage, and the current correction is performed on the ac and dc current set values of the MTPA condition based on the correction value of the motor's ac and dc currents.
8. The method for controlling flux weakening of a motor according to claim 1, wherein a voltage set point is determined based on the ac and dc current set points and the obtained ac and dc current actual values; and after the voltage set value is subjected to coordinate transformation, generating a space vector pulse width modulation signal for driving the operation of the rectangular peak parity type motor.
9. A torque peak parity type motor flux weakening control system, comprising: the system comprises an angular velocity processing module, a rotating speed PI controller, an MTPA module, a flux weakening correction module, a three-phase static coordinate system-two-phase rotating coordinate system converter, a d-axis/q-axis current PI controller, a two-phase rotating coordinate system-two-phase static coordinate system converter, an SVPWM module and an inverter bridge;
the rotating speed PI controller receives the set target rotating speed value and the rotating speed value of the motor rotor transmitted by the angular speed processing module as negative feedback values to calculate and form an electromagnetic torque given value;
the MTPA module calculates and obtains a given value of motor AC/DC shaft current under the MTPA condition according to an electromagnetic torque given value transmitted by the rotating speed PI controller;
the weak magnetic correction module judges whether weak magnetic is required to be started according to the difference value of the limit bearing voltage value of the inverter and the stator voltage, if so, signals are sent to the MTPA module, and the given current values of the alternating axis and the direct axis output by the MTPA module are corrected;
the three-phase stationary coordinate system-two-phase rotating coordinate system converter converts three-phase current values input by the motor into two-phase rotating coordinate system by utilizing an electric angle to obtain an alternating-direct-axis actual current value;
the d-axis/q-axis current PI controller receives the actual values of the alternating-axis current and the direct-axis current and the given value of the dq-axis current of the motor transmitted by the MTPA module, and calculates the given value of the alternating-axis voltage and the direct-axis voltage;
the two-phase rotating coordinate system-two-phase static coordinate system converter converts the given value of the AC/DC axis voltage into the two-phase static coordinate system by utilizing an electric angle to obtain the AC/DC axis voltage value;
the SVPWM module generates a three-phase PWM signal according to the alternating-current and direct-current voltage value;
and the inverter bridge receives PWM signals of the SVPWM module to generate three-phase voltage values to drive the motor to operate.
10. The torque peak parity type motor flux weakening control system according to claim 9, wherein in the MTPA module, to ensure that the permanent magnet is under the auxiliary magnetic stripe, the constraint condition is that:
i d +i q ＞0
wherein i is q For the quadrature current, i d Is a straight axis current;
or, in order to ensure the weak magnetic performance, constraint conditions are added as follows:
i d +i q ≤0
wherein i is q For the quadrature current, i d Is a straight axis current.
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