CN116203639A - Towed frequency domain electromagnetic detection depth focusing emission system and method - Google Patents
Towed frequency domain electromagnetic detection depth focusing emission system and method Download PDFInfo
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Abstract
The invention belongs to the field of towed frequency domain electromagnetic detection, and relates to a towed frequency domain electromagnetic detection depth focusing emission system and a towed frequency domain electromagnetic detection depth focusing emission method, wherein the towed frequency domain electromagnetic detection depth focusing emission method comprises the following steps: determining an effective frequency band required by the exploration target area by utilizing a skin depth formula according to the depth range of the exploration target area; combining the requirement of the investigation target area on the longitudinal detection resolution, and determining the number and distribution of the required main frequencies; the set objective function is obtained by maximizing the objective function z (S M ) Obtaining a switching time sequence; and calculating according to the switching time sequence to obtain the actual frequency spectrum amplitude of the transmitting current. The proposed waveforms have a more reasonable spectral energy distribution and less harmonic pollution. Therefore, high vertical resolution in the target detection range can be ensured. Furthermore, a corresponding wideband resonance compensator is provided to cancel the inductance caused by the transmitting coil in a wideband range, improve the current amplitude of each main frequency in the effective frequency band range, and suppress the harmonic wave outside the main frequency.
Description
Technical Field
The invention belongs to the field of towed frequency domain electromagnetic detection, and particularly relates to a towed frequency domain electromagnetic detection depth focusing and transmitting system and method.
Background
With the rapid development of economy, the resources which are easy to be adopted and explored are reduced year by year. In order to improve the exploitation efficiency and reduce the exploration cost, the research direction of geophysical exploration gradually changes to high-precision and fine exploration. The frequency domain electromagnetic method is an important geophysical detection method, and an automobile dragging transmitting coil provided with a transmitter and a receiver is adopted, and a receiving coil sensor is used for completing detection experiments. The method has the advantages of high efficiency and low cost, and is widely applied to the fields of geological investigation, environmental evaluation and the like. According to the principle of a frequency domain electromagnetic method, a series of electromagnetic field signals with different frequencies can be sent to the ground, so that the electrical information of different depths of the ground can be detected. To ensure high vertical resolution for a target of a particular depth, the excitation frequency should be high and the energy limited to the effective frequency band to achieve depth-focused detection. Meanwhile, the current amplitude of the effective main frequency is higher than the noise floor of the system, so that the acquired data result is meaningful. Because the coil inductance is usually larger, the coil inductance can obviously inhibit the emission current, and the energy of high-frequency detection frequency and the intensity of an electromagnetic field transmitted into the ground are reduced. Meanwhile, the frequency spectrum energy decreases with the increase of frequency, and the detection precision of each main frequency is greatly different [9]. Therefore, high-precision detection experiments place higher demands on the performance of the transmitting system.
In order to improve the detection efficiency, the multifrequency fusion excitation signal is widely adopted, namely, the frequency spectrum of the transmitted waveform contains a plurality of available main frequencies so as to detect stratum with a plurality of depths at the same time. Constable in 1996 proposed a waveform that evenly distributes most of the spectral energy over the first and third harmonics. Srnka proposes a representative waveform with spectral energy equally divided between the first, second and fourth harmonics. The Zhou uses a pseudo-random binary sequence as the transmit waveform, which has a fixed frequency ratio. The signal has a distributed dominant frequency, the spectral energy cannot be limited in the effective frequency band, and the signal is not suitable for high-precision detection in a small-scale area, and the dominant frequency is required to be distributed in the effective narrow frequency band to ensure high vertical resolution. In the high-precision detection experiment of a target with a specific depth, multiple emission experiments with different fundamental frequencies need to be performed by using a pseudo-random waveform. Gao Lihui it is proposed that the deep focusing waveform based on the SHEPWM method focuses the main spectrum energy on the seventh to eleventh harmonics, so that a better detection effect can be obtained in the electrical source detection experiment. However, the ten and higher harmonics are not controllable, and thus high frequency harmonics are severely contaminated. Meanwhile, the spectrum energy of the traditional transmitting current is limited under the influence of the inductance of the transmitting coil, and the requirement of high-precision detection is difficult to meet. Liu Changsheng a three-frequency resonant matching circuit is provided to counteract the impedance caused by the coil inductance, thereby improving the accuracy of magnetic source detection experiments. The circuit is mainly aimed at the main frequency dispersed emission waveform, the passband of each resonance point is narrower, and the circuit is not suitable for the integral promotion of the current value of the depth focusing waveform in the effective frequency band.
Disclosure of Invention
Aiming at the problems that the frequency spectrum energy of the emission current in the prior art is limited and the requirement of high-precision detection is difficult to meet, the invention aims to provide a towed frequency domain electromagnetic detection depth focusing emission system and a towed frequency domain electromagnetic detection depth focusing emission method.
The present invention has been achieved in such a way that,
a towed frequency domain electromagnetic probe depth focus emission method, the method comprising:
determining an effective frequency band required by the exploration target area by utilizing a skin depth formula according to the depth range of the exploration target area;
combining the requirement of the investigation target area on the longitudinal detection resolution, and determining the number and distribution of the required main frequencies;
the set objective function is as follows:
z(S M )＝w 1 ·f 1 (S M )+w 2 ·f 2 (S M )
wherein w is 1 ,w 2 Respectively are weight coefficients, S M For the sequence of switching moments,
the estimated desired spectrum is I n ，J n For current amplitude and phase of the N-th harmonic, the detected dominant frequency is { N 1 ,N 2 ,...,N i ,...,N L }；
The constraint condition is that
t 0 ＜t 1 ＜....＜t M
By maximizing the objective function z (S M ) Obtaining a switching time sequence;
and calculating according to the switching time sequence to obtain the actual frequency spectrum amplitude of the transmitting current.
Further, the calculating according to the switching time sequence to obtain the actual spectrum amplitude of the emission current includes: calculated by the following formula:
wherein L is 0 And R is 0 Equivalent inductance and equivalent resistance of the transmitting coil, J n The current amplitude and phase being the n-order harmonics,
n represents harmonic order, ω n Is the angular frequency of the nth harmonic, T 0 For the period, U (t) is the excitation voltage of the electromagnetic probe.
Further, the method also comprises the step of adjusting the amplitude of the main frequency current by arranging a broadband resonance compensator on the transmitting circuit, wherein the broadband resonance compensator is connected in series with the transmitting coil and comprises an inductance L connected in parallel 1 And capacitor C 1 。
Further, the impedance of the main frequency point in the main frequency band is adjusted by the broadband resonance compensator, specifically, the total impedance of the broadband resonance compensator and the transmitting coil is expressed as:
where// is the parallel operator, ω is the angular frequency of the emission current, L 0 Is the inductance of the transmitting coil, R 0 Is the internal equivalent resistance of the transmitting coil and the bridge circuit;
minimizing the total impedance so that the imaginary part j is 0 and the main frequency band range of the depth focusing waveform is f L ～f H Set the resonance center frequency f r The center frequency f of the resonance band is the center of the main frequency band according to the relationship omega=2pi f of the angular frequency and the frequency r Represented as
The parameters for obtaining the following calculated wideband resonance compensator are:
by adjusting inductance L 1 And capacitor C 1 And (3) adjusting the impedance of the dominant frequency point in the dominant frequency band.
A towed frequency domain electromagnetic probe depth focus transmit system, comprising: a transmit waveform design module:
determining an effective frequency band required by the exploration target area by utilizing a skin depth formula according to the depth range of the exploration target area;
combining the requirement of the investigation target area on the longitudinal detection resolution, and determining the number and distribution of the required main frequencies;
the set objective function is as follows:
z(S M )＝w 1 ·f 1 (S M )+w 2 ·f 2 (S M )
wherein w is 1 ,w 2 Respectively are weight coefficients, S M For the sequence of switching moments,
the estimated desired spectrum is I n ，J n For current amplitude and phase of the N-th harmonic, the detected dominant frequency is { N 1 ,N 2 ,...,N i ,...,N L }；
The constraint condition is that
t 0 ＜t 1 ＜....＜t M
By maximizing the objective function z (S M ) Obtaining a switching time sequence;
and calculating according to the switching time sequence to obtain the actual frequency spectrum amplitude of the transmitting current.
Further, the calculating according to the switching time sequence to obtain the actual spectrum amplitude of the emission current includes: calculated by the following formula:
wherein L is 0 And R is 0 Equivalent inductance and equivalent resistance of the transmitting coil, J n The current amplitude and phase being the n-order harmonics,
n represents harmonic order, ω n Is the angular frequency of the nth harmonic, T 0 For the period, U (t) is the excitation voltage of the electromagnetic probe.
Further, the transmitting system further comprises a wideband resonance compensator arranged on the transmitting circuit for adjusting the amplitude of the main frequency current, wherein the wideband resonance compensator is connected in series with the transmitting coil and comprises an inductance L connected in parallel 1 And capacitor C 1 。
Further, the impedance of the main frequency point in the main frequency band is adjusted by the broadband resonance compensator, specifically, the total impedance of the broadband resonance compensator and the transmitting coil is expressed as:
where// is the parallel operator, ω is the angular frequency of the emission current, L 0 Is the inductance of the transmitting coil, R 0 Is the internal equivalent resistance of the transmitting coil and the bridge circuit;
minimizing the total impedance so that the imaginary part j is 0 and the main frequency band range of the depth focusing waveform is f L ～f H Set the resonance center frequency f r The center frequency f of the resonance band is the center of the main frequency band according to the relationship omega=2pi f of the angular frequency and the frequency r Represented as
The parameters for obtaining the following calculated wideband resonance compensator are:
by adjusting inductance L 1 And capacitor C 1 And (3) adjusting the impedance of the dominant frequency point in the dominant frequency band.
Compared with the prior art, the invention has the beneficial effects that:
the invention designs the depth focusing emission waveform with the main frequency distributed tightly based on a nonlinear programming method. Compared with the emission waveform obtained by the traditional SHEPWM method, the provided waveform has more reasonable spectrum energy distribution and less harmonic pollution. Therefore, high vertical resolution in the target detection range can be ensured. Furthermore, a corresponding wideband resonance compensator is provided to cancel the inductance caused by the transmitting coil in a wideband range, improve the current amplitude of each main frequency in the effective frequency band range, and suppress the harmonic wave outside the main frequency. The designed passband width setting of the resonance compensator has higher flexibility and can be matched with the depth focusing waveforms of different main bandwidths. The feasibility of the proposed scheme is verified through simulation and experiments.
Drawings
FIG. 1 is a schematic diagram of a wideband resonance compensator according to an embodiment of the present invention, wherein the circuit of FIG. 1 (a) is shown in FIG. 1 (b);
fig. 2 is a diagram of a transmitting system structure according to an embodiment of the present invention;
FIG. 3 SHEPWM-based deep focus emission method, simulation results without resonance compensator (a) current waveform (b) current spectrum;
fig. 4 shows a capacitor C according to an embodiment of the present invention 1 Inductance L 1 A variable parameter value curve;
FIG. 5 shows the impedance versus inductance L for different dominant frequencies according to an embodiment of the present invention 1 Is a change curve of (2);
FIG. 6 is a graph showing the impedance versus frequency according to an embodiment of the present invention;
FIG. 7 shows simulation results of a SHEPWM-based deep focus transmit scheme with a resonance compensator (a) current waveform (b) current spectrum, according to an embodiment of the present invention;
FIG. 8 is a simulation result (a) current waveform (b) current spectrum without a resonance compensator for providing the proposed five-frequency deep focus emission scheme according to an embodiment of the present invention;
FIG. 9 shows simulation results of a five-frequency deep focus transmit scheme with a resonance compensator (a) current waveform (b) current spectrum;
FIG. 10 shows simulation results of a five-frequency deep focus transmit scheme with a resonant compensator (a) current waveform (b) current spectrum;
FIG. 11 shows different inductances L in an embodiment of the invention 1 A current gain variation curve with frequency;
fig. 12 illustrates a five-frequency deep focus transmit scheme with current spectra of different resonance compensators (a) wideband resonance compensator (b) narrowband resonance compensator.
Detailed Description
The present invention will be described in further detail with reference to the following examples in order to make the objects, technical solutions and advantages of the present invention more apparent. It should be understood that the specific embodiments described herein are for purposes of illustration only and are not intended to limit the scope of the invention.
The invention provides a towed frequency domain electromagnetic detection depth focusing emission method, which comprises the following steps:
determining an effective frequency band required by the exploration target area by utilizing a skin depth formula according to the depth range of the exploration target area;
combining the requirement of the investigation target area on the longitudinal detection resolution, and determining the number and distribution of the required main frequencies;
the set objective function is as follows:
z(S M )＝w 1 ·f 1 (S M )+w 2 ·f 2 (S M )
wherein w is 1 ,w 2 Respectively are weight coefficients, S M For the sequence of switching moments,
the estimated desired spectrum is I n ，J n For current amplitude and phase of the N-th harmonic, the detected dominant frequency is { N 1 ,N 2 ,...,N i ,...,N L }；
The constraint condition is that
t 0 ＜t 1 ＜....＜t M
By maximizing the objective function z (S M ) Obtaining a switching time sequence;
and calculating according to the switching time sequence to obtain the actual frequency spectrum amplitude of the transmitting current.
The calculating according to the switching time sequence to obtain the actual frequency spectrum amplitude of the emission current includes: calculated by the following formula:
wherein L is 0 And R is 0 Equivalent inductance and equivalent resistance of the transmitting coil, J n The current amplitude and phase being the n-order harmonics,
n represents harmonic order, ω n Is the angular frequency of the nth harmonic, T 0 For the period, U (t) is the excitation voltage of the electromagnetic probe. />
The main frequency current amplitude is regulated by arranging a broadband resonance compensator on the transmitting circuit, wherein the broadband resonance compensator is connected in series with the transmitting coil and comprises an inductance L connected in parallel 1 And capacitor C 1 。
The impedance of a main frequency point in a main frequency band is adjusted through the broadband resonance compensator, and specifically, the total impedance of the broadband resonance compensator and the transmitting coil is expressed as:
where// is the parallel operator, ω is the angular frequency of the emission current, L 0 Is the inductance of the transmitting coil, R 0 Is the internal equivalent resistance of the transmitting coil and the bridge circuit;
minimizing the total impedance so that the imaginary part j is 0 and the main frequency band range of the depth focusing waveform is f L ～f H Set the resonance center frequency f r The center frequency f of the resonance band is the center of the main frequency band according to the relationship omega=2pi f of the angular frequency and the frequency r Represented as
The parameters for obtaining the following calculated wideband resonance compensator are:
by adjusting inductance L 1 And capacitor C 1 And (3) adjusting the impedance of the dominant frequency point in the dominant frequency band.
In frequency domain electromagnetic exploration, an H-bridge inverter is used as a transmitting bridge circuit to ensure high transmitting power. The transmit voltage waveform generally employs a generalized bipolar square wave. At this time, the voltage always works at the peak value, so that the high voltage utilization rate of the direct current power supply and lower waveform generation difficulty can be ensured. The exciting voltage waveform of electromagnetic detection is periodic, and the period is T 0 Can be expressed as:
U(t)＝U(t+T 0 ) (1)
assuming that the voltage waveform switches polarity M times in one period, which is an even number, the switching time sequence can be expressed as:
S M ＝{t 0 ,t 1 ,...,t m ,...,t M } (2)
in this time sequence, t m The moment is the switching moment of voltage polarity, t 0 ＝0，t M ＝T 0 。
Assuming the voltage of the dc power supply is E, then in a single cycle, the transmit voltage waveform may be represented as written:
U(t)＝(-1) m-1 E t m-1 ＜t≤t m (3)
wherein the voltage waveform is selected from t 0 To t 1 Is in positive values in the first interval of (c).
For ease of analysis, the waveform is represented by a sequence of polarity switching instants of bipolar square waves. Then the design of the desired waveform is translated into a design of the switching time sequence. In a single period, fourier series expansion is performed on the formula (3):
the fourier coefficients are respectively:
wherein n represents harmonic order, ω n The angular frequency, which is the nth harmonic, is given by:
in connection with equation (3), equation (5) can be further deduced to be:
the voltage amplitude of each subharmonic can be expressed as:
in magnetic source electromagnetic detection, the influence of coil inductance on the emission current is not negligible. Therefore, the influence of the transmitting coil needs to be included in equation (8). Then the actual spectral amplitude of the emitted current is:
wherein L is 0 And R is 0 Equivalent inductance and equivalent resistance of the transmitting coil, J n Current amplitude and phase for the n-th harmonic.
In combination with equations (7) and (9), the spectral amplitude of the transmit waveform can be determined by a switching time sequence S that characterizes the transmit waveform M Calculated, i.e. the current spectrum is obtained by a sequence of switching moments S M As a function of the argument. The goal of the spectral design is to maximize the energy at the desired frequency with equal current magnitudes. In a deep focusing emission scheme based on the SHEPWM method, a frequency spectrum of an actual current and an expected frequency spectrum are establishedAnd solving a switching time sequence when the matching degree is highest according to the relation equation set. However, the maximum spectral amplitude value of each probe frequency is often difficult to estimate accurately, so that a margin needs to be given to the maximum value of each subharmonic when designing the desired spectrum. Otherwise, once the desired spectrum setting is not reasonable, the iterative solution process may fall into misconvergence. In order to increase the current amplitude of the main frequency as much as possible, the sum of the energy of the main frequency is added as a correction term to the solving target. The nonlinear equation system solving problem is converted into a nonlinear programming problem, and the solving aim is to improve the energy of the main frequency as much as possible while matching the expected frequency spectrum.
Then, assume that the estimated desired spectrum is I n The primary frequency of detection is { N 1 ,N 2 ,...,Ni,...,N L Definitions f 1 (S M ) And f 2 (S M ) The method comprises the following steps of:
the set objective function is as follows:
z(S M )＝w 1 ·f 1 (S M )+w 2 ·f 2 (S M ) (12)
wherein w is 1 ,w 2 Respectively weight coefficients.
The constraint condition is that
t 0 ＜t 1 ＜....＜t M (13)
Equation (10) characterizes the degree of matching of the actual current spectrum to the desired spectrum, while equation (11) characterizes the total energy of the detected dominant frequency. By maximizing the objective function z (S M ) The main frequency amplitude can be corrected while the expected frequency spectrum is matched, so that the main frequency energy is improved as much as possible, and the field intensity of the underground is further improved. Then, the present embodiment can easily solve the problem by using a conventional penalty function algorithm, which is not described herein.
During detection, according to the depth range of the investigation target area, an effective frequency band required by the exploration target area is determined by using a skin depth formula. The number and distribution of the main frequencies required can be determined by combining the requirement of the investigation target area on the longitudinal detection resolution. When the depth focusing waveform is designed based on the SHEPWM method, since the nonlinear equation sets are large in number and are difficult to solve iteratively, only the control equation set of the first eighth harmonic is established, and the amplitude of the higher harmonic is uncontrollable. Since the tenth and eleventh harmonics are close to the spectral magnitudes of the seventh, eighth and ninth harmonics being controlled, they can be utilized. In fact, once the main frequency harmonic needed by the detection target is changed by transformation, the amplitude of the high frequency harmonic is difficult to predict, and even serious interference is brought to parameter extraction, so that the detection precision is reduced. For comparison with the conventional SHEPWM waveform, the seventh to eleventh harmonics are set as the dominant frequency. The switching time sequence of the obtained five-frequency depth focusing waveform is shown in table I with the formula (12) as a solution target. The spectral pairs of the proposed depth focus waveform and the depth focus waveform based on the SHEPWM method are shown in Table II, with a fundamental emission current frequency of 128Hz.
Compared with the SHEPWM scheme, the provided depth focusing waveform has better spectral characteristics. According to Table II, the dominant frequency amplitudes are approximately equal, with the amplitudes exceeding 0.48A. The spectrum energy is distributed more reasonably, and the equal-precision measurement is easier to realize. Meanwhile, the useless harmonic amplitude is low, and the interference of data processing and parameter extraction is reduced. But the low frequency harmonic amplitude is still slightly higher and needs to be further suppressed.
Switching time sequence of five-frequency depth focusing waveform designed in table I
Spectral contrast of the depth focus waveform set forth in Table II and the SHEPWM method-based depth focus waveform
The present embodiment further adjusts the amplitude of the main frequency current by providing a wideband resonance compensator on the transmit circuit.
The amplitude of the dominant frequency of the deep focus waveform is suppressed by the inductive impedance, affected by the emitted magnetic dipole. In order to reduce the impedance of the transmitting coil to a plurality of consecutive main frequency bands and ensure that sufficient energy is transferred into the ground, this embodiment proposes a broadband resonance compensator as shown in fig. 1 (a), and an impedance characteristic as shown in fig. 1 (b). The resonance compensator consists of an inductance L 1 And capacitor C 1 Composition is prepared. L (L) 0 Is the inductance of the transmitting coil, R 0 Is the internal equivalent resistance of the transmitting coil and bridge. The front end is fed with a depth focus waveform by the transmitter.
The total impedance of the coil and the compensator is expressed by:
where// is the parallel operator and ω is the angular frequency of the emission current. In order to minimize equation (14), it is necessary to operate the circuit at resonance, i.e., to have the imaginary part j be 0. At this time, the system impedance is the lowest, and is approximately the internal equivalent resistance R 0 。
Assume that the main band range of the depth focus waveform is f L ～f H . Set the resonance center frequency f r Is the center of the primary frequency band. Based on the relationship ω=2pi f between angular frequency and frequency, the resonance band center frequency f is combined with (14) r Can be expressed as
After determining the radius and turns of the transmitting coil according to engineering requirements, L 0 Is determined. Winding the transmitting coil L 0 After that, it is difficult to fine-tune the inductance value by winding. Therefore, the wideband resonant compensator is designed mainly for the parameter L 1 And C 1 Which need to follow the following relationship:
by adjusting the inductance L according to the formulas (14) and (16) 1 And capacitor C 1 The impedance of the main frequency point in the main frequency band can be adjusted, i.e. the adjustment of the passband of the resonant circuit can be achieved, as shown in fig. 1 (b). L (L) 1 Affecting the bandwidth of the resonant matching circuit. The larger L 1 A wider passband is brought, and a plurality of main frequency current amplitudes in the effective band range are improved. Reasonable selection of compensator parameter inductance L 1 And capacitor C 1 When the imaginary part of the impedance in the effective frequency band is close to zero, a rise of the current values of a plurality of main frequency points in succession of the deep focus waveform is achieved, i.e. resonance in a wide frequency band is provided. Thereby reducing reactive power losses. Compared with an LC series resonant circuit, the compensator has higher control flexibility for the bandwidth of a passband. When the passband is set to be narrower, the low-frequency harmonic wave can be better selected and filtered to match with the depth focusing waveform with narrower effective frequency band, so that the spectral characteristic of the depth focusing waveform is further improved. Thus, the proposed resonance compensator has an effect similar to a bandpass filter, enabling selection of the dominant frequency in the passband range.
The invention also provides a towed frequency domain electromagnetic detection depth focusing emission system, which comprises: a transmit waveform design module:
determining an effective frequency band required by the exploration target area by utilizing a skin depth formula according to the depth range of the exploration target area;
combining the requirement of the investigation target area on the longitudinal detection resolution, and determining the number and distribution of the required main frequencies;
the set objective function is as follows:
z(S M )＝w 1 ·f 1 (S M )+w 2 ·f 2 (S M )
wherein w is 1 ,w 2 Respectively are weight coefficients, S M For the sequence of switching moments,
the estimated desired spectrum is I n ，J n For current amplitude and phase of the N-th harmonic, the detected dominant frequency is { N 1 ,N 2 ,...,N i ,...,N L }；
The constraint condition is that
t 0 ＜t 1 ＜....＜t M
By maximizing the objective function z (S M ) Obtaining a switching time sequence;
and calculating according to the switching time sequence to obtain the actual frequency spectrum amplitude of the transmitting current.
Calculating according to the switching time sequence to obtain the actual frequency spectrum amplitude of the emission current, including: calculated by the following formula:
wherein L is 0 And R is 0 Equivalent inductance and equivalent resistance of the transmitting coil, J n The current amplitude and phase being the n-order harmonics,
n represents harmonic order, ω n Is the angular frequency of the nth harmonic, T 0 For the period, U (t) is the excitation voltage of the electromagnetic probe.
The transmitting system also comprises a wideband resonance compensator arranged on the transmitting circuit for adjusting the amplitude of the main frequency current, wherein the wideband resonance compensator is connected with the transmitting coil in series and comprises an inductance L connected in parallel 1 And capacitor C 1 。
Simulation and analysis
The present embodiment was simulated using MATLAB/Simulink to verify the feasibility of the proposed wideband resonance compensator based transmission scheme. Fig. 2 shows the proposed transmitting system architecture, which mainly comprises a dc power supply, a control signal, an H-bridge transmitting bridge, a resonance compensator and a transmitting coil.
Under the conditions that the power supply voltage is 10V, the inductance of a transmitting coil is 200 mu H, the total internal impedance of the transmitting coil and an inverter bridge is 2 omega, and the fundamental frequency of transmitting current is 128Hz, comparison simulation is developed on a SHEPWM-based deep focusing transmitting scheme before and after adding a resonance compensator. Fig. 3 shows simulation results given by the shewm-based deep focus emission scheme. Fig. 3 (a) shows a current waveform. The bipolar square wave is distorted under the influence of the inductance, and the rising and falling edges of the current become obviously slow. As shown in fig. 3 (b), the frequency points are dense, and high-precision detection within a specific depth range can be realized. Only the first nine harmonics of the waveform are controlled. The seventh, eighth, and ninth harmonics were set to the detected dominant frequencies, with amplitudes of 2.05a,1.97a, and 1.92A, respectively. The amplitude of the main frequency current is suppressed and the amplitude decreases with increasing frequency, under the influence of the inductance of the transmitting coil. The tenth and eleventh harmonics were not controlled, with magnitudes of 1.96A and 2.58A, respectively. The tenth harmonic is close to the set dominant frequency amplitude and can be utilized. However, the eleventh harmonic has too high amplitude to be used for equal accuracy measurement. Meanwhile, because the high-frequency area is uncontrollable, serious high-frequency harmonic pollution exists.
Then, a matched broadband resonance compensator is designed. When the transmitting coil inductance L 0 After being determined, the inductance L 1 And capacitor C 1 The relationship of (2) is in accordance with equation (16). The resonance center point is selected to be 1088Hz in combination with the characteristics of the depth focus waveform. To conveniently select the optimal parameters, L is calculated 2 From 400 μH to 8000 μH, all possible capacitance C values satisfying equation (17) are solved in steps of 10 μH, the solution set being shown in FIG. 4. The inductance L can be seen 1 And capacitor C 1 Is approximately inversely proportional to the value of (c). The value of the inductance should not be too small, otherwise the capacitance value will be too large, which is not suitable from a volume and cost point of view. Meanwhile, if the inductance is too large, the weight and internal resistance thereof are increased, which may cause a high main frequency energy loss. Thus, the inductance L 1 The selection of (2) should be moderate, the range of 400. Mu.H-5000. Mu.H is reasonableA kind of electronic device.
Fig. 5 shows the variation of the main frequency impedance for different resonance compensator parameters. It can be seen that with L 1 The impedance of each dominant frequency decreases. L (L) 1 The variation of (c) has less effect on the eighth and ninth harmonics around the resonance frequency and greater effect on the magnitudes of the seventh, tenth and eleventh harmonics. In order to obtain a wider passband, ensure that each dominant frequency has a lower inductance, a larger L should be selected as much as possible within a reasonable range 1 . The influence on inductance is smaller after the inductance value is higher than 5mH, and the common specification of the device is considered, the inductance L 1 Taking 4mH and capacitance C 1 The value was taken as 120. Mu.F. At this time, the impedance values at the respective main frequencies were 2.081 Ω,2.008 Ω,2.007 Ω,2.052 Ω, and 2.13 Ω, respectively.
Fig. 6 shows the gain curve of the addition of the designed resonant matching circuit. It can be seen that the gains at 6, 7, 8, 9, 10 harmonics are respectively higher than 0.92 at each frequency within the effective frequency band. Because the designed resonant matching circuit is passive, it cannot produce additional gain. However, within the effective band, the output may have little attenuation. Indicating that the designed resonant matching circuit can realize resonance in a wide frequency band. At other frequencies, the output of the system may be significantly reduced due to the high frequency impedance of the load inductance. Therefore, the designed resonant matching circuit can realize lossless output at the designed target frequency and suppress other frequency components.
Fig. 7 shows simulation results after adding a broadband resonant matching circuit. In fig. 7 (a), the peak-to-peak value of the current time domain waveform is raised from 9.8 to 12.4A. According to the spectrum shown in fig. 7 (b), the primary frequency amplitude is respectively raised to 2.26A, 2.33A, 2.35A, 2.44A and 3.23A, and compared with the current spectrum in fig. 3 (b), the primary frequency current value is significantly raised, so that the detection accuracy can be effectively improved. The low-frequency harmonic amplitude is lower, so that the difficulty of data processing and parameter extraction can be reduced.
To continue, a five-frequency deep focus waveform was designed based on the optimization objective of equation (12), and a comparative simulation before and after the addition of the broadband resonance compensator was performed. For ease of comparison, the simulation conditions remain the same as the above simulation. Fig. 8 shows the proposed depth focus waveform with a fundamental frequency of 128Hz. The waveform is obviously distorted under the action of the inductance of the transmitting coil. The dominant frequency amplitudes were 2.18A,2.04a,1.99a,1.92a and 1.88A, respectively. Compared to fig. 3, the spectral energy is more reasonably distributed, with the dominant frequency amplitude being closer. The twelfth and tenth harmonics have low amplitudes, but the harmonic amplitudes at low frequencies are higher. Basically, the feasibility of the proposed waveform generation method is demonstrated. Meanwhile, under the effect of coil inductance inhibition, the frequency spectrum amplitude is reduced along with the increase of frequency.
Fig. 9 shows simulation results after adding the resonance compensator designed by the foregoing simulation. It can be seen that the current peak to peak value increases to 12.6A with dominant frequency amplitudes of 2.37A,2.40a,2.44a,2.42a and 2.37A, respectively. Compared with fig. 8, the spectrum average amplitude is obviously improved, because the resonance brought by the compensator counteracts most of the coil inductance. At this time, the detection energy is stronger, and the vertical resolution of the detection target can be effectively improved. The current amplitude is close, and the equal-precision measurement can be approximately realized. In addition, the resonance compensator has a good suppression effect on low-frequency harmonic waves, and can effectively reduce interference brought by useless harmonic waves to parameter extraction.
Next, a comparison simulation of the proposed two-frequency deep focus emission scheme before and after adding the resonance compensator is made. The simulation results are shown in fig. 10 under the conditions that the power supply voltage is 5V, the inductance of the transmitting coil is 200 muh, the total internal impedance of the transmitting coil and the inverter bridge is 1 Ω, and the fundamental frequency is 128Hz. The current magnitudes at the seventh and eighth times were 2.21A and 2.05A, respectively. Under the suppression effect of the inductance, the frequency spectrum amplitude of the main eighth time is obviously lower than that of the seventh time, and the equal-precision measurement cannot be realized. And there is a relatively significant low frequency harmonic at the fifth harmonic, with an amplitude of about 0.02A.
In order to cancel the inductive reactance brought about by the transmitting coil, the parameters of the resonance compensator are designed based on equation (12). Taking a series of inductances L 1 The current gain curve at the value of (2) and the simulation result are shown in fig. 11. It can be seen that the inductance value L 1 The lower the passband is, the narrower the selectivity of the resonance compensator to the dominant frequency is, and the easier it is to achieve suppression of unwanted harmonics outside the effective band. From the following componentsThe frequency band of the focusing waveform is narrower in two-frequency depth, and a lower inductance value L is needed to be selected when the parameters of the resonance compensator are determined 1 . But inductance value L 1 Cannot be too low otherwise significant suppression of the primary frequency amplitude is brought about.
To further compare the influence of different resonance compensator parameters on the spectrum of the emission current, the spectrum of the emission current in the two cases of the narrow-band resonance compensator and the wide-band resonance compensator is unfolded and compared, and the inductance L is simulated 1 And capacitor C 1 Values were (400. Mu.H, 206. Mu.F) and (5000. Mu.H, 143. Mu.F), and simulation results are shown in FIG. 12. Fig. 12 (a) shows the emission current spectrum when a broadband resonance compensator is added, with main frequency magnitudes of 3.26A and 3.27A, respectively. Compared with the method without adding the resonance compensator, the main frequency amplitude is obviously improved, and higher detection precision is realized. And the current amplitude values of the seventh time and the eighth time are approximately equal, so that the equal-precision measurement can be basically realized. However, the current amplitude at the fifth harmonic is still high, with a magnitude of 0.07A, due to the broader passband. According to fig. 12 (b), when the narrow-band resonance compensator is added, the main frequency amplitude of the seventh and eighth times is 3.20A and 3.24A respectively, and the fifth harmonic amplitude is reduced to 0.02A, which helps to reduce the interference of the useless harmonic on the parameter extraction. Therefore, the resonant compensator has the advantage of adjustable passband and higher flexibility. The passband width can be adjusted according to the detection requirement, so that the frequency selection of the main detection frequency is realized, and undesired harmonics are suppressed.
The foregoing description of the preferred embodiments of the invention is not intended to be limiting, but rather is intended to cover all modifications, equivalents, and alternatives falling within the spirit and principles of the invention.
Claims (8)
1. A towed frequency domain electromagnetic detection depth focusing emission method is characterized by comprising the following steps:
determining an effective frequency band required by the exploration target area by utilizing a skin depth formula according to the depth range of the exploration target area;
combining the requirement of the investigation target area on the longitudinal detection resolution, and determining the number and distribution of the required main frequencies;
the set objective function is as follows:
z(S M )＝w 1 ·f 1 (S M )+w 2 ·f 2 (S M )
wherein w is 1 ,w 2 Respectively are weight coefficients, S M For the sequence of switching moments,
the estimated desired spectrum is I n ，J n For current amplitude and phase of the N-th harmonic, the detected dominant frequency is { N 1 ,N 2 ,...,N i ,...,N L }；
The constraint condition is that
t 0 ＜t 1 ＜....＜t M
By maximizing the objective function z (S M ) Obtaining a switching time sequence;
and calculating according to the switching time sequence to obtain the actual frequency spectrum amplitude of the transmitting current.
2. The towed frequency domain electromagnetic probe depth focusing transmission method of claim 2, wherein said calculating a spectral magnitude of an actual transmission current based on a sequence of switching moments comprises: calculated by the following formula:
wherein L is 0 And R is 0 Equivalent inductance and equivalent resistance of the transmitting coil, J n The current amplitude and phase being the n-order harmonics,
n represents a harmonicThe number of waves omega n Is the angular frequency of the nth harmonic, T 0 For the period, U (t) is the excitation voltage of the electromagnetic probe.
3. The towed frequency domain electromagnetic probe depth focusing transmission method of claim 1, further comprising adjusting a main frequency current amplitude by providing a wideband resonance compensator on a transmission circuit, said wideband resonance compensator being connected in series with a transmission coil and comprising an inductance L connected in parallel 1 And capacitor C 1 。
4. A towed frequency domain electromagnetic probe depth focusing transmission method according to claim 3, wherein the impedance of a main frequency point in a main frequency band is adjusted by a wideband resonance compensator, and specifically, the total impedance of the wideband resonance compensator and a transmitting coil is expressed as:
where// is the parallel operator, ω is the angular frequency of the emission current, L 0 Is the inductance of the transmitting coil, R 0 Is the internal equivalent resistance of the transmitting coil and the bridge circuit;
minimizing the total impedance so that the imaginary part j is 0 and the main frequency band range of the depth focusing waveform is f L ～f H Set the resonance center frequency f r The center frequency f of the resonance band is the center of the main frequency band according to the relationship omega=2pi f of the angular frequency and the frequency r Represented as
The parameters for obtaining the following calculated wideband resonance compensator are:
by adjusting inductance L 1 And capacitor C 1 And (3) adjusting the impedance of the dominant frequency point in the dominant frequency band.
5. A towed frequency domain electromagnetic probe depth focusing transmission system, comprising: a transmit waveform design module:
determining an effective frequency band required by the exploration target area by utilizing a skin depth formula according to the depth range of the exploration target area;
combining the requirement of the investigation target area on the longitudinal detection resolution, and determining the number and distribution of the required main frequencies;
the set objective function is as follows:
z(S M )＝w 1 ·f 1 (S M )+w 2 ·f 2 (S M )
wherein w is 1 ,w 2 Respectively are weight coefficients, S M For the sequence of switching moments,
the estimated desired spectrum is I n ，J n For current amplitude and phase of the N-th harmonic, the detected dominant frequency is { N 1 ,N 2 ,...,N i ,...,N L }；
The constraint condition is that
t 0 ＜t 1 ＜....＜t M
By maximizing the objective function z (S M ) Obtaining a switching time sequence;
and calculating according to the switching time sequence to obtain the actual frequency spectrum amplitude of the transmitting current.
6. A towed frequency domain electromagnetic probe depth focusing transmission system according to claim 5, wherein said calculating a spectral magnitude of an actual transmitted current based on a sequence of switching moments includes: calculated by the following formula:
wherein L is 0 And R is 0 Equivalent inductance and equivalent resistance of the transmitting coil, J n The current amplitude and phase being the n-order harmonics,
n represents harmonic order, ω n Is the angular frequency of the nth harmonic, T 0 For the period, U (t) is the excitation voltage of the electromagnetic probe.
7. The towed frequency domain electromagnetic probe depth focusing transmission system of claim 5, further comprising a wideband resonant compensator disposed on said transmission circuit for adjusting said main frequency current amplitude, said wideband resonant compensator being connected in series with said transmission coil and comprising an inductance L in parallel 1 And capacitor C 1 。
8. A towed frequency domain electromagnetic probe depth focusing transmission system according to claim 7, wherein an impedance of a main frequency point within a main frequency band is adjusted by said wideband resonance compensator, and in particular, a total impedance of the wideband resonance compensator and a transmitting coil is expressed as:
where// is the parallel operator, ω is the angular frequency of the emission current, L 0 Is the inductance of the transmitting coil, R 0 Is the internal equivalent resistance of the transmitting coil and the bridge circuit;
minimizing the total impedance so that the imaginary part j is 0 and the main frequency band range of the depth focusing waveform is f L ～f H Set the resonance center frequency f r Is the center of the main frequency bandBased on the relationship ω=2pi f between angular frequency and frequency, the center frequency f of the resonance band r Represented as
The parameters for obtaining the following calculated wideband resonance compensator are:
by adjusting inductance L 1 And capacitor C 1 And (3) adjusting the impedance of the dominant frequency point in the dominant frequency band.
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Cited By (3)
| Publication number | Priority date | Publication date | Assignee | Title |
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| CN117169880A (en) * | 2023-11-03 | 2023-12-05 | 深圳市吉奥地球科技有限公司 | Pseudo-random signal transmitting method, receiving method and system |
| CN117492099A (en) * | 2024-01-02 | 2024-02-02 | 吉林大学 | Urban underground space towed time-frequency combined electromagnetic detection system and method |
| CN117691561A (en) * | 2024-01-31 | 2024-03-12 | 华中科技大学 | Secondary equipment cooperative protection method for resonance overvoltage |
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Cited By (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN117169880A (en) * | 2023-11-03 | 2023-12-05 | 深圳市吉奥地球科技有限公司 | Pseudo-random signal transmitting method, receiving method and system |
| CN117169880B (en) * | 2023-11-03 | 2024-02-09 | 深圳市吉奥地球科技有限公司 | Pseudo-random signal transmitting method, receiving method and system |
| CN117492099A (en) * | 2024-01-02 | 2024-02-02 | 吉林大学 | Urban underground space towed time-frequency combined electromagnetic detection system and method |
| CN117492099B (en) * | 2024-01-02 | 2024-04-19 | 吉林大学 | Urban underground space towed time-frequency combined electromagnetic detection system and method |
| CN117691561A (en) * | 2024-01-31 | 2024-03-12 | 华中科技大学 | Secondary equipment cooperative protection method for resonance overvoltage |
| CN117691561B (en) * | 2024-01-31 | 2024-04-26 | 华中科技大学 | Secondary equipment cooperative protection method for resonance overvoltage |
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