CN116191912A - Current control method for energy storage converter - Google Patents
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/4815—Resonant converters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J3/00—Circuit arrangements for ac mains or ac distribution networks
- H02J3/01—Arrangements for reducing harmonics or ripples
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J3/00—Circuit arrangements for ac mains or ac distribution networks
- H02J3/28—Arrangements for balancing of the load in a network by storage of energy
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J3/00—Circuit arrangements for ac mains or ac distribution networks
- H02J3/28—Arrangements for balancing of the load in a network by storage of energy
- H02J3/32—Arrangements for balancing of the load in a network by storage of energy using batteries with converting means
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J3/00—Circuit arrangements for ac mains or ac distribution networks
- H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/12—Arrangements for reducing harmonics from ac input or output
- H02M1/126—Arrangements for reducing harmonics from ac input or output using passive filters
Abstract
The invention provides a current control method of an energy storage converter, which belongs to the technical field of energy storage conversion control and comprises the following steps: acquiring a first error signal of a filter inductance current and a reference current of an energy storage converter, and a second error signal of a grid-connected side current and a zero reference current; the first error signal is processed by the direct current controller and the fundamental wave resonance controller respectively, and the second error signal is processed by the harmonic wave resonance controller and added to obtain a controller output signal; the filter capacitor branch is connected in parallel with an actively damped feedback signal, and the resonance voltage in the filter capacitor voltage is extracted by using an adaptive filter based on a second-order generalized integrator; the controller output signal is added with the sampled feedforward capacitor voltage signal and the resonance voltage in the filter capacitor voltage is subtracted to generate a modulation voltage. The invention improves the static characteristic and the dynamic response speed of the energy storage converter, improves the adaptability of the grid-connected environment, and enhances the flexibility of active damping adjustment.
Description
Technical Field
The invention belongs to the technical field of energy storage variable current control, and particularly relates to a current control method of an energy storage converter.
Background
In the current emerging electrochemical energy storage system, the energy storage converter is an energy router in the whole energy storage system, for example, a patent document with the publication number of CN114696347B, performs energy interaction between a battery and a power grid according to a scheduling instruction of a superior and the battery, can realize that the output of the energy storage system is not interfered when the power grid normally operates, and can accelerate the island judgment of the system when the main power grid loses power. The energy storage converter determines the stability of charging and discharging of the battery, also determines the grid-connected performance of the whole energy storage system, and is a key part of the energy storage system. The control of the energy storage converter is divided into an inner loop and an outer loop, the outer loop controls transmission power, and the inner loop controls grid-connected current. The current control characteristic of the energy storage converter determines the grid-connected performance of the energy storage converter, including the system power factor, the current harmonic content, the direct current bias, the dynamic response speed of the system and the like.
At present, the energy storage converter using the LC or LCL grid-connected filter is widely applied to various scenes, so that the grid-connected environment is large in difference, current control needs to have certain robustness, and resonance of the energy storage converter and a power grid can be restrained. Therefore, how to give a current control method meeting the above requirements is a key issue in the control of energy storage converters.
Disclosure of Invention
In order to solve the above-mentioned shortcomings of the prior art, the present invention provides a current control method for an energy storage converter, so as to solve the above-mentioned technical problems.
The invention provides a current control method of an energy storage converter, which comprises the following steps:
acquiring a first error signal of a filter inductance current and a reference current of an energy storage converter, and a second error signal of a grid-connected side current and a zero reference current;
the first error signal is processed by the direct current controller and the fundamental wave resonance controller respectively, and the second error signal is processed by the harmonic wave resonance controller and added to obtain a controller output signal;
the filter capacitor branch is connected in parallel with an actively damped feedback signal, and the resonance voltage in the filter capacitor voltage is extracted by using an adaptive filter based on a second-order generalized integrator;
and the output signal of the controller is added with the sampling feedforward capacitance voltage signal, and the resonance voltage in the filter capacitance voltage is subtracted to generate a modulation voltage, so that the control of the energy storage converter is realized.
Further, the transfer function of the controlled object of the energy storage converter is as follows:
the corresponding transfer function of the filter inductance is:
，L s is the inductance value of the filter inductor, R s Is L s Equivalent resistance on the resistor;
the filter capacitance corresponds to a transfer function of:
，C s is the capacitance value of the filter capacitor;
the corresponding transfer function of the grid-connected side inductance is as follows:
，L e is the total inductance value of the filter inductance at the grid-connected side, the grid-connected line inductance and the secondary side leakage inductance of the transformer, R e Is L e Equivalent resistance on the resistor;
the equivalent transfer function corresponding to the modulation is:
，T s is the control period of the converter. />
Further, the method further comprises the following steps: designing a current control loop of a filter inductor of the converter:
the nonlinear link, the filter capacitor voltage sampling feedforward and the active damping branch in the modulation process are taken as disturbance d in a current control loop of a filter inductor of the converter;
the controlled object of the current control loop of the filter inductor of the converter is the filter inductor, and the transfer function is 1/Z Ls (s)；
The first error signal is processed by a direct current controller and a fundamental resonance controller respectively and then added with disturbance d to generate a modulation voltage;
the direct current controller is a proportional integral controller, namely a PI controller, and the transfer function of the direct current controller is as follows:
wherein ,Rs Filtering inductance L for current transformer s Equivalent resistance, K p For the gain of the PI controller, the zero point of the PI controller is designed as s= -R s /L s Compensating a pole of a controlled object corresponding to the filter inductance transfer function;
the fundamental wave resonance controller is a vector proportional integral resonance controller, namely a VPI resonance controller, and the transfer function of the vector proportional integral resonance controller is as follows:
in the formula ,Rs Filtering inductance L for current transformer s Equivalent resistance, K v Gain, ω of the controller for vector proportional integral resonance e For the grid voltage angular frequency, the VPI resonant controller has a s= -R on the molecule s /L s Can be used for compensating the pole of the controlled object corresponding to the filter inductance transfer function.
Further, the fundamental resonance controller considers modulation delay effects, i.e. modulation equivalent transfer function G PWM (s) the lower the control frequency, the greater the impact on the system dynamic response, with phase delay; therefore, when the control frequency is 15 times less than the fundamental frequency, the vector proportional product with delay compensation is selectedA split resonance controller having a transfer function of:
in the formula φm Is the fundamental wave phase compensation angle, R s Filtering inductance L for current transformer s Equivalent resistance, omega e For the angular frequency of the voltage of the power grid, K vc Is the gain of the VPIc controller.
Further, the designed converter filter inductor current control loop is simplified:
after the controller in the filter inductor current control loop is determined, the open loop transfer function of the filter inductor current control loop can be obtained as follows:
； in the formula Ci (s) = C i0 (s) + C i1 (s)；
The closed loop transfer function of the current control loop of the filter inductor is as follows:
further, the method further comprises the step of designing a grid-connected current control loop:
defining a grid-connected current control loop controlled object as G L (s) the transfer function is:
the transfer function of the filter capacitor voltage feed-forward branch H(s) is:
wherein ,GSOGI (s) is a filter capacitor voltage sampling filter transfer function:
in the formula ks For SOGI coefficient, ω e Is the grid voltage angular frequency; the second error signal is processed by a harmonic resonance controller to obtain a harmonic control signal, and each subharmonic in the actual grid-connected current is a sine signal, so that n resonance R controllers are selected for control according to different subharmonics, and the transfer functions of the n resonance R controllers are as follows:
ω e for the angular frequency of the voltage of the power grid, K vc For the gain of the VPIc controller,
gain for n resonant R controllers.
Further, the method further comprises the following steps:
active damping is added into the control system, the active damping is derived from parallel passive damping on a filter capacitor and is realized by using voltage sampling of the filter capacitor, and the differential link and 1/G of the full frequency band in the actual system are considered PWM The advanced link of(s) cannot be realized, the resonance voltage in the filter capacitor is obtained by using the adaptive filter based on the SOGI, and only the resonance voltage is subjected to differential processing, so that the transfer function of the active damping branch is obtained as follows: g SOGIrf (s)Z Ls (s)/R d ；
wherein ,
；
in the formula ,ωr For the system resonant angular frequency, k sr Is SOGI coefficient, phi r Is G PWM (s) at ω r Is used for the phase delay introduced in the process.
Further, the method is characterized by further comprising: feedback branches using multiple SOGI-based adaptive filters, each feedback branch G SOGIrx (s)Z Ls (s)/R dx (x=1, 2, … M) each having an independent SOGI-based adaptive filter and an independent active damping R dx, wherein 
； in the formula ksr For SOGI coefficient, ω rx Is the central resonance angular frequency and is regulated by a frequency locking ring, phi rx Is G PWM (s) at ω rx Phase delay introduced at that point.
Further, the method further comprises the following steps: the feedback transfer function of the whole current control method after active damping is considered is as follows:
。
the control method for the energy storage converter has the beneficial effects that the fundamental wave and direct current of the filter inductance current and grid-connected side harmonic waves are controlled, the static characteristic and the dynamic response speed of the energy storage converter are improved, and the adaptability to the grid-connected environment is improved. The resonance voltage in the filter capacitor voltage is tracked by using a plurality of self-adaptive filters, so that full-frequency multi-section adjustable resistive damping is realized, the problem that the application scene of the energy storage converter is not fixed and the dynamic response of the traditional active damping realization method is slow is solved, and the flexibility of active damping adjustment is enhanced.
Drawings
In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the drawings that are required to be used in the description of the embodiments or the prior art will be briefly described below, and it will be obvious to those skilled in the art that other drawings can be obtained from these drawings without inventive effort.
FIG. 1 is a schematic diagram of a grid-connected universal energy storage converter in the present invention;
FIG. 2 is a control block diagram of the present invention constructed using a current control method;
FIG. 3 is a block diagram of a filter inductor current control according to one embodiment of the present invention;
FIG. 4 is a grid-tie current control block diagram of one embodiment of the present invention;
FIG. 5 is a control block diagram of a passive damping implementation of one embodiment of the present invention;
FIG. 6 is a control block diagram of an active damping single-branch implementation of one embodiment of the present invention;
FIG. 7 is a control block diagram of an active damping single-arm implementation of another embodiment of the present invention;
FIG. 8 is a control block diagram of an active damping multi-branch implementation of one embodiment of the present invention;
fig. 9 is a schematic diagram of a plurality of adaptive filter frequency bins according to one embodiment of the invention.
Detailed Description
In order to make the technical solution of the present invention better understood by those skilled in the art, the technical solution of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention, and it is apparent that the described embodiments are only some embodiments of the present invention, not all embodiments. All other embodiments, which can be made by those skilled in the art based on the embodiments of the present invention without making any inventive effort, shall fall within the scope of the present invention.
The present invention will now be described with reference to the following keywords.
VPI resonant controller: a vector proportional integral resonance controller, a linear controller, forms control deviation according to the given value and the actual output value, controls the sine signal, and realizes the non-static-difference control of the sine signal.
PI controller: the proportional-integral controller is a linear controller, which forms control deviation according to a given value and an actual output value, forms control quantity by linearly combining the proportional and integral of the deviation, and controls a controlled object to realize the static-difference-free control of a direct current signal.
SOGI adaptive filter: and an adaptive filter based on a second-order generalized integrator is used for realizing the dead-difference-free filtering and tracking of the sinusoidal signals.
The invention provides a current control method of an energy storage converter, and aims at the energy storage converter using an LC or LCL type grid-connected filter shown in figure 1. Wherein, the filter inductance near the energy storage converter is defined as L s ，L s R for equivalent resistance s Representation, three phases L s The current on is defined as i s . The filter capacitance is defined as C s (Star connection) three-phase filter capacitor voltage is defined as u c . The filter inductance of the grid-connected side of the energy storage converter, the grid-connected line inductance and the secondary side leakage inductance of the transformer are defined as L together e ，L e R for equivalent resistance e Representation, three phases L e The current on is respectively defined as i e 。
The present invention proposes a current control method shown in the dashed box of fig. 2 for the energy storage converter system shown in fig. 1. Because zero-sequence fundamental components are not existed in the energy storage converter system, the control of fundamental current is carried out under a two-phase static coordinate system, so that the occurrence of rotation coordinate transformation is avoided, and the system is used as a linear time-invariant system for carrying out large-signal analysis.
As shown in fig. 2, the current control method includes:
obtaining filter inductance L of energy storage converter s Current i s And a reference current i sref First error signal of (1), grid-connected side current i e And a second error signal of zero reference current;
the first error signal is processed by the direct current controller and the fundamental wave resonance controller respectively, and the second error signal is processed by the harmonic wave resonance controller and added to obtain a controller output signal;
the filter capacitor branch is connected in parallel with an actively damped feedback signal, and the resonance voltage in the filter capacitor voltage is extracted by using an adaptive filter based on a second-order generalized integrator;
and the output signal of the controller is added with the sampling feedforward capacitance voltage signal, and the resonance voltage in the filter capacitance voltage is subtracted to generate a modulation voltage, so that the control of the energy storage converter is realized.
In the present embodiment, the reference current i sref The active component amplitude of the (a) is from an active outer ring, the reactive component amplitude is from a reactive outer ring, and the power grid voltage phase information is from a power grid synchronization strategy; the command value of the zero reference current, i.e., the harmonic current, is set to 0. Wherein the first error signal is subjected to three-phase to two-phase conversion before being processed by the fundamental wave controller, and the second error signal is subjected to harmonic controlThe three-phase-two-phase conversion is carried out before the processing of the controller, and the two-phase-three-phase conversion is carried out after the processing of the controller.
The transfer function of the controlled object of the energy storage converter is as follows:
the corresponding transfer function of the filter inductance is:
，L s is the inductance value of the filter inductor, R s Is L s Equivalent resistance on the resistor;
the filter capacitance corresponds to a transfer function of:
，C s is the capacitance value of the filter capacitor;
the corresponding transfer function of the grid-connected side inductance is as follows:
，L e is the total inductance value of the filter inductance at the grid-connected side, the grid-connected line inductance and the secondary side leakage inductance of the transformer, R e Is L e Equivalent resistance on the resistor;
the equivalent transfer function corresponding to the modulation is:
，T s is the control period of the converter.
The specific content of the current control method comprises the design of a current controller and the realization of active damping. After the controlled object is defined, the current controller and the active damping are designed.
The current control of the filter inductor of the converter is designed:
ignoring non-linear links in the modulation process, assuming u c Sampling feedforward can substantially cancel u in a circuit c The effect of this branch, together with the active damping branch, is considered as a disturbance in the control loop. The filter inductance control loop can be simplified to the control block diagram shown in fig. 3. Wherein C is i0 (s) is a DC controller, C i1 (s) is the fundamental controller and the disturbance is denoted by d.
The first error signal is processed by a direct current controller and a fundamental resonance controller respectively and then added with disturbance d to generate a modulation voltage;
the fundamental wave resonance controller is a vector proportional integral resonance controller, namely a VPI resonance controller, and the transfer function of the vector proportional integral resonance controller is as follows:
in the formula ,Kv Is the gain, omega of the VPI controller e Is the grid voltage angular frequency. The use of a VPI controller enables a dead-beat-free control of the fundamental current, and in addition, there is a s= -R on the VPI controller molecule s /L s The corresponding poles in the system can be compensated, so that the dynamic response speed of the system can be improved by using the VPI controller.
Alternatively, as an embodiment of the present application, in actual use, G PWM (s) phase delay exists, so that the dynamic performance is influenced, and therefore, when the control frequency is low, super-front phase compensation can be added into the VPI harmonic controller to form a VPic harmonic controller with delay compensation, and the transfer function of modulation is as follows:
in the formula φm Is the fundamental phase compensation angle. The VPIc controller adds a zero point in the s-domain right half plane, and the zero point also has an influence on the dynamic performance of the system. Therefore, the invention only uses the VPic controller when the control frequency is lower than 15 times the grid frequency.
The function of the dc controller is to suppress dc bias, so long as no static difference control is achieved, in this embodiment, the dc controller is a PI controller:
in the formula ,Rs Filtering inductance L for current transformer s Equivalent resistance, K p Is the gain of the PI controller. PI controlThe zero point of the device is also designed as s= -R s /L s And compensating the pole of the controlled object corresponding to the filter inductance transfer function. After the fundamental wave controller and the direct current controller are determined, the gain of the controller can be designed through a Bode diagram, a Nyquist diagram, a Nichols diagram, a root track and the like, and K is determined v and Kp 。
The grid-connected current controller is designed as follows:
the open loop transfer function of the filter inductor current control loop is:
in the formula Ci (s) = C i0 (s) + C i1 (s). The closed loop transfer function of the filter inductance current control loop is derived from the open loop transfer function as follows:
the schematic diagram shown in FIG. 2 is then further simplified to yield FIG. 4, G L (s) is a transfer function of all controlled objects, which is:
h(s) is the feedback loop transfer function:
G SOGI (s) is u c The transfer function of the sampling feedforward filter is filtered by using an adaptive filter based on SOGI, and the transfer function is as follows:
in the formula ks For SOGI coefficient, ω e Is the grid voltage angular frequency.
The current control method provided by the embodiment of the invention uses harmonicsWave resonance controller, C i,n (s) is a controller for suppressing each sub-harmonic in the grid-connected current. Because the system has no requirement on the dynamic response speed for restraining the harmonic waves, the controller design can be directly carried out in a closed-loop system. Each subharmonic is also a sine signal, the resonance controller is selected to realize no static difference control, and the transfer function of a single resonance R controller is as follows:
for different subharmonics, n resonant controllers need to be added. Gain K of resonant controller r,n The selection can be made by looking at the bode plot.
So far, the current controller in the whole system is designed. The controller aims at the energy storage converter using the LC or LCL grid-connected filter, improves the static characteristic and dynamic response speed of the energy storage converter, and improves the adaptability of the grid-connected environment. In order to ensure the dynamic response speed of the system, the system controls the fundamental wave of the filter inductor current. The controller selects a vector proportional integral resonance controller to realize the static-difference-free tracking of sinusoidal signals, and meanwhile, the influence of a system pole introduced by a filter inductor on the system performance is eliminated, and the dynamic response speed of the system is improved. In order to avoid direct current bias in each phase of current of the energy storage converter, a proportional-integral controller is used for restraining the direct current bias. In order to reduce the total harmonic distortion rate of the grid-connected current of the energy storage converter, a resonance controller is used for controlling low-order harmonics in the grid-connected current.
However, for an energy storage converter using an LC or LCL filter, the resonance problem during grid connection of the energy storage converter must also be solved. Besides the method of adding passive damping in the filter, the method of adding active damping to inhibit system resonance is a method without increasing system cost and loss, and has more advantages.
The active damping is to add an extra branch in the control loop, which is equivalent to adding a resistor in the filter capacitor branch to restrain the system oscillation. And thus can be back-pushed by passive damping when deriving. FIG. 5 shows the control of the shunt resistance of the capacitive branchA block diagram, wherein the dashed box is a passive damping R added outside the initial control block diagram d 。
The active damping is realized by simulating the passive damping, so that the active damping can be reversely deduced from the passive damping when the active damping is added, and the feedback for adjusting the damping is moved to the control side as shown in fig. 6. C due to control of DC and fundamental i (s) is substantially unresponsive to resonant frequency signals, and therefore the uppermost-C of FIG. 6 i (s)/Z R The(s) feedback branch may be directly ignored. And the other Z L (s)/R d G PWM (s) in the feedback branch, Z Ls (s) differentiation links equivalent to full frequency band, 1/G PWM And(s) is an advanced link, and both cannot be realized in a digital control system, namely active damping of pure resistance characteristics of a full frequency band cannot be realized.
Therefore, the embodiment of the invention uses the SOGI-based adaptive filter to track and filter the resonant frequency voltage, and the resonant frequency voltage is in Z L (s)/R d G PWM (s) adding a zero and two poles in the feedback branch, so that the branch can be realized; meanwhile, the SOGI is realized by using a digital realization method based on two integrators, three sinusoidal signals with the same phase as an input signal, advanced by 90 degrees and delayed by 90 degrees can be simultaneously output, a differential link at the resonance frequency can be realized, and the sampling delay, the calculation delay and the PWM delay can be subjected to phase compensation, so that pure resistance characteristic damping is injected at the system resonance point.
The transfer function of the filtering part of the SOGI-based adaptive filter is defined as G after considering the PWM phase influence SOGIrf (s), i.e
in the formula ωr Obtaining resonant angular frequency, k, for frequency-locked loop (FLL) sr The SOGI coefficient was chosen to be 1.414. Phi (phi) r Is G PWM (s) at ω r Phase delay introduced at that point. At this time, the feedback branch becomes G SOGIrf (s)Z Ls (s)/R d As shown in fig. 7.
For the followingThe converter with fixed application scene is fixed in grid-connected side inductance value, the resonant frequency is fixed, and the FLL of the adaptive filter can determine the central resonant angular frequency. In implementation, only G is needed SOGIrf (s)Z Ls (s)/R d An actively damped feedback branch is sufficient. However, the PCS has wide application scene, the grid-connected side characteristic is not fixed, the nonlinear characteristic of the frequency locking ring matched with the adaptive filter can be amplified, the dynamic response speed of the system is slow, and PCS fault protection can be possibly caused.
The present invention therefore proposes to solve this problem using a plurality of feedback branches of SOGI based adaptive filters, as shown in fig. 8. Each feedback branch G SOGIrx (s)Z Ls (s)/R dx (x=1, 2, … M) each having an independent SOGI-based adaptive filter and an independent active damping R dx The transfer function of the filtering part of the adaptive filter is that
in the formula ωrx For the center resonant angular frequency, the tuning is responsible for by the Frequency Locked Loop (FLL). k (k) sr The SOGI coefficient was chosen to be 1.414. Phi (phi) rx Is G PWM (s) at ω rx Phase delay introduced at that point.
The following is how to determine the center resonance angular frequency ω of each adaptive filter rx Further description will be given. A plurality of adaptive filters used for the embodiment of the invention, the input signal of each adaptive filter is composed of a sampling feedforward voltage u c Subtracting its fundamental voltage u c1 Obtained by equivalently stringing 1-G at the input side SOGI (s) a filter. The frequency bins for which each adaptive filter is responsible are shown in FIG. 9, where ω rLC For a resonant frequency where the system has only LC filters, i.e. grid-tie inductances of 0, in particular,
the method comprises the steps of carrying out a first treatment on the surface of the Assuming that M adaptive filters are used, the range of variation of the resonant angular frequency of each adaptive filter is defined as Δω= (ω) c −ω rLC )/M，ω c Is the nyquist frequency of the system. The upper and lower limits of the resonant angular frequency are respectively:
；/>
the method comprises the steps of carrying out a first treatment on the surface of the The center resonance angular frequency of the adaptive filter obtained by averaging is as follows: />
。
It should be noted that M may be selected according to the computing resources of the controller, and in particular, M is determined by the system control frequency and the bandwidth of the adaptive filter.
The embodiment of the invention provides the method for realizing the active damping, at least one of M adaptive filters tracks resonance to voltage, and the tracking speed is far faster than that of the adaptive filter because the output frequency deviation from FLL and the resonance frequency are delta omega at maximum. For other adaptive filters which do not play a role in tracking, as no voltage signal with corresponding frequency exists in the system, the filter does not output, and the control system is not influenced. Therefore, all branch outputs are added to the controller output. Compared with the traditional active damping realization method, the method increases the flexibility of active damping adjustment.
After adding active damping, only the feedback transfer function H(s) is affected in the overall current control of FIG. 4, the feedback transfer function H of active damping is considered Rd (s) is:
after the transfer function is determined, the gain and active damping resistance of the controller can be designed through a Bode diagram, a Nyquist diagram, a Nichols diagram, a root track and the like, and meanwhile, the stability of the system can be determined directly through a closed loop zero pole.
The method for realizing the active damping uses a plurality of self-adaptive filters to track the resonance voltage, realizes the full-frequency multi-section adjustable resistive damping, solves the problem that the application scene of the energy storage converter is not fixed and the dynamic response of the traditional method for realizing the active damping is slow, and enhances the flexibility of the adjustment of the active damping. Each adaptive filter filters and tracks the resonance voltage, and the adaptive frequency ranges of the adaptive filters are set to different upper and lower limits and are not overlapped with each other. The resonance voltage input of the self-adaptive filter is obtained by subtracting the fundamental wave voltage obtained by filtering from the filter capacitor voltage sample, so that the influence of active damping on fundamental wave control is avoided.
Although the present invention has been described in detail by way of preferred embodiments with reference to the accompanying drawings, the present invention is not limited thereto. Various equivalent modifications and substitutions may be made in the embodiments of the present invention by those skilled in the art without departing from the spirit and substance of the present invention, and it is intended that the present invention encompass all such modifications and substitutions as would be within the scope of the present invention as defined by the appended claims. Therefore, the protection scope of the invention is subject to the protection scope of the claims.
Claims (9)
1. The method for controlling the current of the energy storage converter is characterized by comprising the following steps of:
acquiring a first error signal of a filter inductance current and a reference current of an energy storage converter, and a second error signal of a grid-connected side current and a zero reference current;
the first error signal is processed by the direct current controller and the fundamental wave resonance controller respectively, and the second error signal is processed by the harmonic wave resonance controller and added to obtain a controller output signal;
the filter capacitor branch is connected in parallel with an actively damped feedback signal, and the resonance voltage in the filter capacitor voltage is extracted by using an adaptive filter based on a second-order generalized integrator;
and the output signal of the controller is added with the sampling feedforward capacitance voltage signal, and the resonance voltage in the filter capacitance voltage is subtracted to generate a modulation voltage, so that the control of the energy storage converter is realized.
2. The energy storage converter current control method of claim 1, wherein the energy storage converter controlled object transfer function is as follows:
the corresponding transfer function of the filter inductance is:
，L s is the inductance value of the filter inductor, R s Is L s Equivalent resistance on the resistor;
the filter capacitance corresponds to a transfer function of:
，C s is the capacitance value of the filter capacitor;
the corresponding transfer function of the grid-connected side inductance is as follows:
，L e is the total inductance value of the filter inductance at the grid-connected side, the grid-connected line inductance and the secondary side leakage inductance of the transformer, R e Is L e Equivalent resistance on the resistor;
the equivalent transfer function corresponding to the modulation is:
，T s is the control period of the converter.
3. The energy storage converter current control method of claim 2, further comprising: designing a current control loop of a filter inductor of the converter:
the nonlinear link, the filter capacitor voltage sampling feedforward and the active damping branch in the modulation process are taken as disturbance d in a current control loop of a filter inductor of the converter;
the controlled object of the current control loop of the filter inductor of the converter is the filter inductor, and the transfer function is 1/Z Ls (s)；
The first error signal is processed by a direct current controller and a fundamental resonance controller respectively and then added with disturbance d to generate a modulation voltage;
the direct current controller is a proportional integral controller, namely a PI controller, and the transfer function of the direct current controller is as follows:
wherein ,Rs Filtering inductance L for current transformer s Equivalent resistance, K p For the gain of the PI controller, the zero point of the PI controller is designed as s= -R s /L s Compensating a pole of a controlled object corresponding to the filter inductance transfer function;
the fundamental wave resonance controller is a vector proportional integral resonance controller, namely a VPI resonance controller, and the transfer function of the vector proportional integral resonance controller is as follows:
in the formula ,Rs Filtering inductance L for current transformer s Equivalent resistance, K v Gain, ω of the controller for vector proportional integral resonance e For the grid voltage angular frequency, the VPI resonant controller has a s= -R on the molecule s /L s Can be used for compensating the pole of the controlled object corresponding to the filter inductance transfer function.
4. A method of controlling an energy storage converter current according to claim 3, wherein the fundamental resonance controller considers modulation delay effects, namely a modulation equivalent transfer function G PWM (s) the lower the control frequency, the greater the impact on the system dynamic response, with phase delay; therefore, when the control frequency is 15 times less than the fundamental wave frequency, a vector proportional integral resonance controller with delay compensation is selected, and the transfer function is as follows:
in the formula φm Is the fundamental wave phase compensation angle, R s Is a variable flowFilter inductance L s Equivalent resistance, omega e For the angular frequency of the voltage of the power grid, K vc Is the gain of the VPIc controller.
5. A method according to claim 3, further comprising: simplifying a designed current control loop of the filter inductor of the converter:
after the controller in the filter inductor current control loop is determined, the open loop transfer function of the filter inductor current control loop can be obtained as follows:
； in the formula Ci (s) = C i0 (s) + C i1 (s)；
The closed loop transfer function of the current control loop of the filter inductor is as follows:
6. the method of claim 5, further comprising designing a grid-tie current control loop:
defining a grid-connected current control loop controlled object as G L (s) the transfer function is:
the transfer function of the filter capacitor voltage feed-forward branch H(s) is:
wherein ,GSOGI (s) is a filter capacitor voltage sampling filter transfer function:
in the formula ks For SOGI coefficient, ω e Is the grid voltage angular frequency; the second error signal is processed by a harmonic resonance controller to obtain a harmonic control signal, and each subharmonic in the actual grid-connected current is a sine signal, so that n resonance R controllers are selected for control according to different subharmonics, and the transfer functions of the n resonance R controllers are as follows:
ω e for the angular frequency of the voltage of the power grid, K vc For the gain of the VPIc controller,
gain for n resonant R controllers.
7. The method as recited in claim 2, further comprising:
active damping is added into the control system, the active damping is derived from parallel passive damping on a filter capacitor and is realized by using voltage sampling of the filter capacitor, and the differential link and 1/G of the full frequency band in the actual system are considered PWM The advanced link of(s) cannot be realized, the resonance voltage in the filter capacitor is obtained by using the adaptive filter based on the SOGI, and only the resonance voltage is subjected to differential processing, so that the transfer function of the active damping branch is obtained as follows: g SOGIrf (s)Z Ls (s)/R d ；
wherein ,
；
in the formula ,ωr For the system resonant angular frequency, k sr Is SOGI coefficient, phi r Is G PWM (s) at ω r Is used for the phase delay introduced in the process.
8. The method as recited in claim 7, further comprising: using multiple SOGI-basedFeedback branches of the adaptive filter of (a), each feedback branch G SOGIrx (s)Z Ls (s)/R dx (x=1, 2, … M) each having an independent SOGI-based adaptive filter and an independent active damping R dx, wherein 
； in the formula ksr For SOGI coefficient, ω rx Is the central resonance angular frequency and is regulated by a frequency locking ring, phi rx Is G PWM (s) at ω rx Phase delay introduced at that point.
9. The method according to claim 6 or 8, further comprising: the feedback transfer function of the whole current control method after active damping is considered is as follows:
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