CN116191576A - Network-structured three-phase converter and impedance and admittance model modeling method thereof - Google Patents

Network-structured three-phase converter and impedance and admittance model modeling method thereof Download PDF

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CN116191576A
CN116191576A CN202310340499.XA CN202310340499A CN116191576A CN 116191576 A CN116191576 A CN 116191576A CN 202310340499 A CN202310340499 A CN 202310340499A CN 116191576 A CN116191576 A CN 116191576A
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converter
current
axis
voltage
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李先允
郑雨萱
徐意
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Nanjing Institute of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H02J3/46—Controlling of the sharing of output between the generators, converters, or transformers
    • H02J3/48—Controlling the sharing of the in-phase component
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H02J3/46—Controlling of the sharing of output between the generators, converters, or transformers
    • H02J3/50—Controlling the sharing of the out-of-phase component
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J2203/00—Indexing scheme relating to details of circuit arrangements for AC mains or AC distribution networks
    • H02J2203/20—Simulating, e g planning, reliability check, modelling or computer assisted design [CAD]

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Abstract

The invention discloses a network three-phase converter and an impedance and admittance model modeling method thereof, which adopts a calculation method for constructing a dq model step by step, adopts modularized analysis, accumulates layer by layer, integrates calculation, and forms the dq impedance and admittance model of the network three-phase converter, wherein the method comprises the following steps: constructing a small signal dq model of a main circuit of the converter; constructing a small signal dq impedance model of a converter control link, and sequentially constructing a voltage-current variable coordinate transformation dq model, an active power reactive power dq model, a sampling low-pass filtering dq model, a power control loop phase angle dq model, a voltage reference signal dq model, a voltage control outer loop dq model, a current control inner loop dq model and a PWM control delay dq model; and simplifying calculation to form a dq impedance model and an admittance model expression of the network three-phase converter. The network-structured three-phase current transformer impedance and admittance model calculated by the invention can be expanded to impedance admittance models of a plurality of current transformers connected in parallel, and provides a model foundation for analyzing the stability of the system in the off-grid and grid-connected states of the current transformers.

Description

Network-structured three-phase converter and impedance and admittance model modeling method thereof
Technical Field
The invention relates to the technical field of new energy power systems, in particular to a network three-phase converter and an impedance and admittance model modeling method thereof.
Background
Renewable energy sources (mainly wind energy and solar energy) are developed and utilized rapidly, a large amount of new energy sources are connected in grid, and a converter is widely used as a key component for new energy source grid connection. The high-proportion new energy determines high-proportion power electronic equipment, so that the economical efficiency and the flexibility of the operation of the power system are greatly enhanced, but the capacity ratio of the synchronous generator in the power system is gradually reduced, the integral inertia of the power system is greatly reduced, and the stability of the power system is reduced.
Currently, existing converters for connecting renewable energy sources mostly use grid-connected control, and phase information of a grid-connected Point (PCC) is measured by a phase-locked loop (PLL) to achieve synchronization with a power grid. The grid-connected converters, although having a fast power response capability, cannot operate normally in islanding situations, and at the same time cannot usually establish frequency and voltage in weak grids with widely distributed renewable energy sources, have stability problems, which inhibit their grid support functions and interactions with other generator sets. Therefore, there is an urgent need for a grid-connected three-phase converter, instead of a grid-connected converter, which can operate under off-grid and on-grid conditions, so as to reduce the influence of the grid-connected converter on the system stability.
The stability analysis of the system relies on an accurate, simple, flexible mathematical model. The main methods at present are state space methods and impedance analysis methods. However, the state space method often integrates a system power supply and a load, builds an overall model, and has to reconstruct a new system model when the load topology changes, thereby greatly increasing the complexity of building and analyzing the model. Meanwhile, the constructed state equation is often difficult to analyze due to the fact that the order is too high. According to the impedance analysis method, a single converter is regarded as a voltage source or a current source, an impedance model of the whole system can be calculated only by calculating output impedance, and a simple, convenient and flexible modeling thought is provided for researching parallel operation of different converters and parallel operation of multiple converters of the same type. Therefore, on the basis of providing a network three-phase converter, the network three-phase converter impedance and admittance model modeling method is provided, and a model foundation is laid for researching system stability and parallel operation of multiple converters.
Disclosure of Invention
Aiming at the defects in the prior art, the invention provides a grid-structured three-phase converter and an impedance and admittance model modeling method thereof, which replace a grid-following converter, can operate under off-grid and grid-connected conditions, reduce the influence of the grid-following converter on system stability, and lay a model foundation for researching system stability and parallel operation of the multi-converter.
In order to achieve the above purpose, the present invention adopts the following technical scheme:
a network three-phase converter comprises a main circuit link and a control link;
the main circuit link comprises a three-phase six-arm inverter and an LC filter electrically connected with the three-phase six-arm inverter; the LC filter comprises an inductance L f And filter capacitor C f The inductance L f One end of the inductor L is connected with the output end of the three-phase six-arm inverter f The other end of (a) is connected with a filter capacitor C f And a bus bar, a capacitor C f The other end of the first electrode is grounded;
the control link comprises a first abc/dq coordinate transformation module, a second abc/dq coordinate transformation module, a third abc/dq coordinate transformation module, a fourth abc/dq coordinate transformation module, a power calculation module, a low-pass filter, a power control loop, a voltage control outer loop, a current control inner loop and a PWM control module;
the inductance L f Is of the inductance current i of (2) L The power calculation module receives a d-axis component voltage value and a q-axis component voltage value output by the abc/dq coordinate conversion module II and a d-axis component current value and a q-axis component current value output by the abc/dq coordinate conversion module III, and outputs instantaneous active power P e And instantaneous reactive power Q e The low-pass filter filters the instantaneous active power P output by the calculation module e And instantaneous reactive power Q e And outputting the average active power and the average reactive power to the power control loop; the power control loop receives active power parameters based on a virtual synchronization control strategyTest value P ref And reactive power reference value Q ref Outputting synchronous phase angles theta of a main circuit to a first abc/dq coordinate conversion module, a second abc/dq coordinate conversion module, a third abc/dq coordinate conversion module and a fourth abc/dq coordinate conversion module, and outputting a dq axis voltage control reference value v ref To the voltage control outer loop; the voltage control outer ring outputs dq axis current control reference value i ref The coupling between the voltage and current dq axes of the main circuit is decoupled by adopting a feedforward decoupling method on the basis of a PI controller through the current control inner ring, the voltage control outer ring and the current control inner ring, and PWM modulation voltage e is output r And outputting a signal to a PWM control module by the abc/dq coordinate conversion module IV, wherein the PWM control module outputs a control signal for controlling the main circuit inverter.
The invention also provides a method for modeling the impedance admittance of the network-structured three-phase converter, which comprises the following steps:
s1: constructing a complex frequency domain small signal dq model of a main circuit link after grid connection of the converter, and drawing a transfer function structure block diagram of the LC main circuit small signal model;
s2: constructing a complex frequency domain small signal dq model of a control link of the converter, wherein the complex frequency domain small signal dq model of the control link comprises a filtered output voltage v and a PWM modulation voltage e of the converter r Voltage coordinate transformation dq model of (2), inductance current i L The system comprises a power grid current i, a current coordinate conversion dq model, an active power reactive power dq model, a sampling low-pass filter dq model, a power control loop phase angle dq model, a voltage reference value dq model, a voltage control outer loop dq model, a current control inner loop dq model and a PWM control delay dq model;
s3: drawing a transfer function structure block diagram of a control link on the basis of the transfer function structure block diagram of the LC main circuit small signal model to obtain a full-system small signal structure block diagram, integrating and summarizing a complex frequency domain small signal dq model of the main circuit link and a complex frequency domain small signal dq model of a converter control link into an output voltage v filtered by the converter, an injected power grid current i and an inductance current i L And small signal models related to all set reference powers, and drawingAnd (3) a simplified full-system small signal structure block diagram is obtained, and a network three-phase converter impedance model and a network three-phase converter admittance model are formed according to the simplified full-system small signal structure block diagram.
In order to optimize the technical scheme, the specific measures adopted further comprise:
further, in S1, the complex frequency domain small signal dq model of the main circuit link specifically includes:
Figure BDA0004158001630000031
wherein Δe d And Δe q The disturbance quantity of the component of the unfiltered output voltage e of the converter on the d axis and the disturbance quantity of the component on the q axis are s being complex frequency L f And R is R f Inductance and parasitic resistance of LC filter, w 0 For the rated angular frequency of the converter, deltai Ld And Δi Lq Respectively the inductance current i L Disturbance variable of component on d-axis and disturbance variable of component on q-axis, Δv d And Deltav q The disturbance quantity of the component of the output voltage v of the converter on the d axis and the disturbance quantity of the component on the q axis through the filter are respectively; Δi d And Δi q The disturbance quantity of the component of the injection grid current i on the d axis and the disturbance quantity of the component on the q axis are respectively C f For filtering capacitance, Z L Is a filter inductance impedance transfer function matrix expressed as
Figure BDA0004158001630000032
Y C Is a filter capacitance admittance transfer function matrix expressed as +.>
Figure BDA0004158001630000033
Further in S2, the filtered output voltage v and PWM modulated voltage e of the converter r Is transformed dq model of the voltage coordinate of (d) and the inductance current i L And the current coordinate conversion dq model of the injected grid current i is respectively as follows:
Figure BDA0004158001630000041
in the method, in the process of the invention,
Figure BDA0004158001630000042
the small signal disturbance component of the filtered output voltage v of the converter on the d axis under the dq rotation coordinate system of the converter;
Figure BDA0004158001630000043
The small signal disturbance component of the filtered output voltage v of the converter on the q axis under the dq rotation coordinate system of the converter;
Figure BDA0004158001630000044
For small signal disturbance component of the filtered output voltage v of the converter on d axis under the system dq rotation coordinate system,
Figure BDA0004158001630000045
the small signal disturbance component of the filtered output voltage v of the converter on the q axis under the system dq rotating coordinate system; delta theta is the rotation angle phase difference of two coordinate systems, D v Is the coordinate transformation coefficient of the filtered output voltage v of the converter, expressed as
Figure BDA0004158001630000046
Figure BDA0004158001630000047
For the component of the steady-state value of the filtered output voltage v of the converter on the q-axis in the system dq rotation coordinate system,/->
Figure BDA0004158001630000048
The component of the steady-state value of the filtered output voltage v of the converter on the d-axis under the system dq rotation coordinate system;
Figure BDA0004158001630000049
Injecting power grid electricity into current transformer dq rotation coordinate systemSmall signal disturbance component of stream i on d-axis,/->
Figure BDA00041580016300000410
Injecting a small signal disturbance component of the power grid current i on the q axis into the current transformer dq rotation coordinate system;
Figure BDA00041580016300000411
And->
Figure BDA00041580016300000412
Respectively injecting small signal disturbance components of the grid current i on d axis and q axis under the system dq rotating coordinate system; d (D) i Is the coordinate transformation coefficient of the current i of the converter injected into the grid, expressed as +.>
Figure BDA00041580016300000413
Figure BDA00041580016300000414
And->
Figure BDA00041580016300000415
Respectively injecting components of steady-state values of the power grid current i on d-axis and q-axis under the system dq rotating coordinate system;
Figure BDA00041580016300000416
And->
Figure BDA00041580016300000417
Respectively, the inductive current i under the rotating coordinate system of the current transformer dq L Small signal disturbance component on d-axis and q-axis,/->
Figure BDA00041580016300000418
And->
Figure BDA00041580016300000419
Respectively, inductance current i under system dq rotating coordinate system L Small signal disturbance component on D-axis and q-axis, D iL Is the inductance current i L Coordinate transformation of (a)Coefficient, expressed as->
Figure BDA0004158001630000051
Figure BDA0004158001630000052
And->
Figure BDA0004158001630000053
Respectively, inductance current i under system dq rotating coordinate system L The components of the steady state values of (c) on the d-axis and q-axis;
Figure BDA0004158001630000054
And->
Figure BDA0004158001630000055
PWM modulation voltage e under system dq rotation coordinate system r Small signal disturbance component on d-axis and q-axis,/->
Figure BDA0004158001630000056
And->
Figure BDA0004158001630000057
PWM modulation voltage e under the rotation coordinate system of current transformer dq respectively r Small signal disturbance component in D-axis and q-axis, D er Is PWM modulation voltage e r Is expressed as +.>
Figure BDA0004158001630000058
Figure BDA0004158001630000059
And->
Figure BDA00041580016300000510
PWM modulation voltage e under system dq rotation coordinate system r The components of the steady state values of (c) on the d-axis and q-axis.
Further, in S2, the active power reactive power dq model specifically includes:
Figure BDA00041580016300000511
in the formula DeltaS e Is a complex power matrix, expressed as
Figure BDA00041580016300000512
ΔP e And DeltaQ e Respectively calculating an instantaneous active power small signal value and an instantaneous reactive power small signal value by a power calculation module; g S-V Is the power voltage coefficient, expressed as
Figure BDA00041580016300000513
G S-I Is the power current coefficient, expressed as +.>
Figure BDA00041580016300000514
Further, in S2, the sampling low-pass filter dq model is specifically:
Figure BDA00041580016300000515
wherein G is LPF A dq model representing the sampled low pass filter; w (w) c Is the cut-off angular frequency of the low pass filter.
Further, in S2, the power control loop phase angle dq model is specifically:
Figure BDA00041580016300000516
the voltage reference value dq model is specifically:
Figure BDA0004158001630000061
wherein G is A Is the relation coefficient between the phase difference and the power and represents
Figure BDA0004158001630000062
w 0 The rated angular frequency of the current transformer operation; j is the virtual inertia of the active power loop; ΔS ref Representation->
Figure BDA0004158001630000063
ΔP ref And DeltaQ ref The active power input reference value small signal variable and the reactive power input reference value small signal variable are respectively represented; deltav dref And Deltav qref Respectively the reference voltage small signal value Deltav ref The component K on the dq axis is the reactive power loop integral coefficient; v ref Is the reference voltage; g B And G C Respectively indicate->
Figure BDA0004158001630000064
And->
Figure BDA0004158001630000065
The relation coefficient between the reference voltage and the power and the relation coefficient between the output voltage of the converter through the filter are respectively; d (D) p And D q The active damping coefficient and the reactive damping coefficient are respectively.
Further, in S2, the voltage control outer loop dq model specifically includes:
Figure BDA0004158001630000066
the current control inner ring dq model specifically comprises:
Figure BDA0004158001630000067
wherein Δi dref And Δi qref Reference current small signal values delta i respectively output by the voltage control outer ring ref Components on the d-axis and q-axis;
Figure BDA0004158001630000071
representation of
Figure BDA0004158001630000072
For the voltage PI control transfer function, < >>
Figure BDA0004158001630000073
And->
Figure BDA0004158001630000074
The proportional gain coefficient and the integral gain coefficient of the voltage control outer ring are respectively;
Figure BDA0004158001630000075
Representation->
Figure BDA0004158001630000076
The decoupling coefficient of the outer ring is controlled by voltage;
Figure BDA00041580016300000718
Representation->
Figure BDA0004158001630000077
For the current PI control transfer function, < >>
Figure BDA0004158001630000078
And->
Figure BDA0004158001630000079
The proportional gain coefficient and the integral gain coefficient of the current control inner loop are respectively;
Figure BDA00041580016300000710
Representation->
Figure BDA00041580016300000711
The decoupling coefficient is controlled for the current.
Further, in S2, the dq model of the PWM control delay is specifically:
Figure BDA00041580016300000712
wherein G is d For PWM control of a matrix of delay transfer functions, the matrix is represented
Figure BDA00041580016300000713
K d For the transfer function of the delay element +.>
Figure BDA00041580016300000714
Approximation by second order Pade->
Figure BDA00041580016300000715
Figure BDA00041580016300000716
To delay period f s Is the switching frequency.
Further in S3, the filtered output voltage v of the and converter is injected into the grid current i and the inductance current i L The small signal model related to each set reference power is specifically:
Figure BDA00041580016300000717
Figure BDA0004158001630000081
wherein H is v The relation between PWM modulation voltage and the output voltage of the converter through the filter under the system coordinate system is obtained; h i The relation between PWM modulation voltage and grid current injected into the converter is adopted; h Sref Setting a relation between PWM modulation voltage and reference power of the converter; h iL The relation between PWM modulation voltage and main circuit inductance current is adopted;
the impedance model of the network-structured three-phase converter is as follows:
Δv s ＝M·ΔS ref -Z·Δi s
Figure BDA0004158001630000082
the admittance model of the network-structured three-phase converter is as follows:
Δi s ＝N·ΔS ref -Y·Δv s
Figure BDA0004158001630000083
wherein M is a small signal relation expression between the output voltage of the converter and a power reference value under a system coordinate system; z is the output impedance expression of the converter; n is a small signal relation expression between the output current of the converter and a power reference value under a system coordinate system; y is the output admittance expression of the converter.
The beneficial effects of the invention are as follows:
(1) Aiming at the technical problems existing in the prior art, the invention provides a grid-structured three-phase converter which can operate under off-grid and on-grid conditions and has stronger applicability compared with the grid-structured converter;
(2) According to the invention, dq modeling is carried out on the network-structured three-phase current transformer, the constructed impedance and admittance model adopts modularized calculation, the realization method is simple, the precision is higher, the calling application is flexible, the network-structured three-phase current transformer can be used for constructing the parallel impedance admittance model of a plurality of current transformers, and the network-structured three-phase current transformer is beneficial to providing model reference for researching the stability problem of the current transformer.
Drawings
Fig. 1 is a topological diagram of a network three-phase converter according to the present invention;
FIG. 2 is a power control loop control block diagram;
FIG. 3 is a diagram of dq decoupling relationships between the voltage control outer loop and the current control inner loop control block diagrams, and the control loop and the main circuit controlled object;
FIG. 4 is a schematic diagram of a flow chart of a modeling method for modeling impedance and admittance models of a network-structured three-phase current transformer;
FIG. 5 is a block diagram of the transfer function of the LC main circuit small signal model;
FIG. 6 is a block diagram of a full system small signal architecture including a main circuit and control links;
fig. 7 is a simplified block diagram of a full system small signal architecture.
Detailed Description
The invention will now be described in further detail with reference to the accompanying drawings.
In one embodiment, the invention provides a network three-phase converter, and a topological diagram of the three-phase converter is shown in fig. 1, and the network three-phase converter comprises a main circuit link and a control link;
the main circuit link comprises a three-phase six-arm inverter and an LC filter electrically connected with the three-phase six-arm inverter; the LC filter comprises an inductance L f And filter capacitor C f The inductance L f One end of the inductor L is connected with the output end of the three-phase six-arm inverter f The other end of (a) is connected with a filter capacitor C f And a bus bar, a capacitor C f The other end of the first electrode is grounded;
the control link comprises a first abc/dq coordinate transformation module, a second abc/dq coordinate transformation module, a third abc/dq coordinate transformation module, a fourth abc/dq coordinate transformation module, a power calculation module, a low-pass filter, a power control loop, a voltage control outer loop, a current control inner loop and a PWM control module;
the inductance L f Is of the inductance current i of (2) L The power calculation module receives a d-axis component voltage value and a q-axis component voltage value output by the abc/dq coordinate conversion module II and a d-axis component current value and a q-axis component current value output by the abc/dq coordinate conversion module III, and outputs instantaneous active power P e And instantaneous reactive power Q e The low-pass filter filters the instantaneous active power P output by the calculation module e And instantaneous reactive power Q e And outputting the average active power and the average reactive power to the power control loop; the power control loop is based on virtualQuasi-synchronous control strategy for receiving active power reference value P ref And reactive power reference value Q ref Outputting synchronous phase angles theta of a main circuit to a first abc/dq coordinate conversion module, a second abc/dq coordinate conversion module, a third abc/dq coordinate conversion module and a fourth abc/dq coordinate conversion module, and outputting a dq axis voltage control reference value v ref To the voltage control outer loop; as shown in fig. 2, a power control loop control block diagram; the voltage control outer ring outputs dq axis current control reference value i ref The coupling between the voltage and current dq axes of the main circuit is decoupled by adopting a feedforward decoupling method on the basis of a PI controller through the current control inner ring, the voltage control outer ring and the current control inner ring, and PWM modulation voltage e is output r And outputting a signal to a PWM control module by the abc/dq coordinate conversion module IV, wherein the PWM control module outputs a control signal for controlling the main circuit inverter. Fig. 3 shows the dq decoupling relationship between the voltage control outer loop and the current control inner loop, and the control loop and the main circuit controlled object.
In another embodiment, the invention provides a network three-phase converter impedance admittance modeling method, in order to construct dq impedance and admittance models of the network three-phase converter, the invention adopts a calculation method for constructing the dq models step by step, performs modular analysis, accumulates layer by layer, integrates calculation, and forms a network three-phase converter dq impedance model based on virtual synchronous power control, taking power sampling low-pass filtering into consideration and taking dq decoupling voltage and current PI double-loop control into consideration, and the process is shown in fig. 4, and specifically comprises the following steps:
s1: constructing a complex frequency domain small signal dq model of a main circuit link after grid connection of the converter, and drawing a transfer function structure block diagram of the LC main circuit small signal model;
s2: constructing a complex frequency domain small signal dq model of a control link of the converter, wherein the complex frequency domain small signal dq model of the control link comprises a filtered output voltage v and a PWM modulation voltage e of the converter r Voltage coordinate transformation dq model of (2), inductance current i L And a current coordinate conversion dq model, an active power reactive power dq model, a sampling low-pass filter dq model, a power control loop phase angle dq model of the injected grid current iA dq model of a voltage reference value, a dq model of a voltage control outer ring, a dq model of a current control inner ring and a dq model of PWM control delay;
s3: drawing a transfer function structure block diagram of a control link on the basis of the transfer function structure block diagram of the LC main circuit small signal model to obtain a full-system small signal structure block diagram, integrating and summarizing a complex frequency domain small signal dq model of the main circuit link and a complex frequency domain small signal dq model of a converter control link into an output voltage v filtered by the converter, an injected power grid current i and an inductance current i L And drawing a simplified full-system small signal structure block diagram according to each small signal model related to the set reference power, and arranging and forming a network three-phase converter impedance model and a network three-phase converter admittance model according to the simplified full-system small signal structure block diagram.
Specifically, a complex frequency domain small signal dq model of a main circuit link is firstly constructed according to the following formula:
Figure BDA0004158001630000111
wherein Δe d And Δe q The disturbance quantity of the component of the unfiltered output voltage e of the converter on the d axis and the disturbance quantity of the component on the q axis are s being complex frequency L f And R is R f Inductance and parasitic resistance of LC filter, w 0 For the rated angular frequency of the converter, deltai Ld And Δi Lq Respectively the inductance current i L Disturbance variable of component on d-axis and disturbance variable of component on q-axis, Δv d And Deltav q The disturbance quantity of the component of the output voltage v of the converter on the d axis and the disturbance quantity of the component on the q axis through the filter are respectively; Δi d And Δi q The disturbance quantity of the component of the injection grid current i on the d axis and the disturbance quantity of the component on the q axis are respectively C f For filtering capacitance, Z L Is a filter inductance impedance transfer function matrix expressed as
Figure BDA0004158001630000112
Y C Is a filter capacitance admittance transfer function matrix expressed as +.>
Figure BDA0004158001630000113
The main circuit equation is thus:
Figure BDA0004158001630000114
to draw a transfer function block diagram of the LC main circuit small signal model as shown in fig. 5.
And secondly, considering a complex frequency domain small signal impedance model of the control link. The whole system is divided into a system dq rotating coordinate system and a converter dq rotating coordinate system, and in a stable running state, the two coordinate systems are overlapped, and in an actual power grid, when the power grid voltage is slightly disturbed, a phase difference exists between the two coordinate systems. Each variable in the coordinate system of the system is x s The phase angle is theta, and each variable in the converter coordinate system is x c Phase angle of theta c The method comprises the steps of carrying out a first treatment on the surface of the The rotation angle phase difference of the two coordinate systems is delta theta, and the phase angle relation between the system coordinate system and the converter coordinate system is theta c =θ+Δθ. The Park transformation matrix can be used for obtaining a transformation matrix T for changing the system coordinates into the converter coordinates Δq The following is shown:
Figure BDA0004158001630000121
considering Δθ≡0, sin (Δθ) ≡Δθ, cos (Δθ) ≡1 may be approximated to obtain the conversion relation expression between small signal disturbance of voltage (or current) variable in two coordinate systems of the system and the controller as follows:
Figure BDA0004158001630000122
wherein,,
Figure BDA0004158001630000123
and +.>
Figure BDA0004158001630000124
The small signal disturbance components of the voltage (or current) variable on the dq axis under the coordinate system of the converter and the coordinate system of the system are respectively;
Figure BDA0004158001630000125
And->
Figure BDA0004158001630000126
Is the steady-state value X of voltage (or current) under the system coordinate system s Components on the dq axis.
The output voltage v and the filter inductance current i of the converter are sequentially constructed according to the method L Current i, current of converter injection into grid and PWM modulation voltage e r The voltage-current variable coordinate conversion dq model of (2) is as follows:
Figure BDA0004158001630000127
in the method, in the process of the invention,
Figure BDA0004158001630000128
the small signal disturbance component of the filtered output voltage v of the converter on the d axis under the dq rotation coordinate system of the converter;
Figure BDA0004158001630000129
The small signal disturbance component of the filtered output voltage v of the converter on the q axis under the dq rotation coordinate system of the converter;
Figure BDA00041580016300001210
For small signal disturbance component of the filtered output voltage v of the converter on d axis under the system dq rotation coordinate system,
Figure BDA00041580016300001211
the small signal disturbance component of the filtered output voltage v of the converter on the q axis under the system dq rotating coordinate system; delta theta is twoRotational angle phase difference of coordinate system, D v Is the coordinate transformation coefficient of the filtered output voltage v of the converter, expressed as
Figure BDA00041580016300001212
Figure BDA00041580016300001213
For the component of the steady-state value of the filtered output voltage v of the converter on the q-axis in the system dq rotation coordinate system,/->
Figure BDA0004158001630000131
The component of the steady-state value of the filtered output voltage v of the converter on the d-axis under the system dq rotation coordinate system;
Figure BDA0004158001630000132
Injecting a small signal disturbance component of a grid current i on a d axis into a dq rotating coordinate system of the converter, < >>
Figure BDA00041580016300001322
Injecting a small signal disturbance component of the power grid current i on the q axis into the current transformer dq rotation coordinate system;
Figure BDA0004158001630000133
And->
Figure BDA0004158001630000134
Respectively injecting small signal disturbance components of the grid current i on d axis and q axis under the system dq rotating coordinate system; d (D) i Is the coordinate transformation coefficient of the current i of the converter injected into the grid, expressed as +.>
Figure BDA0004158001630000135
Figure BDA0004158001630000136
And->
Figure BDA0004158001630000137
Respectively, system dq rotating seatInjecting components of steady-state values of the power grid current i on d-axis and q-axis under the standard system;
Figure BDA0004158001630000138
And->
Figure BDA0004158001630000139
Respectively, the inductive current i under the rotating coordinate system of the current transformer dq L Small signal disturbance component on d-axis and q-axis,/->
Figure BDA00041580016300001310
And->
Figure BDA00041580016300001323
Respectively, inductance current i under system dq rotating coordinate system L Small signal disturbance component on D-axis and q-axis, D iL Is the inductance current i L Is expressed as +.>
Figure BDA00041580016300001311
Figure BDA00041580016300001312
And->
Figure BDA00041580016300001313
Respectively, inductance current i under system dq rotating coordinate system L The components of the steady state values of (c) on the d-axis and q-axis;
Figure BDA00041580016300001314
And->
Figure BDA00041580016300001315
PWM modulation voltage e under system dq rotation coordinate system r Small signal disturbance component on d-axis and q-axis,/->
Figure BDA00041580016300001316
And->
Figure BDA00041580016300001317
PWM modulation voltage e under the rotation coordinate system of current transformer dq respectively r Small signal disturbance component in D-axis and q-axis, D er Is PWM modulation voltage e r Is expressed as +.>
Figure BDA00041580016300001318
Figure BDA00041580016300001319
And->
Figure BDA00041580016300001320
PWM modulation voltage e under system dq rotation coordinate system r The components of the steady state values of (c) on the d-axis and q-axis.
The voltage-current variable coordinate conversion equation can be written as:
Figure BDA00041580016300001321
obviously the converter output voltage v, the filter inductor current i L Current i, current of converter injection into grid and PWM modulation voltage e r The small signal conversion relation of the three-phase converter is not separated from the rotation angle phase difference delta theta of the two coordinate systems, and the phase difference delta theta is found to be closely related to the output power of the main circuit of the converter according to the control strategy block diagram of the network-structured three-phase converter.
The active power and reactive power dq model of the network-structured three-phase converter is constructed according to the following steps:
Figure BDA0004158001630000141
in the formula DeltaS e Is a complex power matrix, expressed as
Figure BDA0004158001630000142
ΔP e And DeltaQ e The instantaneous active power small signal value and the instantaneous reactive power small signal value calculated by the power calculation module are respectively；G S-V Is the power voltage coefficient, expressed as
Figure BDA0004158001630000143
G S-I Is the power current coefficient, expressed as +.>
Figure BDA0004158001630000144
The power calculation equation for shorthand is obtained as follows:
ΔS e ＝G S-V ·Δv s +G S-I ·Δi s (8)
further, if the first order low pass filter calculated by taking the output power samples into account, the low pass filter transfer function is expressed as:
Figure BDA0004158001630000145
wherein K is LPF Is a low pass filter transfer function; w (w) c Is the cut-off angular frequency of the low pass filter.
Combining active power and reactive power into a complex power matrix for integral calculation, and constructing a sampling low-pass filter dq model according to the following steps:
Figure BDA0004158001630000146
wherein G is LPF Representing the dq model of the sampled low pass filter.
Further according to
Figure BDA0004158001630000147
And->
Figure BDA0004158001630000148
The described power control loop strategy, the column write power control loop small signal control equation set is:
Figure BDA0004158001630000151
wherein w and w 0 The actual angular frequency and the rated angular frequency of the current transformer are respectively operated; the Δw converter runs an actual angular frequency small signal variable; p (P) ref And Q ref Inputting a reference value for active power and reactive power; ΔP ref And DeltaQ ref Representing the input reference value small signal variable of the active power and the reactive power; j is the virtual inertia of the active power loop; k is the integral coefficient of the reactive power loop; d (D) p And D q The active damping coefficient and the reactive damping coefficient are respectively; v N And v is the actual capacitor voltage and the nominal voltage, respectively; v ref Is the reference voltage.
According to the described power control loop small signal equation set, the active power and the reactive power are regarded as the whole to be operated, so that the phase small signal variable of the converter is repeatedly listed as a matrix of 1*2, and a phase angle dq model of the power control loop is constructed as follows:
Figure BDA0004158001630000152
voltage reference dq model:
Figure BDA0004158001630000153
wherein G is A Is the relation coefficient between the phase difference and the power and represents
Figure BDA0004158001630000154
w 0 The rated angular frequency of the current transformer operation; j is the virtual inertia of the active power loop; ΔS ref Representation->
Figure BDA0004158001630000155
ΔP ref And DeltaQ ref The active power input reference value small signal variable and the reactive power input reference value small signal variable are respectively represented; deltav dref And Deltav qref Respectively are provided withIs the small signal value Deltav of the reference voltage ref The component K on the dq axis is the reactive power loop integral coefficient; v ref Is the reference voltage; g B And G C Respectively indicate->
Figure BDA0004158001630000161
And->
Figure BDA0004158001630000162
The relation coefficient between the reference voltage and the power and the relation coefficient between the output voltage of the converter through the filter are respectively; d (D) p And D q The active damping coefficient and the reactive damping coefficient are respectively.
The power control loop equation, abbreviated, is obtained as:
Figure BDA0004158001630000163
in this embodiment, the main circuit inductance current i in the dq coordinate system is found according to the main circuit equation L And coupling exists between dq axes of the grid-connected voltage v, so that in order to eliminate coupling interaction, control precision and dynamic response characteristics are improved, and the influence of a coupling term is reduced by adopting feedforward decoupling control. Meanwhile, in order to track the voltage reference signal output by the power control loop, zero steady-state error is realized, and the voltage control outer loop adopts PI control; in order to improve the dynamic response performance of the system and realize zero steady-state error, the current control inner loop also adopts PI control. The dq model of the voltage control outer loop is constructed as follows:
Figure BDA0004158001630000164
the dq model of the current control inner loop is constructed as follows:
Figure BDA0004158001630000165
wherein Δi dref And Δi qref Reference current small signal values delta i respectively output by the voltage control outer ring ref Components on the d-axis and q-axis;
Figure BDA0004158001630000166
representation->
Figure BDA0004158001630000167
For the voltage PI control transfer function, < >>
Figure BDA0004158001630000168
And->
Figure BDA0004158001630000169
The proportional gain coefficient and the integral gain coefficient of the voltage control outer ring are respectively;
Figure BDA0004158001630000171
Representation->
Figure BDA0004158001630000172
The decoupling coefficient of the outer ring is controlled by voltage;
Figure BDA0004158001630000173
Representation->
Figure BDA0004158001630000174
For the current PI control transfer function, < >>
Figure BDA0004158001630000175
And->
Figure BDA0004158001630000176
The proportional gain coefficient and the integral gain coefficient of the current control inner loop are respectively;
Figure BDA0004158001630000177
Representation->
Figure BDA0004158001630000178
For controlling the inner loop decoupling coefficient for current。/>
The voltage-current double-loop control equation is:
Figure BDA0004158001630000179
further, in this embodiment, a 1.5 sampling period delay generated by PWM digital control is considered, and the transfer function expression of the delay link is:
Figure BDA00041580016300001710
wherein K is d Is a transfer function of a time delay link;
Figure BDA00041580016300001711
to delay period f s Is the switching frequency.
Due to K d As a nonlinear function, a new transfer function of the delay link can be obtained by second-order pad approximation according to the following formula:
Figure BDA00041580016300001712
the small signal model of PWM control delay is established according to the following steps:
Figure BDA00041580016300001713
wherein G is d For PWM control of a matrix of delay transfer functions, the matrix is represented
Figure BDA00041580016300001714
The PWM control delay equation is obtained by short:
Δe＝G d ·Δe r (21)
finally, a system-wide small signal block diagram including the main circuit and the control link is drawn as shown in fig. 6. Will be described in(6) (8) (10) (14) (17) integrating and summarizing to obtain output voltage v of the converter through a filter, injection grid current i and main circuit inductance current i L Setting a reference power related small signal model as follows:
Figure BDA0004158001630000181
Figure BDA0004158001630000182
wherein H is v The relation between PWM modulation voltage and the output voltage of the converter through the filter under the system coordinate system is obtained; h i The relation between PWM modulation voltage and grid current injected into the converter is adopted; h Sref Setting a relation between PWM modulation voltage and reference power of the converter; h iL Is the relationship between the PWM modulation voltage and the main circuit inductor current.
Drawing a simplified full-system small signal structure block diagram shown in fig. 7 according to formulas (2) (22) and (21), and arranging to obtain a network-structured three-phase converter impedance model as follows:
Δv s ＝M·ΔS ref -Z·Δi s
Figure BDA0004158001630000183
the admittance model of the net-structured three-phase converter is obtained by arrangement:
Δi s ＝N·ΔS ref -Y·Δv s
Figure BDA0004158001630000184
wherein M is a small signal relation expression between the output voltage of the converter and a power reference value under a system coordinate system; z is the output impedance expression of the converter; n is a small signal relation expression between the output current of the converter and a power reference value under a system coordinate system; y is the output admittance expression of the converter.
The above is only a preferred embodiment of the present invention, and the protection scope of the present invention is not limited to the above examples, and all technical solutions belonging to the concept of the present invention belong to the protection scope of the present invention. It should be noted that modifications and adaptations to the invention without departing from the principles thereof are intended to be within the scope of the invention as set forth in the following claims.

Claims (10)

1. The network-structured three-phase converter is characterized by comprising a main circuit link and a control link;
the main circuit link comprises a three-phase six-arm inverter and an LC filter electrically connected with the three-phase six-arm inverter; the LC filter comprises an inductance L f And filter capacitor C f The inductance L f One end of the inductor L is connected with the output end of the three-phase six-arm inverter f The other end of (a) is connected with a filter capacitor C f And a bus bar, a capacitor C f The other end of the first electrode is grounded;
the control link comprises a first abc/dq coordinate transformation module, a second abc/dq coordinate transformation module, a third abc/dq coordinate transformation module, a fourth abc/dq coordinate transformation module, a power calculation module, a low-pass filter, a power control loop, a voltage control outer loop, a current control inner loop and a PWM control module;
the inductance L f Is of the inductance current i of (2) L The power calculation module receives a d-axis component voltage value and a q-axis component voltage value output by the abc/dq coordinate conversion module II and a d-axis component current value and a q-axis component current value output by the abc/dq coordinate conversion module III, and outputs instantaneous active power P e And instantaneous reactive power Q e The low-pass filter filters the instantaneous active power P output by the calculation module e And instantaneous reactive power Q e And output averagePower and average reactive power to power control loop; the power control loop receives the active power reference value P based on a virtual synchronization control strategy ref And reactive power reference value Q ref Outputting synchronous phase angles theta of a main circuit to a first abc/dq coordinate conversion module, a second abc/dq coordinate conversion module, a third abc/dq coordinate conversion module and a fourth abc/dq coordinate conversion module, and outputting a dq axis voltage control reference value v ref To the voltage control outer loop; the voltage control outer ring outputs dq axis current control reference value i ref The coupling between the voltage and current dq axes of the main circuit is decoupled by adopting a feedforward decoupling method on the basis of a PI controller through the current control inner ring, the voltage control outer ring and the current control inner ring, and PWM modulation voltage e is output r And outputting a signal to a PWM control module by the abc/dq coordinate conversion module IV, wherein the PWM control module outputs a control signal for controlling the main circuit inverter.
2. The method for modeling the impedance admittance of the network-structured three-phase converter is characterized by comprising the following steps of:
s1: constructing a complex frequency domain small signal dq model of a main circuit link after grid connection of the converter to obtain a main circuit equation, and drawing a transfer function structure block diagram of the LC main circuit small signal model according to the main circuit equation;
s2: constructing a complex frequency domain small signal dq model of a control link of the converter, wherein the complex frequency domain small signal dq model of the control link comprises a filtered output voltage v and a PWM modulation voltage e of the converter r Voltage coordinate transformation dq model of (2), inductance current i L The system comprises a power grid current i, a current coordinate conversion dq model, an active power reactive power dq model, a sampling low-pass filter dq model, a power control loop phase angle dq model, a voltage reference value dq model, a voltage control outer loop dq model, a current control inner loop dq model and a PWM control delay dq model;
s3: drawing a transfer function structure block diagram of a control link on the basis of the transfer function structure block diagram of the LC main circuit small signal model to obtain a full-system small signal structure block diagram, and carrying out complex frequency domain small signal dq model and current transformation on the main circuit linkThe complex frequency domain small signal dq model of the controller control link is integrated and summarized into output voltage v, injection grid current i and inductance current i which are filtered by the converter L And drawing a simplified full-system small signal structure block diagram according to each small signal model related to the set reference power, and arranging and forming a network three-phase converter impedance model and a network three-phase converter admittance model according to the simplified full-system small signal structure block diagram.
3. The method for modeling the impedance admittance of the network-structured three-phase converter according to claim 2, wherein in S1, the complex frequency domain small signal dq model of the main circuit link is specifically:
Figure FDA0004158001620000021
wherein Δe d And Δe q The disturbance quantity of the component of the unfiltered output voltage e of the converter on the d axis and the disturbance quantity of the component on the q axis are s being complex frequency L f And R is R f Inductance and parasitic resistance of LC filter, w 0 For the rated angular frequency of the converter, deltai Ld And Δi Lq Respectively the inductance current i L Disturbance variable of component on d-axis and disturbance variable of component on q-axis, Δv d And Deltav q The disturbance quantity of the component of the output voltage v of the converter on the d axis and the disturbance quantity of the component on the q axis through the filter are respectively; Δi d And Δi q The disturbance quantity of the component of the injection grid current i on the d axis and the disturbance quantity of the component on the q axis are respectively C f For filtering capacitance, Z L Is a filter inductance impedance transfer function matrix expressed as
Figure FDA0004158001620000022
Y C Is a filter capacitance admittance transfer function matrix expressed as +.>
Figure FDA0004158001620000023
4. The method of modeling impedance admittance of a three-phase grid-structured converter according to claim 2, characterized in that in S2 the filtered output voltage v and PWM modulated voltage e of the converter r Is transformed dq model of the voltage coordinate of (d) and the inductance current i L And the current coordinate conversion dq model of the injected grid current i is respectively as follows:
Figure FDA0004158001620000031
in the method, in the process of the invention,
Figure FDA0004158001620000032
the small signal disturbance component of the filtered output voltage v of the converter on the d axis under the dq rotation coordinate system of the converter;
Figure FDA0004158001620000033
The small signal disturbance component of the filtered output voltage v of the converter on the q axis under the dq rotation coordinate system of the converter;
Figure FDA0004158001620000034
for small signal disturbance component of the filtered output voltage v of the converter in the system dq rotation coordinate system on d-axis,/->
Figure FDA0004158001620000035
The small signal disturbance component of the filtered output voltage v of the converter on the q axis under the system dq rotating coordinate system; Δq is the rotation angle phase difference of two coordinate systems, D v Is the coordinate transformation coefficient of the filtered output voltage v of the converter, expressed as +.>
Figure FDA0004158001620000036
Figure FDA0004158001620000037
For the component of the steady-state value of the filtered output voltage v of the converter on the q-axis in the system dq rotation coordinate system,/->
Figure FDA0004158001620000038
The component of the steady-state value of the filtered output voltage v of the converter on the d-axis under the system dq rotation coordinate system;
Figure FDA0004158001620000039
Injecting a small signal disturbance component of a grid current i on a d axis into a dq rotating coordinate system of the converter, < >>
Figure FDA00041580016200000310
Injecting a small signal disturbance component of the power grid current i on the q axis into the current transformer dq rotation coordinate system;
Figure FDA00041580016200000311
And->
Figure FDA00041580016200000312
Respectively injecting small signal disturbance components of the grid current i on d axis and q axis under the system dq rotating coordinate system; d (D) i Is the coordinate transformation coefficient of the current i injected into the power grid by the converter, expressed as
Figure FDA00041580016200000313
Figure FDA00041580016200000314
And->
Figure FDA00041580016200000315
Respectively injecting components of steady-state values of the power grid current i on d-axis and q-axis under the system dq rotating coordinate system;
Figure FDA00041580016200000316
And->
Figure FDA00041580016200000317
Respectively, the inductive current i under the rotating coordinate system of the current transformer dq L Small signal disturbance components on the d-axis and q-axis,
Figure FDA00041580016200000318
and->
Figure FDA00041580016200000319
Respectively, inductance current i under system dq rotating coordinate system L Small signal disturbance component on D-axis and q-axis, D iL Is the inductance current i L Is expressed as +.>
Figure FDA0004158001620000041
Figure FDA0004158001620000042
And->
Figure FDA0004158001620000043
Respectively, inductance current i under system dq rotating coordinate system L The components of the steady state values of (c) on the d-axis and q-axis;
Figure FDA0004158001620000044
And->
Figure FDA0004158001620000045
PWM modulation voltage e under system dq rotation coordinate system r Small signal disturbance component on d-axis and q-axis,/->
Figure FDA0004158001620000046
And->
Figure FDA0004158001620000047
PWM modulation voltage e under the rotation coordinate system of current transformer dq respectively r Small signal disturbance components on the d-axis and q-axis,D er is PWM modulation voltage e r Is expressed as
Figure FDA0004158001620000048
Figure FDA0004158001620000049
And->
Figure FDA00041580016200000410
PWM modulation voltage e under system dq rotation coordinate system r The components of the steady state values of (c) on the d-axis and q-axis.
5. The method for modeling the impedance admittance of a three-phase network converter according to claim 2, wherein in S2, the active power reactive power dq model is specifically:
Figure FDA00041580016200000411
in the formula DeltaS e Is a complex power matrix, expressed as
Figure FDA00041580016200000412
ΔP e And DeltaQ e Respectively calculating an instantaneous active power small signal value and an instantaneous reactive power small signal value by a power calculation module; g S-V Is the power voltage coefficient, expressed as +.>
Figure FDA00041580016200000413
G S-I Is the power current coefficient, expressed as +.>
Figure FDA00041580016200000414
6. The method for modeling impedance admittance of a three-phase network converter according to claim 2, wherein in S2, the sampling low-pass filter dq model is specifically:
Figure FDA00041580016200000415
wherein G is LPF A dq model representing the sampled low pass filter; w (w) c Is the cut-off angular frequency of the low pass filter.
7. The method for modeling impedance admittance of a three-phase network converter according to claim 2, wherein in S2, the power control loop phase angle dq model is specifically:
Figure FDA0004158001620000051
the voltage reference value dq model is specifically:
Figure FDA0004158001620000052
wherein G is A Is the relation coefficient between the phase difference and the power and represents
Figure FDA0004158001620000053
w 0 The rated angular frequency of the current transformer operation; j is the virtual inertia of the active power loop; ΔS ref Representation->
Figure FDA0004158001620000054
ΔP ref And DeltaQ ref The active power input reference value small signal variable and the reactive power input reference value small signal variable are respectively represented; deltav dref And Deltav qref Respectively the reference voltage small signal value Deltav ref The component K on the dq axis is the reactive power loop integral coefficient; v ref Is the reference voltage; g B And G C Respectively indicate->
Figure FDA0004158001620000055
And->
Figure FDA0004158001620000056
The relation coefficient between the reference voltage and the power and the relation coefficient between the output voltage of the converter through the filter are respectively; d (D) p And D q The active damping coefficient and the reactive damping coefficient are respectively.
8. The method for modeling impedance admittance of a three-phase network converter according to claim 2, wherein in S2, the voltage control outer loop dq model is specifically:
Figure FDA0004158001620000057
the current control inner ring dq model specifically comprises:
Figure FDA0004158001620000061
wherein Δi dref And Δi qref Reference current small signal values delta i respectively output by the voltage control outer ring ref Components on the d-axis and q-axis;
Figure FDA0004158001620000062
representation->
Figure FDA0004158001620000063
For the voltage PI control transfer function, < >>
Figure FDA0004158001620000064
And->
Figure FDA0004158001620000065
Proportional gain for voltage controlled outer loopCoefficients and integral gain coefficients;
Figure FDA0004158001620000066
Representation->
Figure FDA0004158001620000067
The decoupling coefficient of the outer ring is controlled by voltage;
Figure FDA0004158001620000068
Representation->
Figure FDA0004158001620000069
For the current PI control transfer function, < >>
Figure FDA00041580016200000610
And->
Figure FDA00041580016200000611
The proportional gain coefficient and the integral gain coefficient of the current control inner loop are respectively;
Figure FDA00041580016200000612
Representation->
Figure FDA00041580016200000613
The decoupling coefficient is controlled for the current.
9. The method for modeling impedance admittance of a three-phase network converter according to claim 2, wherein in S2, the dq model of the PWM control delay is specifically:
Figure FDA00041580016200000614
wherein G is d For PWM control of a matrix of delay transfer functions, the matrix is represented
Figure FDA00041580016200000615
K d As a transfer function of the delay element,
Figure FDA00041580016200000616
approximation by second order Pade->
Figure FDA00041580016200000617
Figure FDA00041580016200000618
To delay period f s Is the switching frequency.
10. The method of modeling impedance admittance of three-phase grid-connected converter according to claim 2, wherein in S3, the filtered output voltage v of said converter, the injected grid current i, the inductor current i L The small signal model related to each set reference power is specifically:
Figure FDA0004158001620000071
Figure FDA0004158001620000072
wherein H is v The relation between PWM modulation voltage and the output voltage of the converter through the filter under the system coordinate system is obtained; h i The relation between PWM modulation voltage and grid current injected into the converter is adopted; h Sref Setting a relation between PWM modulation voltage and reference power of the converter; h iL The relation between PWM modulation voltage and main circuit inductance current is adopted;
the impedance model of the network-structured three-phase converter is as follows:
Δv s ＝M·ΔS ref -Z·Δi s
Figure FDA0004158001620000073
the admittance model of the network-structured three-phase converter is as follows:
Δi s ＝N·ΔS ref -Y·Δv s
Figure FDA0004158001620000074
wherein M is a small signal relation expression between the output voltage of the converter and a power reference value under a system coordinate system; z is the output impedance expression of the converter; n is a small signal relation expression between the output current of the converter and a power reference value under a system coordinate system; y is the output admittance expression of the converter.
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Publication number Priority date Publication date Assignee Title
CN116418049A (en) * 2023-06-08 2023-07-11 四川大学 Accurate admittance modeling method for sagging-controlled three-phase grid-connected inverter

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN116418049A (en) * 2023-06-08 2023-07-11 四川大学 Accurate admittance modeling method for sagging-controlled three-phase grid-connected inverter
CN116418049B (en) * 2023-06-08 2023-08-11 四川大学 Accurate admittance modeling method for sagging-controlled three-phase grid-connected inverter

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