CN116131475A - IPT system with highly integrated magnetic coupler and IPT system integration method - Google Patents

IPT system with highly integrated magnetic coupler and IPT system integration method Download PDF

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Publication number
CN116131475A
CN116131475A CN202310145023.0A CN202310145023A CN116131475A CN 116131475 A CN116131475 A CN 116131475A CN 202310145023 A CN202310145023 A CN 202310145023A CN 116131475 A CN116131475 A CN 116131475A
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coil
integrated
mutual inductance
transmitting
magnetic coupler
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Inventor
史可
冯天旭
蒋金橙
王佩月
洪承镐
唐春森
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Chongqing University of Post and Telecommunications
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Chongqing University of Post and Telecommunications
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/005—Mechanical details of housing or structure aiming to accommodate the power transfer means, e.g. mechanical integration of coils, antennas or transducers into emitting or receiving devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/10—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling
    • H02J50/12—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling of the resonant type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/40—Circuit arrangements or systems for wireless supply or distribution of electric power using two or more transmitting or receiving devices
    • H02J50/402—Circuit arrangements or systems for wireless supply or distribution of electric power using two or more transmitting or receiving devices the two or more transmitting or the two or more receiving devices being integrated in the same unit, e.g. power mats with several coils or antennas with several sub-antennas
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/70—Circuit arrangements or systems for wireless supply or distribution of electric power involving the reduction of electric, magnetic or electromagnetic leakage fields
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
    • H02M7/068—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode mounted on a transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/70—Energy storage systems for electromobility, e.g. batteries
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/7072—Electromobility specific charging systems or methods for batteries, ultracapacitors, supercapacitors or double-layer capacitors
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T90/00—Enabling technologies or technologies with a potential or indirect contribution to GHG emissions mitigation
    • Y02T90/10—Technologies relating to charging of electric vehicles
    • Y02T90/14—Plug-in electric vehicles

Abstract

The invention relates to a wireless power transmission technology, in particular to an IPT system with a highly integrated magnetic coupler and an IPT system integration method; three kinds of integrated coils and the design flow of the integrated coils are provided in the method; wherein, the integrated reverse transmitting coil is used for realizing better anti-offset characteristic and high efficiency; the secondary side integrated inductance coil aims to realize zero-voltage switching condition configuration, so that the output power and the anti-offset characteristic are not affected; the primary side integrated inductor coil is intended to achieve decoupling from other coils, further improving compactness without affecting characteristics.

Description

IPT system with highly integrated magnetic coupler and IPT system integration method
Technical Field
The invention relates to a wireless power transmission technology, in particular to an IPT system of a specific highly integrated magnetic coupler and an IPT system integration method.
Background
With the rapid growth of population and economy, the problems of shortage of traditional energy sources and environmental pollution are increasingly prominent. Magnetically coupled wireless power transfer (IPT) systems have proven to be an excellent solution with the features of avoiding cumbersome cables, availability of galvanic isolation, more free operation, weather protection, low maintenance and higher security. It has been widely used in biomedical implants, consumer electronics and electric vehicles.
The IPT system is free of the physical media, allowing considerable flexibility in the primary and secondary coils. However, mechanical independence makes perfect alignment difficult. The offset may cause a variation in mutual inductance between the coupled coils, resulting in a reduction in transmission power, instability, and more power loss. Thus, anti-migration performance is an important performance indicator of IPT systems.
Based on the former study, the improvement of the anti-migration performance of IPT systems was mainly studied from the following three aspects. The first is to reduce the variation in mutual inductance between the transmitting coil and the receiving coil by changing the shape of the magnetic coupling mechanism. The focus of these methods is to achieve stable single-coupled power transfer through the design of coil shape, core structure, coil polarity and winding method. The second is to use a control scheme to match the mutual inductance under offset. The extra control modules add complexity to the IPT system and the custom requirements for different systems impair versatility. The third is to design the compensation topology and parameters to achieve stable transmission power under mutual inductance variations. The constant output current characteristics are changed in the parameter configuration of these methods. Since the coupling channel is not changed in nature, its anti-offset effect is limited.
LCC compensation topologies have been widely adopted because they provide power proportional to the coupling coefficient and achieve a constant output current mode of operation for battery charging applications. However, it requires more inductive elements, which increases the complexity of the system. Meanwhile, the external compensation inductor has the following problems:
1) It requires additional space, which makes the system bulky;
2) When a large current passes through the compensation inductor, the heating problem is difficult to solve;
3) More magnetic cores are used, so that the cost is increased;
4) The modularization of the wireless power transmission device is not facilitated.
Disclosure of Invention
To solve the above problems, the present invention proposes an IPT system with a highly integrated magnetic coupler and an IPT system integration method in which three integrated coils are proposed. Wherein, the integrated reverse transmitting coil is used for realizing better anti-offset characteristic and high efficiency; the secondary side integrated inductor is intended to achieve Zero Voltage Switching (ZVS) condition configuration, leaving the output power and anti-offset characteristics unaffected; the primary side integrated inductor coil is intended to achieve decoupling from other coils, further improving compactness without affecting characteristics.
In a first aspect, the present invention provides a composition having a high degree ofAn IPT system of a highly integrated magnetic coupler, the IPT system comprising an inverter, a highly integrated magnetic coupler, a rectifier and a load; the highly integrated magnetic coupler comprises a transmitting end and a receiving end; the transmitting end comprises a transmitting coil L p1 Integrated reverse coil L p2 Primary side integrated inductor L pf A first magnetic core plate and a first shield plate; a first magnetic core plate is arranged on the first shielding plate, and a transmitting coil L p1 Is arranged on a first magnetic core plate and integrates a reverse coil L p2 Is arranged on the transmitting coil L p1 Inside, and integrate the reverse coil L p2 And a transmitting coil L p1 Is of a central symmetrical structure;
the receiving end comprises a receiving coil L s Secondary side integrated inductor L sf A second magnetic core plate and a second shield plate; a second magnetic core plate is arranged on the second shielding plate and is used for receiving the coil L s Is arranged on the second magnetic core plate, and the secondary side is integrated with an inductance coil L sf Is arranged on the receiving coil L s Inside.
Further, primary side integrated inductor L pf Is a double-stage coil, a transmitting coil L p1 And an integrated reverse coil L p2 4 double-stage coils are arranged between the two coils; the 1 double-stage coil has 2 current loops, the 4 double-stage coils have 8 current loops, and the polarities of every two current loops are opposite.
In a second aspect, based on the highly integrated magnetic coupler structure set forth in the first aspect, the present invention proposes an IPT system integration method with a highly integrated magnetic coupler, comprising the steps of:
s1, constructing an initial magnetic coupler according to the working frequency, the input voltage and the output power of an IPT system; the initial magnetic coupler comprises a transmitting end and a receiving end; the transmitting end is provided with a transmitting coil L p1 The receiving end is provided with a receiving coil L s The method comprises the steps of carrying out a first treatment on the surface of the Transmitting coil L p1 And a receiving coil L s The number of turns and the size of the coil are preset, and the transmitting coil L is defined p1 And a receiving coil L s The mutual inductance between them is M p1s ；
S2, arranging an integrated reverse coil L at a transmitting end of the initial magnetic coupler p2 And determines the integrated reverse coil L according to the output power of the IPT system p2 Turns of (2); definition of the transmitting coil L p1 Integrated reverse coil L p2 And receiving coil L s The mutual inductance difference between the two is the first mutual inductance M ps ；
S3, performing finite element analysis through MAXWELL, traversing the integrated reverse coil L p2 Determine the size of the first mutual inductance M ps If the alignment deviation is stable within 40%, the step S4 is entered if yes, otherwise, the traversal is continued;
s4, arranging a secondary side integrated inductance coil L at the receiving end of the initial magnetic coupler sf Defining a transmitting coil L p1 Integrated reverse coil L p2 With secondary side collective inductance coil L sf The mutual inductance difference between the two is the second mutual inductance M psf ；
S5, the second mutual inductance M psf With a first mutual inductance M ps The ratio of (2) is defined as epsilon; traversing secondary side integrated inductor L sf Judging whether epsilon is less than or equal to zeta and zeta is a desired susceptibility index; if yes, entering a step S5, otherwise, continuing traversing;
s6, arranging a primary side integrated inductance coil L at the transmitting end of the initial magnetic coupler pf Traversing primary side integrated inductor L based on output power of transmitting end of initial magnetic coupler pf And the number and size of turns in (a) to yield a final optimized highly integrated magnetic coupler.
Further, the circuit topology of the IPT system with the highly integrated magnetic coupler includes a primary side LCC topology compensation network and a secondary side LCC topology compensation network; the primary side LCC topology compensation network comprises a transmitting coil L p1 Integrated reverse coil L p2 Primary side integrated inductor L pf Capacitance C p And capacitor C pf The method comprises the steps of carrying out a first treatment on the surface of the The secondary side LCC topology compensation network comprises a receiving coil L s Secondary side integrated inductor L sf Capacitance C s And capacitor C sf ；
The primary side integrates an inductance coil L pf Capacitance C p Transmitting coil L p1 And an integrated reverse coil L p2 In series, capacitor C p Transmitting coil L p1 And an integrated reverse coil L p2 Circuit and capacitor C pf Parallel connection; wherein the coil L is transmitted p1 And an integrated reverse coil L p2 Is reversely connected in series; the secondary side is integrated with the inductance coil L sf Capacitance C s And a receiving coil L s In series, capacitor C s And a receiving coil L s Circuit and capacitor C sf Parallel connection;
transmitting coil L p1 Integrated reverse coil L p2 And receiving coil L s The mutual inductance difference between the two is the first mutual inductance M ps The method comprises the steps of carrying out a first treatment on the surface of the Transmitting coil L p1 Integrated reverse coil L p2 With secondary side collective inductance coil L sf The mutual inductance difference between the two is the second mutual inductance M psf The method comprises the steps of carrying out a first treatment on the surface of the Transmitting coil L p1 And an integrated reverse coil L p2 The mutual inductance between the two coils is the first same-side internal mutual inductance M p12 The method comprises the steps of carrying out a first treatment on the surface of the Secondary side collective inductance coil L sf And a receiving coil L s The mutual inductance between the two coils is the second same-side internal mutual inductance M ssf 。
Further, the second same side internal mutual inductance M ssf The equivalent is a T-shaped network, and is integrated with a secondary side LCC topology compensation network, and is decoupled by the following formula:
Figure BDA0004088891560000041
L sfe ＝L sf +M ssf
Figure BDA0004088891560000042
first same side internal mutual inductance M p12 The decoupling formula of (2) is expressed as:
Figure BDA0004088891560000043
wherein ,Lse Representing a receiving coilL s And a series compensation capacitor C s And ω represents the system resonant angular frequency, L sfe Representing a secondary side integrated inductor L sf And a second same-side internal mutual inductance M ssf Is equivalent to the superposition of C sfe Representing the secondary side parallel compensation capacitance C sf And a second same-side internal mutual inductance M ssf Is equivalent to the superposition of L p Representing the transmitting coil L p1 Integrated reverse coil L p2 And primary side series compensation capacitor C p Is equivalent to the superposition of (a).
Further, an equivalent circuit of a circuit topology package of the IPT system is obtained by adopting a basic harmonic approximation method, and the equivalent circuit is obtained by using a kirchhoff law:
Figure BDA0004088891560000051
the resonance relationship is expressed as:
ω 2 L pf C pf ＝ω 2 L p C pf ＝ω 2 L se C sfe ＝ω 2 L sfe C sfe ＝1
wherein ,
Figure BDA0004088891560000052
representing the inverted output current, ">
Figure BDA0004088891560000053
Indicating the current through the transmitting coil, < >>
Figure BDA0004088891560000054
Indicating the current through the receiving coil, < > and->
Figure BDA0004088891560000055
Representing the output current +.>
Figure BDA0004088891560000056
Represents the inversion output voltage, R e Representing the load that is commonly equivalent to the rectifier and the load.
Further, a kirchhoff law formula is simplified through a resonance relation, and the method is obtained:
Figure BDA0004088891560000057
derived from the simplified formula:
Figure BDA0004088891560000058
Figure BDA0004088891560000059
Figure BDA00040888915600000510
Figure BDA00040888915600000511
Figure BDA0004088891560000061
Figure BDA0004088891560000062
wherein ,
Figure BDA0004088891560000063
representing the inverted output voltage, ">
Figure BDA0004088891560000064
Modulo representing inverted output voltage +.>
Figure BDA0004088891560000065
Representing transmit coil currentIs (are) mould>
Figure BDA0004088891560000066
Mode, P, representing output current out Represents output power, K 1 Representing coefficients defined by system parameters, Z in Representing the total equivalent impedance of the system from the input.
The invention has the beneficial effects that:
in the highly integrated magnetic coupler provided by the invention, the three integrated coils (the integrated reverse coil, the primary side integrated inductance coil and the secondary side integrated inductance coil) and the main coil (the transmitting coil and the receiving coil) share the magnetic core and the space in the layers, so that the compactness of the coupler is greatly improved.
In the IPT system integration method with the highly integrated magnetic coupler, firstly, the coupling change characteristic is changed by arranging an integrated reverse coil at a transmitting end, so as to obtain equivalent mutual inductance difference to replace the original single coupling. Second, a secondary side integrated inductor is provided at the receiving end and introduces a design variable to achieve the desired inverter output phase angle to ensure ZVS conditions. Finally, the primary side integrated inductance coil is arranged at the transmitting end, so that the primary side integrated inductance coil can be decoupled from other coils in an offset state and an alignment state, and full magnetic integration is more feasible.
Drawings
FIG. 1 is a schematic perspective view of a highly integrated magnetic coupler according to the present invention;
FIG. 2 is a circuit topology of an IPT system based on a highly integrated magnetic coupler of the present invention;
FIG. 3 is an equivalent circuit diagram of the ipsilateral internal mutual inductance decoupling of the secondary side LCC topology compensation network of the present invention;
FIG. 4 is an equivalent circuit diagram of an IPT system based on a highly integrated magnetic coupler of the present invention;
FIG. 5 is a flow chart of a highly integrated magnetic coupler design of the present invention;
FIG. 6 is a schematic plan view of a highly integrated magnetic coupler of the present invention;
FIG. 7 shows a mutual inductance M of the present invention p1s A profile from perfect alignment to 180mm (40%) offset;
FIG. 8 is a diagram of the coil structure definition of a highly integrated magnetic coupler of the present invention;
FIG. 9 is a schematic diagram of a magnetic flux density analysis according to the present invention;
FIG. 10 is a schematic view of the angle definition of the present invention;
FIG. 11 shows the equivalent mutual inductance difference M under deflection according to the present invention ps Is a change curve of (2);
FIG. 12 shows the number of turns n of the present invention w5 And length l 5 An affected secondary side inductor parameter;
FIG. 13 is a diagram of offset M according to the present invention psf 、M ps And a variation graph of epsilon;
FIG. 14 is a primary side integrated inductor design of the present invention;
FIG. 15 is a graph of system output power and efficiency for the offset case of the present invention;
the magnetic resonance circuit comprises a 1-receiving coil, a 2-transmitting coil, a 3-second shielding plate, a 4-primary side integrated induction coil, a 5-integrated reverse coil, a 6-secondary side integrated induction coil, a 7-second magnetic core plate, an 8-first magnetic core plate and a 9-first shielding plate.
Detailed Description
The following description of the embodiments of the present invention will be made clearly and completely with reference to the accompanying drawings, in which it is apparent that the embodiments described are only some embodiments of the present invention, but not all embodiments. All other embodiments, which can be made by those skilled in the art based on the embodiments of the invention without making any inventive effort, are intended to be within the scope of the invention.
The invention provides an IPT system with a highly integrated magnetic coupler and an IPT system integration method, wherein the IPT system with the highly integrated magnetic coupler comprises an inverter, the highly integrated magnetic coupler, a rectifier and a load.
In one embodiment, the highly integrated magnetic coupler structure according to the present invention is shown in fig. 1, and includes a transmitting end and a receiving end; the transmitting end comprises a transmitting coil 2, an integrated reverse coil 5, a primary side integrated inductance coil 4, a first magnetic core plate 8 and a first shielding plate 9; the first magnetic core plate 8 is arranged on the first shielding plate 9, the transmitting coil 2 is arranged on the first magnetic core plate 8, the integrated reverse coil 5 is arranged inside the transmitting coil 2, and the integrated reverse coil 5 and the transmitting coil 2 are of a central symmetrical structure;
the receiving end comprises a receiving coil 1, a secondary side integrated inductance coil 6, a second magnetic core plate 7 and a second shielding plate 3; the second magnetic core plate 7 is placed on the second shield plate 3, the receiving coil 1 is disposed on the second magnetic core plate 7, and the secondary side integrated inductance coil 6 is disposed inside the receiving coil 1.
Wherein, first magnetic core board and second magnetic core board are all used for reinforcing coupling, and first shield plate and second shield plate are all in order to reduce electromagnetic leakage. The highly integrated magnetic coupler provided by the invention aims to improve the anti-offset performance in the x direction and the y direction and realize the complete integration of an LCC compensation IPT system.
In one embodiment, the circuit topology of an IPT system based on a highly integrated magnetic coupler is shown in fig. 2, U dc Is the voltage of a direct current power supply, and the voltage feed inverter consists of MOSFETs S1-S4, C o R is filter capacitance L Is an equivalent load, the rectifier is composed of diodes D1-D4, I in For inverter current vector, I p To transmit the coil current vector, I s To receive the coil current vector, I out Is an output current vector; the system operates at a frequency f and at an angular frequency ω=2pi f.
As shown by fig. 2, the circuit topology of the highly integrated magnetic coupler includes a primary side LCC topology compensation network and a secondary side LCC topology compensation network; the primary side LCC topology compensation network comprises a transmitting coil L p1 Integrated reverse coil L p2 Primary side integrated inductor L pf Capacitance C p And capacitor C pf The method comprises the steps of carrying out a first treatment on the surface of the The secondary side LCC topology compensation network comprises a receiving coil L s Secondary side integrated inductor L sf Capacitance C s And capacitor C sf ；
The primary side integrated inductorCoil L pf Capacitance C p Transmitting coil L p1 And an integrated reverse coil L p2 In series, capacitor C p Transmitting coil L p1 And an integrated reverse coil L p2 Circuit and capacitor C pf Parallel connection; wherein the coil L is transmitted p1 And an integrated reverse coil L p2 Is reversely connected in series; the secondary side is integrated with the inductance coil L sf Capacitance C s And a receiving coil L s In series, capacitor C s And a receiving coil L s Circuit and capacitor C sf Parallel connection;
transmitting coil L p1 Integrated reverse coil L p2 And receiving coil L s The mutual inductance difference between the two is the first mutual inductance M ps The method comprises the steps of carrying out a first treatment on the surface of the Transmitting coil L p1 Integrated reverse coil L p2 With secondary side collective inductance coil L sf The mutual inductance difference between the two is the second mutual inductance M psf The method comprises the steps of carrying out a first treatment on the surface of the Transmitting coil L p1 And an integrated reverse coil L p2 The mutual inductance between the two coils is the first same-side internal mutual inductance M p12 The method comprises the steps of carrying out a first treatment on the surface of the Secondary side collective inductance coil L sf And a receiving coil L s The mutual inductance between the two coils is the second same-side internal mutual inductance M ssf 。
In this embodiment, for the sake of brevity and simplification of the calculation process, the first mutual inductance and the second mutual inductance are defined as:
M ps ＝M p1s -M p2s (1)
M psf ＝M p1sf -M p2sf (2)
wherein ,Mp1s For transmitting coil L p1 And a receiving coil L s Mutual inductance between M p2s For integrating the counter-coil L p2 And a receiving coil L s Mutual inductance between M p1sf For transmitting coil L p1 And a secondary side integrated inductance coil L sf Mutual inductance between M p2sf For integrating the counter-coil L p2 And a secondary side integrated inductance coil L sf Mutual inductance between them.
Specifically, the second same side inner mutual inductance M ssf May be equivalently a T-network, as shown in FIG. 3, M ssf As secondary side LCC topology compensation networkA part is integrated together and decoupled by the following formulae (3) - (5):
Figure BDA0004088891560000091
L sfe ＝L sf +M ssf (4)
Figure BDA0004088891560000092
wherein ,Lse Representing a receiving coil L s And a series compensation capacitor C s Equivalent value of L sfe Representing the secondary side integrated inductance L sf And mutual inductance M ssf Is equivalent to the superposition of C sfe Representing the secondary side parallel compensation capacitance C sf And mutual inductance M ssf Is equivalent to the superposition of (a).
In addition, M is also the mutual inductance inside the same side p12 Decoupling can be achieved by the following equation (6):
Figure BDA0004088891560000093
wherein ,Lp Representing the primary side transmitting coil L p1 Primary side reverse coil L p2 And primary side series compensation capacitor C p Is equivalent to the superposition of (a).
In one embodiment, an equivalent circuit of the circuit topology of the IPT system is obtained based on a Fundamental Harmonic Approximation (FHA) method, as shown in fig. 4, the square wave voltage is approximately a sinusoidal source U1, the rectifier and the resistive load together are equivalent to R e ＝8R L /pi 2. The coupling in the circuit is represented by the relevant source. In this example the theoretical feasibility of the proposed method is focused on, ignoring the power loss of the element.
Based on the equivalent circuit, the method is obtained according to kirchhoff's law:
Figure BDA0004088891560000101
/>
meanwhile, the resonance relationship is expressed as:
ω 2 L pf C pf ＝ω 2 L p C pf ＝ω 2 L se C sfe ＝ω 2 L sfe C sfe ＝1 (8)
wherein ,
Figure BDA0004088891560000102
representing the inverted output current, ">
Figure BDA0004088891560000103
Indicating the current through the transmitting coil, < >>
Figure BDA0004088891560000104
Indicating the current through the receiving coil, < > and->
Figure BDA0004088891560000105
Representing the output current +.>
Figure BDA0004088891560000106
Represents the inversion output voltage, R e Representing the load that is commonly equivalent to the rectifier and the load.
The kirchhoff law formula is simplified through a resonance relation formula, and the method comprises the following steps of:
Figure BDA0004088891560000107
derived from the simplified formula:
Figure BDA0004088891560000108
Figure BDA0004088891560000109
Figure BDA00040888915600001010
Figure BDA00040888915600001011
Figure BDA0004088891560000117
Figure BDA0004088891560000111
wherein ,
Figure BDA0004088891560000112
representing the inverted output voltage, ">
Figure BDA0004088891560000113
Modulo representing inverted output voltage +.>
Figure BDA0004088891560000114
Mode representing the current of the transmitting coil,/->
Figure BDA0004088891560000115
Mode, P, representing output current out Represents output power, K 1 Representing coefficients defined by system parameters, Z in Representing the total equivalent impedance of the system from the input.
From the derived current correlation formulas (10), (11), I p and Iout The system has constant current output characteristics irrespective of the load. From the formulas (12) and (13), K 1 Depending on the system parameters of the design, the system will not change in the case of offset, so the anti-offset performance of the system depends on M ps 2 . (12) The mathematical derivation of the equation is compatible with a single coupled energy transfer mode, and the nature of the power transfer is unchanged. For general single sheetsCoupling IPT systems, the ability to improve anti-migration through the design of the coupling mechanism is limited. In contrast, mutual inductance difference M ps It is expected to achieve stable output power in the case of offset. The integrated back coil is designed in detail herein to optimize M ps Is stable.
In particular, from (14) the calculated inverter current and (15) the calculated IPT system input impedance, it can be seen that M psf The output power is not affected but the imaginary part of the total input impedance of the system is affected. In this embodiment, a phase angle between the inverter voltage and the inverter current is defined as α, positive and negative values thereof represent inductance and capacitance, respectively, and an absolute value represents a degree to which the system deviates from a resonance point; definition M psf And M is as follows ps Where epsilon is the ratio of alpha, the tangent value of alpha is expressed as:
Figure BDA0004088891560000116
according to (16), by M psf The resonance state of the IPT system can be quantified. Under normal parametric configurations, the resonance state may be limited by the design of the magnetic coupler. Epsilon can be designed to be small to ensure normal output power and ZVS operating conditions.
In one embodiment, based on the above theory, an IPT system anti-offset parameter optimization method based on a highly integrated magnetic coupler is provided, wherein an initial magnetic coupling mechanism is constructed as a design background, and then an integrated reverse coil is set to determine a variable M ps To improve the tolerance of the offset; setting a secondary side integrated inductance coil to determine epsilon for configuring ZVS conditions; finally, a primary side integrated inductance coil is arranged to realize decoupling. In this embodiment, the highly integrated magnetic coupler structure modeled using MAXWELL is shown in fig. 6, and parameters describing the dimensions of the highly integrated magnetic coupler are defined in table 1:
table 1 parameters of highly integrated magnetic coupler
Figure BDA0004088891560000121
In one embodiment, the parameter optimization design method is explained with a typical example of an IPT system suitable for a power level of 3kW, and the specific process is shown in fig. 5, comprising:
s1, constructing an initial magnetic coupler according to the working frequency, the input voltage and the output power of an IPT system; the initial magnetic coupler comprises a transmitting end and a receiving end; the transmitting end is provided with a transmitting coil L p1 The receiving end is provided with a receiving coil L s The method comprises the steps of carrying out a first treatment on the surface of the Transmitting coil L p1 And a receiving coil L s The number of turns and the size of the coil are preset, and the transmitting coil L is defined p1 And a receiving coil L s The mutual inductance between them is M p1s ；
Specifically, the transmitting coil L p1 Is 450mm by 5mm, and has a number of turns n w1 10; receiving coil L s Is 300mm by 5mm, and has a number of turns n w4 12, the magnetic core plate is a ferrite plate for enhancing a magnetic field, the shielding plate is an aluminum shielding plate for electromagnetic shielding, and the transmission distance d=150 mm between the transmitting end and the receiving end.
FIG. 7 shows mutual inductance M p1s The profile from alignment to 180mm (40%) offset. From the well aligned case to the 40% offset case, the mutual inductance drops by nearly 40%. For IPT systems with single mutual inductance for power transfer, it is difficult to substantially improve anti-offset performance through the parametric design of the magnetic coupling mechanism. The invention thus adds an integrated back-winding on the primary side to alter the coupling of the power transfer, thereby improving the anti-offset performance.
S2, arranging an integrated reverse coil L at a transmitting end of the initial magnetic coupler p2 And determines the integrated reverse coil L according to the output power of the IPT system p2 Turns of (2); definition of the transmitting coil L p1 Integrated reverse coil L p2 And receiving coil L s The mutual inductance difference between the two is the first mutual inductance M ps ；
S3, performing finite element analysis through MAXWELL, traversing the integrated reverse coil L p2 Determine the size of the first mutual inductance M ps Whether the alignment deviation is stabilized within 40%, if so, the step S4 is entered, otherwise, the process is continuedTraversing;
the reverse coil is integrated to improve coupling performance between the transmit coil and the receive coil. Thus, the coupling characteristics between the integrated reverse coil and the receiving coil are expected to be substantially the same as the coupling characteristics between the transmitting coil and the receiving coil. In this way, the equivalent mutual inductance obtained by superposition of the two coupling mechanisms can be kept stable enough under the offset condition. Transmitting coil L p1 Integrated reverse coil L p2 And receiving coil L s Mutual inductance difference M between ps The variation of (2) determines the anti-offset performance of the IPT system. The position, size and number of turns of the integrated counter coil play a critical role when the main coil is fixed. Thus, to obtain a more stable M ps The integrated back coil needs to be carefully designed.
Specifically, as shown in fig. 8, the integrated reverse coils are designed inside the transmit coil, they are center symmetrical, as shown in fig. 8 (a). Each side of the coil with multiple turns is equivalently the center position for analysis. The corresponding parameters are marked in FIG. 8 (b), the four sides of the transmit coil are l ai (i=1, 2,3, 4); the four sides of the integrated reversing coil are l bj (j=1, 2,3, 4); the four sides of the receiving coil are l cv (v=1, 2,3, 4). The end points of the coil are marked for definition. Wherein A1, B1, C1 and D1 are four endpoints of the transmitting coil; a2, B2, C2, and D2 are four endpoints of the integrated reverse coil; F. g, H and J are the four end points of the receive coil. Taking into account l c1 and la1 Is parallel, from l c1 Any point up to l a1 Is equal, defined as r.
The magnetic flux density analysis is shown in FIG. 9, l a1 The magnetic field strength generated at point F is:
Figure BDA0004088891560000141
wherein I represents the current values of the transmitting coil and the integrated counter coil,
Figure BDA0004088891560000142
representation squareVector of directions, l 1 Represents the length of the side length of the transmitting coil, theta 1 The angles marked in fig. 9 (a).
Q1 is the distance F to A (referring to A1, A2). Q2 is the superposition of the lengths of lc2 and Q1. In FIG. 9 (a), l c1 The average value of the magnetic flux density is calculated by integration as:
Figure BDA0004088891560000143
wherein ,θ2 The angles marked in fig. 9 (a).
l a1 The magnetic flux generated at the receiving coil Ls is:
Figure BDA0004088891560000144
l a1 and Ls The mutual inductance between the two is as follows:
Figure BDA0004088891560000145
similarly, to obtain mutual inductance between the other side and the receiving coil, the relevant angle is defined in fig. 10, and the same can be said:
Figure BDA0004088891560000151
Figure BDA0004088891560000152
Figure BDA0004088891560000153
Figure BDA0004088891560000154
Figure BDA0004088891560000155
wherein ,μ0 Represents the permeability, θ, of the vacuum 5 、θ 6 、θ 9 、θ 10 、θ 15 、θ 16 、θ 11 、θ 12 、θ 13 and θ14 The angle information of (2) is shown in fig. 10.
Thus, for qualitative analysis, the mutual inductance M1 between the receiving coil and the transmitting coil, and the mutual inductance M2 between it and the integrated counter coil can be expressed as
Figure BDA0004088891560000156
To more intuitively express the characteristic of equivalent mutual inductance, the corresponding mutual inductances are combined:
Figure BDA0004088891560000157
each item includes mutual inductance generated by the transmitting coil and the integrated reversing coil, which are opposite in direction. Further, according to (20), (21) and fig. 8, the following relation can be obtained:
Figure BDA0004088891560000161
θ 2 -θ 1 ＜θ 4 -θ 3 ,θ 8 -θ 7 ＜θ 6 -θ 5 ,
θ 16 -θ 15 ＜θ 14 -θ 13 ,θ 10 -θ 9 ＜θ 12 -θ 11
according to (23) and (24), the integrated reverse coil effectively converts the original single mutual inductance into an equivalent mutual inductance difference, thereby improving the anti-offset performance. In addition, based on the qualitative analysis of the mutual inductance expression under the offset considered in (20) and (21), two features are revealed, which can further guide the design of the integrated back coil. First, the number of turns of the integrated reverse coil has a negative gain effect on the equivalent mutual inductance under offset, with a greater impact. Second, the equivalent side length of the integrated back-coil has a slight positive gain effect on the equivalent mutual inductance during misalignment due to the variation in side length ratio and angle difference in (24). Thus, the number of turns and size can be used as the thickness parameter. Specifically, for the number of turns parameter, the larger the number of turns of the integrated reverse coil, the more stable the equivalent mutual inductance difference, but the more the mutual inductance cancels. For the dimensional parameters, the larger the size of the integrated reverse coil, the more stable the equivalent mutual inductance value, but the more mutual inductance is eliminated. For different turns there is a size optimum corresponding to the best anti-migration performance. The final optimized equivalent mutual inductance value varies with the choice of turns. In the existing reverse coil study, after the number of turns is designed, the corresponding unique dimension optimum is obtained. This requires a compromise between optimal performance and power transfer capability. However, if the number of layers of the integrated reverse coil is increased, the equivalent size can be increased with a fixed number of turns, and the power transmission capability can be improved while achieving the improvement in performance.
In the 3kW power stage prototype of this embodiment, the number of turns of the integrated reverse coil is designed to be 12. To achieve stable M during misalignment ps Side length l 2 Optimization was performed. In FIG. 11 (a), with l 2 The difference in mutual inductance increases in offset cases are more pronounced than in the case of perfect alignment. When l 2 =200mm and l 2 At =220 mm, the mutual inductance difference under offset even exceeds that of perfect alignment. When l 2 At =180 mm, the mutual inductance difference remains stable from good alignment to 40% misalignment, and IPT systems using this mutual inductance difference for power transfer have excellent anti-offset performance. FIG. 11 (b) further shows that when 2 M when=180 mm p1s and Mp2s Is a coupling characteristic of (a). Thus, the integrated reverse coil is sized 180mm by 5mm.
S4, arranging a secondary side integrated inductance coil at the receiving end of the initial magnetic couplerL sf Defining a transmitting coil L p1 Integrated reverse coil L p2 With secondary side collective inductance coil L sf The mutual inductance difference between the two is the second mutual inductance M psf ；
S5, the second mutual inductance M psf With a first mutual inductance M ps The ratio of (2) is defined as epsilon; traversing secondary side integrated inductor L sf Judging whether epsilon is less than or equal to zeta and zeta is a desired susceptibility index; if yes, entering a step S5, otherwise, continuing traversing;
after the integrated back coil is designed, the anti-offset performance is improved due to the poor stable mutual inductance. The integrated inductor on the secondary side can be further designed to optimize ZVS operating conditions. Under the usual 50% duty cycle control strategy, ZVS operating state is closely related to the phase angle α of the inverter output voltage and current. According to fig. 3, the equivalent inductance in the secondary side LCC topology compensation network of the ipt system is L sfe It is L sf and Mssf Is a superposition of (3). Therefore, the key design of the secondary side integrated inductor is L sf and Mssf Is not limited. With n w5 and l5 L is obtained by a traversal method as a variable sf and Mssf As shown in fig. 12. The result shows that under the same parameters, the internal mutual inductance M ssf Proximity self-inductance L sf And (5) configuration. In realizing smaller self-inductance L sf At the same time, the required equivalent self-inductance L can be obtained sfe . According to (16), the size of the secondary side integrated inductor coil is expected to be sufficiently small. In a 3kW power prototype, the secondary side integrated inductor is sized 140mm x 5mm. Number of turns n w4 Designed as 4.
Under the designed secondary side integrated inductance coil, M psf 、M ps And the variation of the ratio epsilon upon shifting is shown in fig. 13. The results show that epsilon remains at a minimum above 0 and a maximum within 0.1. The ZVS working state can be ensured, and the turn-off current can not be too large.
S6, arranging a primary side integrated inductance coil L at the transmitting end of the initial magnetic coupler pf Traversing according to output power of transmitting end of initial magnetic couplerPrimary side integrated inductor L pf And the number and size of turns in (a) to yield a final optimized highly integrated magnetic coupler.
Specifically, the required primary side (primary side) circuit impedance is determined according to the total power level (including voltage and current) output by the transmitting end, so as to determine the required integrated coil inductance value, and the primary side integrated inductor L is traversed through simulation software pf The number of turns and the size of (a) are such that the self-inductance value is close to the desired value.
With a view to improving the compactness of the coupler, the inductor in the primary side LCC topology compensation network is implemented by one coupling coil without affecting the system characteristics. The primary side integrated inductor is designed as a special bipolar coil in consideration of adaptability to x-direction and y-direction offset, as shown in fig. 14 (a). Unlike a typical bipolar structure, the bipolar coil proposed herein has eight current loops. Wherein the polarities of two adjacent current loops are opposite. The transmit coil and the integrated reverse coil are monopolar. Due to the central symmetry, the primary side integrated inductor coils are naturally decoupled from them.
The design of the primary side integrated inductor is represented by l 3 、l 4 and nw3 And (5) determining. Different n w3 and l3 l corresponds to primary side integrated inductance, and in selecting l 4 In the case of=60 mm, it is obtained by a traversal method, as shown in fig. 14. The size of the primary side integrated inductor can be designed in a 3kW power level IPT system according to the required inductance value. n is n w3 Designed as 3,l 3 Designed to be 200mm.
In one embodiment, experiments were performed using the highly integrated magnetic coupler described above with an IPT system suitable for a 3kW power level. At the input, a dc power supply and a high power inverter are used to provide 300V ac excitation to the resonant circuit. The inverter adopts CREE silicon carbide MOSFET (C2M 0025120D) with the internal resistance of 20mΩ, so that the power loss is reduced, and the output stability is improved. The PWM control signal for the MOSFET is generated by the control chip DSP 28335. The output is a fixed frequency of 85 kHz. At the output, the CREE C3D20060D diode is used for a rectifier to provide dc current to the EA-CPS-8080 electronic load. The output resistance was set to 30Ω. The coil was made of 1000 strands of AWG 38 litz wire. The magnetic material PC95 is used to construct the ferrite plate. The measured parameters and calculated resonance parameters of the coupling mechanism are shown in table 2:
table 2 measurement parameters of highly integrated magnetic coupler
Figure BDA0004088891560000181
Figure BDA0004088891560000191
/>
Two points are revealed from the experimental results. First, the system works effectively with good alignment and different offset distances. Second, the inverter remains operating in an ideal micro-inductive load state, and the secondary side integrated coil effectively optimizes ZVS conditions.
Fig. 15 gives the output power and the DC-DC overall system efficiency for an IPT system for an offset when rl=30Ω. The output power ranges from 2.9kw to 3.1kw over an offset range of 18 cm. The average output power was 3kW. Under the working condition of 18cm offset, the output power was reduced to 2.9kW, which is 97% of the well aligned value (3.1 kW). Under the same conditions, the overall efficiency of the IPT system can still be maintained above 93%. The results verify that the proposed method has good anti-migration performance.
Although embodiments of the present invention have been shown and described, it will be understood by those skilled in the art that various changes, modifications, substitutions and alterations can be made therein without departing from the principles and spirit of the invention, the scope of which is defined in the appended claims and their equivalents.

Claims (9)

1. An IPT system having a highly integrated magnetic coupler, the IPT system comprising an inverter, a highly integrated magnetic coupler, a rectifier and a load; the highly integrated magnetic coupler comprises a transmitting end and a receiving end; the transmitting end comprises a transmitting coil L p1 Integrated reverse coil L p2 Primary side integrated circuitInductance coil L pf A first magnetic core plate and a first shield plate; a first magnetic core plate is arranged on the first shielding plate, and a transmitting coil L p1 Is arranged on a first magnetic core plate and integrates a reverse coil L p2 Is arranged on the transmitting coil L p1 Inside, and integrate the reverse coil L p2 And a transmitting coil L p1 Is of a central symmetrical structure;
the receiving end comprises a receiving coil L s Secondary side integrated inductor L sf A second magnetic core plate and a second shield plate; a second magnetic core plate is arranged on the second shielding plate and is used for receiving the coil L s Is arranged on the second magnetic core plate, and the secondary side is integrated with an inductance coil L sf Is arranged on the receiving coil L s Inside.
2. An IPT system with a highly integrated magnetic coupler as claimed in claim 1 wherein the primary side integrates an inductor L pf Is a double-stage coil, a transmitting coil L p1 And an integrated reverse coil L p2 4 double-stage coils are arranged between the two coils; the 1 double-stage coil has 2 current loops, the 4 double-stage coils have 8 current loops, and the polarities of every two current loops are opposite.
3. An IPT system with a highly integrated magnetic coupler as claimed in claim 1 wherein the inverter is a voltage fed inverter consisting of MOSFET S1, MOSFET S2, MOSFET S3 and MOSFET S4; the rectifier includes a diode D1, a diode D2, a diode D3, and a diode D4.
4. An IPT system integration method with a highly integrated magnetic coupler, comprising the steps of:
s1, constructing an initial magnetic coupler according to the working frequency, the input voltage and the output power of an IPT system; the initial magnetic coupler comprises a transmitting end and a receiving end; the transmitting end is provided with a transmitting coil L p1 The receiving end is provided with a receiving coil L s The method comprises the steps of carrying out a first treatment on the surface of the Transmitting coil L p1 And a receiving coil L s The number of turns and the size of the coil are preset and definedTransmitting coil L p1 And a receiving coil L s The mutual inductance between them is M p1s ；
S2, arranging an integrated reverse coil L at a transmitting end of the initial magnetic coupler p2 And determines the integrated reverse coil L according to the output power of the IPT system p2 Turns of (2); definition of the transmitting coil L p1 Integrated reverse coil L p2 And receiving coil L s The mutual inductance difference between the two is the first mutual inductance M ps ；
S3, performing finite element analysis through MAXWELL, traversing the integrated reverse coil L p2 Determine the size of the first mutual inductance M ps If the alignment deviation is stable within 40%, the step S4 is entered if yes, otherwise, the traversal is continued;
s4, arranging a secondary side integrated inductance coil L at the receiving end of the initial magnetic coupler sf Defining a transmitting coil L p1 Integrated reverse coil L p2 With secondary side collective inductance coil L sf The mutual inductance difference between the two is the second mutual inductance M psf ；
S5, the second mutual inductance M psf With a first mutual inductance M ps The ratio of (2) is defined as epsilon; traversing secondary side integrated inductor L sf Judging whether epsilon is less than or equal to zeta and zeta is a desired susceptibility index; if yes, entering a step S5, otherwise, continuing traversing;
s6, arranging a primary side integrated inductance coil L at the transmitting end of the initial magnetic coupler pf Traversing primary side integrated inductor L based on output power of transmitting end of initial magnetic coupler pf And the number and size of turns in (a) to yield a final optimized highly integrated magnetic coupler.
5. An IPT system integration method with a highly integrated magnetic coupler as claimed in claim 4 wherein the circuit topology of the IPT system comprises a primary side LCC topology compensation network and a secondary side LCC topology compensation network; the primary side LCC topology compensation network comprises a transmitting coil L p1 Integrated reverse coil L p2 Primary side integrated inductor L pf Capacitance C p And capacitor C pf The method comprises the steps of carrying out a first treatment on the surface of the The secondary side LCC topology complementThe compensation network comprises a receiving coil L s Secondary side integrated inductor L sf Capacitance C s And capacitor C sf ；
The primary side integrates an inductance coil L pf Capacitance C p Transmitting coil L p1 And an integrated reverse coil L p2 In series, capacitor C p Transmitting coil L p1 And an integrated reverse coil L p2 Circuit and capacitor C pf Parallel connection; wherein the coil L is transmitted p1 And an integrated reverse coil L p2 Is reversely connected in series; the secondary side is integrated with the inductance coil L sf Capacitance C s And a receiving coil L s In series, capacitor C s And a receiving coil L s Circuit and capacitor C sf Parallel connection;
transmitting coil L p1 Integrated reverse coil L p2 And receiving coil L s The mutual inductance difference between the two is the first mutual inductance M ps The method comprises the steps of carrying out a first treatment on the surface of the Transmitting coil L p1 Integrated reverse coil L p2 With secondary side collective inductance coil L sf The mutual inductance difference between the two is the second mutual inductance M psf The method comprises the steps of carrying out a first treatment on the surface of the Transmitting coil L p1 And an integrated reverse coil L p2 The mutual inductance between the two coils is the first same-side internal mutual inductance M p12 The method comprises the steps of carrying out a first treatment on the surface of the Secondary side collective inductance coil L sf And a receiving coil L s The mutual inductance between the two coils is the second same-side internal mutual inductance M ssf 。
6. An IPT system integration method with a highly integrated magnetic coupler as claimed in claim 5 wherein the second common side internal mutual inductance M ssf The equivalent is a T-shaped network, and is integrated with a secondary side LCC topology compensation network, and is decoupled by the following formula:
Figure FDA0004088891550000031
L sfe ＝L sf +M ssf
Figure FDA0004088891550000032
first same side internal mutual inductance M p12 The decoupling formula of (2) is expressed as:
Figure FDA0004088891550000033
wherein ,Lse Representing a receiving coil L s And a series compensation capacitor C s And ω represents the system resonant angular frequency, L sfe Representing a secondary side integrated inductor L sf And a second same-side internal mutual inductance M ssf Is equivalent to the superposition of C sfe Representing the secondary side parallel compensation capacitance C sf And a second same-side internal mutual inductance M ssf Is equivalent to the superposition of L p Representing the transmitting coil L p1 Integrated reverse coil L p2 And primary side series compensation capacitor C p Is equivalent to the superposition of (a).
7. The method of claim 6, wherein the equivalent circuit of the circuit topology package of the IPT system is obtained by using a basic harmonic approximation method, and is obtained by kirchhoff's law:
Figure FDA0004088891550000041
the resonance relationship is expressed as:
ω 2 L pf C pf ＝ω 2 L p C pf ＝ω 2 L se C sfe ＝ω 2 L sfe C sfe ＝1
wherein ,
Figure FDA0004088891550000042
representing the inverted output current, ">
Figure FDA0004088891550000043
Indicating the current through the transmitting coil, < >>
Figure FDA0004088891550000044
Indicating the current through the receiving coil, < > and->
Figure FDA0004088891550000045
Representing the output current +.>
Figure FDA0004088891550000046
Represents the inversion output voltage, R e Representing the load that is commonly equivalent to the rectifier bridge and the load.
8. An IPT system integration method with a highly integrated magnetic coupler as claimed in claim 7 wherein the kirchhoff's law equation is simplified by a resonant relationship to obtain:
Figure FDA0004088891550000047
derived from the simplified formula:
Figure FDA0004088891550000048
Figure FDA0004088891550000049
Figure FDA00040888915500000410
Figure FDA00040888915500000411
Figure FDA0004088891550000051
Figure FDA0004088891550000052
wherein ,
Figure FDA0004088891550000053
representing the inverted output voltage, ">
Figure FDA0004088891550000054
Modulo representing inverted output voltage +.>
Figure FDA0004088891550000055
A mode representing the current of the transmit coil,
Figure FDA0004088891550000056
mode, P, representing output current out Represents output power, K 1 Representing coefficients defined by system parameters, Z in Representing the total equivalent impedance of the system from the input.
9. An IPT system integration method with a highly integrated magnetic coupler as claimed in claim 8 wherein the derivation of the reduced formula reveals M psf The output power is not affected, but the imaginary part of the total input impedance of the IPT system is affected; defining a phase angle between the inverter voltage and the inverter current as alpha, wherein positive and negative values respectively represent the sensitivity and the capacitance, and an absolute value represents the degree of deviation of the IPT system from a resonance point; by M psf Quantifying the resonance state of IPT system, expressed as:
Figure FDA0004088891550000057
/>
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