CN116054656A - Method, system and medium for suppressing harmonic disturbance of counter electromotive force of permanent magnet synchronous motor - Google Patents

Method, system and medium for suppressing harmonic disturbance of counter electromotive force of permanent magnet synchronous motor Download PDF

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CN116054656A
CN116054656A CN202211634322.2A CN202211634322A CN116054656A CN 116054656 A CN116054656 A CN 116054656A CN 202211634322 A CN202211634322 A CN 202211634322A CN 116054656 A CN116054656 A CN 116054656A
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electromotive force
back electromotive
estimated value
permanent magnet
synchronous motor
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吴轩
王超
吴婷
余旭
黄守道
黄晓辉
吕铭晟
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Hunan University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/50—Reduction of harmonics

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Abstract

The invention discloses a method, a system and a medium for suppressing the harmonic disturbance of counter electromotive force of a permanent magnet synchronous motor, wherein the method for suppressing the harmonic disturbance of the counter electromotive force of the permanent magnet synchronous motor comprises the steps of acquiring an estimated value of the counter electromotive force by adopting a sliding mode algorithm based on working voltage and current of the permanent magnet synchronous motor, wherein the sliding mode algorithm adopts a preset sliding mode switching function to reduce high-frequency buffeting of the counter electromotive force estimation; filtering the estimated value of the back electromotive force to obtain a fundamental component of the back electromotive force; the estimation of rotor position angle and speed is performed by a phase-locked loop based on the fundamental component of back emf. The invention aims to inhibit harmonic interference of back electromotive force estimation, improve estimation accuracy of rotor position angle and speed, and is independent of a specific rotor position estimation method, and can be matched with the existing rotor position estimation method.

Description

Method, system and medium for suppressing harmonic disturbance of counter electromotive force of permanent magnet synchronous motor
Technical Field
The invention relates to the technical field of motor control, in particular to a method, a system and a medium for suppressing harmonic disturbance of back electromotive force of a permanent magnet synchronous motor.
Background
The permanent magnet synchronous motor has the advantages of high power density, high efficiency, quick torque response, small volume and the like, and is widely applied to industries such as industrial equipment manufacturing, aerospace, automobiles, processing and manufacturing and the like in addition to development of permanent magnet materials, high-performance processors and power electronic components. In the control system of the permanent magnet synchronous motor, the acquisition of the rotor position angle and the rotor speed is indispensable, and the installation of the traditional mechanical sensor can bring great disadvantages, and even some special occasions do not allow the installation of the mechanical sensor. Therefore, the position and speed information of the rotor can be estimated through sampling the electric signals (such as voltage and current) of the motor during operation and through a specific mathematical model, and the method is generally called as a sensorless technology of the permanent magnet synchronous motor.
In the position estimation method based on the permanent magnet synchronous motor model, the sliding mode algorithm is simple, robustness to interference and quick in dynamic response, the method is commonly used for estimating the counter electromotive force of the permanent magnet synchronous motor, and further rotor position angle and speed information are extracted from the counter electromotive force through a phase-locked loop. However, the back electromotive force estimated by the sliding mode contains larger 5 th and 7 th harmonic components in addition to the fundamental wave due to the nonlinearity of the inverter and the interference of the spatial magnetic field harmonic, and thus 6 th harmonic pulsation is introduced into the estimated rotor position. The 6 pulses of the position estimation can cause inaccurate magnetic field decoupling of motor control, increase extra energy loss of the motor, and deteriorate the performance of the control without the position sensor. Since the direct source of the estimated rotor position harmonic error is harmonic interference of the back emf, it is necessary to filter the back emf. Common back emf filtering methods are multilayer bandpass filters, such as multilayer generalized second order integrators, multilayer proportional resonator filters, etc. The method adopts a multi-layer parallel structure, parameters of each layer are mutually influenced, parameter adjustment is difficult, and discretization is not easy. There are also documents that propose filtering structures based on adaptive learning mechanisms, but to some extent increase the computational complexity.
Disclosure of Invention
The invention aims to solve the technical problems: aiming at the problems in the prior art, the invention provides a method, a system and a medium for suppressing harmonic disturbance of counter electromotive force of a permanent magnet synchronous motor, which aim to suppress the harmonic disturbance of the counter electromotive force, improve the estimation precision of rotor position angle and speed, and do not depend on a specific rotor position estimation method, and can be matched with the existing rotor position estimation method.
In order to solve the technical problems, the invention adopts the following technical scheme:
a method for suppressing harmonic disturbance of back electromotive force of a permanent magnet synchronous motor comprises the following steps:
s101, acquiring an estimated value of a back electromotive force by adopting a sliding mode algorithm based on the working voltage and current of a permanent magnet synchronous motor, wherein the sliding mode algorithm adopts a preset sliding mode switching function to reduce high-frequency buffeting of back electromotive force estimation;
s102, filtering the estimated value of the back electromotive force to obtain a fundamental component of the back electromotive force;
s103, estimation of rotor position angle and speed is performed by a phase-locked loop based on the fundamental component of back electromotive force.
Optionally, step S101 includes:
s201, three-phase current of the permanent magnet synchronous motor obtained through current sensor sampling is subjected to coordinate transformation to obtain two-phase current under a static two-phase coordinate system;
s202, taking two-phase current and voltage under a static two-phase coordinate system as input quantities, constructing a sliding mode observer based on a back electromotive force equation, selecting a preset sliding mode switching function, wherein the selected sliding mode surface S is the difference value between the two-phase current under the static two-phase coordinate system and the observed value thereof, and obtaining an estimated value of the back electromotive force under the static two-phase coordinate system when the sliding mode converges.
Optionally, the preset sliding mode switching function is a hyperbolic tangent function.
Optionally, the functional expression of the back emf equation in step S202 is:
Figure BDA0004006999220000021
Figure BDA0004006999220000022
in the above, u α And u β Is the working voltage of the permanent magnet synchronous motor, R s Is the resistance of the permanent magnet synchronous motor, L d And L q Respectively the d-axis inductance and the q-axis inductance of the permanent magnet synchronous motor, p is a differential operator, omega e Is the rotor rotating speed, i of the permanent magnet synchronous motor α And i β E is the two-phase current in the stationary two-phase coordinate system α And e β Is the back electromotive force psi in a static two-phase coordinate system f Permanent magnet flux linkage of permanent magnet synchronous motor, theta e Is the rotor angle.
Optionally, filtering the estimated value of the back electromotive force in step S102 means filtering out 5 th order and 7 th order harmonic components in the estimated value of the back electromotive force to obtain a fundamental component of the back electromotive force.
Optionally, a filter used for filtering the estimated value of the counter electromotive force in step S102 is a frequency adaptive comb filter, a center frequency of the frequency adaptive comb filter is the same as a motor rotor rotation speed of the permanent magnet synchronous motor, and the motor rotor rotation speed of the permanent magnet synchronous motor is fed back to the frequency adaptive comb filter in real time through a phase-locked loop to achieve frequency adaptation.
Optionally, the transfer function G(s) of the frequency adaptive comb filter has a functional expression:
Figure BDA0004006999220000023
in the above-mentioned method, the step of,
Figure BDA0004006999220000024
is the fundamental component of back EMF in a stationary two-phase coordinate system, z αβ Is the estimated value of the back electromotive force under the static two-phase coordinate system, s is an integral operator, e is a constant, j is an imaginary unit,>
Figure BDA0004006999220000025
is an estimated value of the rotor rotation speed period, and comprises: />
Figure BDA0004006999220000026
Wherein->
Figure BDA0004006999220000027
The rotor speed of the frequency adaptive comb filter is fed back to the phase-locked loop in real time.
Optionally, the frequency adaptive comb filter includes:
discrete delay module for respectively inputting estimated value z of alpha-axis back electromotive force α Estimated value z of beta-axis back electromotive force β Delay time
Figure BDA0004006999220000031
And discrete delay sampling is carried out by adopting a linear interpolation calculation method to obtain an estimated value z of alpha-axis back electromotive force after delay αd Estimated value z of beta-axis back electromotive force βd ；
An angle rotation module for estimating z of the alpha-axis back electromotive force after delay αd Estimated value z of beta-axis back electromotive force βd Multiplying a Park inverse transformation constant coefficient matrix to carry out Park inverse transformation to obtain inverse motorPotential estimation value
Figure BDA0004006999220000032
And->
Figure BDA0004006999220000033
A summation module for adding the estimated value z of the alpha-axis back electromotive force α Its back electromotive force estimated value obtained by angle rotation module
Figure BDA0004006999220000034
Adding, and multiplying by 1/2 to obtain back EMF fundamental wave +.>
Figure BDA00040069992200000311
Estimate z of the beta-axis back EMF to be input β And its back electromotive force estimated value +.>
Figure BDA0004006999220000036
Adding, and multiplying by 1/2 to obtain back electromotive force fundamental wave of beta axis +.>
Figure BDA0004006999220000037
Optionally, the function expression of discrete delay sampling by adopting a linear interpolation calculation method is as follows:
z αd [k-N]＝f r ·z α [k-N c ]+(1-f r )·z α [k-N f ]
z βd [k-N]＝f r ·z β [k-N c ]+(1-f r )·z β [k-N f ]
in the above, z ad [k-N]、z βd [k-N]As the back electromotive force estimated value z α And back electromotive force estimation value z β The value, z, that has undergone linear interpolation processing at the kth-N sampling instants α [k-N c ]、z β [k-N c ]Back emf estimate z for the alpha axis α And a back emf estimate z of the beta axis β In the k-N c The value of each sampling instant, z α [k-N f ]、z β [k-N f ]Back emf estimate z for the alpha axis α And a back emf estimate z of the beta axis β In the k-N f The value of each sampling moment, N is the discrete sampling length, can be the decimal, f r ＝N-N f ，N f ＝floor(N)，N c ＝ceil(N)＝N f +1; wherein:
Figure BDA0004006999220000038
in the above formula, N is a discrete sampling length,
Figure BDA0004006999220000039
for the estimated value of the rotor rotation speed period, T s For the sampling period +.>
Figure BDA00040069992200000310
The rotor speed of the frequency adaptive comb filter is fed back to the phase-locked loop in real time.
Optionally, step S103 includes:
s301, carrying out per unit processing on fundamental wave components of back electromotive force under a static two-phase coordinate system to respectively obtain a fundamental wave component per unit value of alpha-axis back electromotive force and a fundamental wave component per unit value of beta-axis back electromotive force;
s302, multiplying the per unit value of the alpha-axis back electromotive force fundamental wave component by the cosine value of the rotor position angle currently output to be used as a first product term, multiplying the per unit value of the beta-axis back electromotive force fundamental wave component by the sine value of the rotor position angle currently output to be used as a second product term, multiplying the two product terms by-1 respectively, and adding to obtain a rotor position angle error;
s303, the rotor position angle error is passed through a PI regulator to obtain a rotor speed estimated value, and the rotor speed estimated value is subjected to integral calculation to obtain an estimated rotor position angle.
In addition, the invention also provides a system for suppressing the back electromotive force harmonic disturbance of the permanent magnet synchronous motor, which comprises a microprocessor and a memory which are connected with each other, wherein the microprocessor is programmed or configured to execute the method for suppressing the back electromotive force harmonic disturbance of the permanent magnet synchronous motor.
Furthermore, the present invention provides a computer readable storage medium having stored therein a computer program for being programmed or configured by a microprocessor to perform the method of back EMF harmonic disturbance suppression of a permanent magnet synchronous motor.
Compared with the prior art, the invention has the following advantages:
1. the method comprises the steps of obtaining an estimated value of counter electromotive force by adopting a sliding mode algorithm based on working voltage and current of a permanent magnet synchronous motor, wherein the sliding mode algorithm adopts a preset sliding mode switching function to reduce high-frequency buffeting of counter electromotive force estimation; the estimated value of the counter electromotive force is filtered to obtain the fundamental component of the counter electromotive force, the rotor position angle and the speed are estimated through the phase-locked loop based on the fundamental component of the counter electromotive force, the high-frequency buffeting and the harmonic component of the counter electromotive force can be eliminated, and the estimation precision of the rotor position angle and the speed can be effectively improved.
2. The invention comprises the estimation of rotor position angle and speed by a phase-locked loop based on the fundamental component of back electromotive force, and can be matched with the existing rotor position estimation method without depending on a specific rotor position estimation method.
Drawings
FIG. 1 is a schematic diagram of a basic flow of a method according to an embodiment of the present invention.
Fig. 2 is a schematic diagram of a control principle of a method according to an embodiment of the present invention.
Fig. 3 is a schematic diagram of a frequency adaptive comb filter according to an embodiment of the present invention.
Fig. 4 is a discrete schematic diagram of a discrete delay module of a frequency adaptive comb filter according to an embodiment of the present invention.
Fig. 5 is a bode diagram corresponding to a transfer function of a frequency adaptive comb filter according to an embodiment of the present invention.
Detailed Description
As shown in fig. 1 and 2, the method for suppressing the harmonic disturbance of the back electromotive force of the permanent magnet synchronous motor according to the embodiment includes:
s101, acquiring an estimated value of a back electromotive force by adopting a sliding mode algorithm based on the working voltage and current of a permanent magnet synchronous motor, wherein the sliding mode algorithm adopts a preset sliding mode switching function to reduce high-frequency buffeting of back electromotive force estimation;
s102, filtering the estimated value of the back electromotive force to obtain a fundamental component of the back electromotive force;
s103, estimation of rotor position angle and speed is performed by a phase-locked loop based on the fundamental component of back electromotive force.
As shown in fig. 1 and 2, step S101 in the present embodiment includes:
s201, three-phase current of the permanent magnet synchronous motor obtained through current sensor sampling is subjected to coordinate transformation to obtain two-phase current under a static two-phase coordinate system;
s202, taking two-phase current and voltage under a static two-phase coordinate system as input quantities, constructing a sliding mode observer based on a back electromotive force equation, selecting a preset sliding mode switching function, wherein the selected sliding mode surface S is the difference value between the two-phase current under the static two-phase coordinate system and the observed value thereof, and obtaining an estimated value of the back electromotive force under the static two-phase coordinate system when the sliding mode converges.
The three-phase current is subjected to coordinate transformation to obtain two-phase current under a static two-phase coordinate system, namely Clark transformation, and a function expression of a Clark transformation matrix adopted by the Clark transformation is as follows:
Figure BDA0004006999220000051
in the above, T 3s/2s Is a Clark transformation matrix.
In this embodiment, the predetermined sliding mode switching function is a hyperbolic tangent function, and the function expression is:
Figure BDA0004006999220000052
in the above formula, F (x) represents a sliding mode switching function, tanh (x) represents a hyperbolic tangent function, m is a coefficient for adjusting a boundary layer of the hyperbolic tangent function, x is an independent variable, and e is a constant.
In this embodiment, the functional expression of the back emf equation in step S202 is:
Figure BDA0004006999220000053
/>
Figure BDA0004006999220000054
in the above, u α And u β Is the working voltage of the permanent magnet synchronous motor, R s Is the resistance of the permanent magnet synchronous motor, L d And L q Respectively the d-axis inductance and the q-axis inductance of the permanent magnet synchronous motor, p is a differential operator, omega e Is the rotor rotating speed, i of the permanent magnet synchronous motor α And i β E is the two-phase current in the stationary two-phase coordinate system α And e β Is the back electromotive force psi in a static two-phase coordinate system f Permanent magnet flux linkage of permanent magnet synchronous motor, theta e Is the rotor angle.
In this embodiment, filtering the estimated value of the back electromotive force in step S102 means filtering out the 5 th order and 7 th order harmonic components in the estimated value of the back electromotive force to obtain the fundamental component of the back electromotive force. Considering the traditional filter-free method, the back electromotive force z observed by the sliding mode α 、z β The equation containing the fundamental component and the 5 th and 7 th harmonics is as follows:
Figure BDA0004006999220000055
in the above, e 1 E is the amplitude of the fundamental component of the back EMF 5 、e 7 Amplitude, omega of 5 th and 7 th harmonic components of back electromotive force e Is the rotor rotating speed theta of the permanent magnet synchronous motor 1 Initial rotor position angle for fundamental component，θ 5 And theta 7 Initial position angles of 5 th order and 7 th order harmonic components respectively. Considering the traditional method without using a filter, the equation for directly transmitting the sliding mode observed back electromotive force to the phase-locked loop for phase estimation is as follows:
Figure BDA0004006999220000061
in the above, epsilon is the rotor position angle error, theta e For the actual rotor position angle, θ e ＝ω e t+θ 1 ，
Figure BDA0004006999220000062
For the estimated rotor position angle (currently output rotor position angle),>
Figure BDA0004006999220000063
the rotor position angle is initialized for the estimated fundamental component. When the system reaches steady state, consider that the estimated rotational speed of the phase-locked loop matches the actual rotational speed, i.e. +.>
Figure BDA0004006999220000064
The rotor position error can be reduced to:
Figure BDA0004006999220000065
as can be seen from the above equation, when the 5 th and 7 th harmonics in the sliding mode estimated back electromotive force are directly used to estimate the rotor position by the phase-locked loop without being processed by the filter, the 6 th harmonic is introduced into the rotor position error, and since the PI regulator in the phase-locked loop has low suppression capability for the 6 th ac harmonic, the 6 th harmonic is also present in the extracted estimated position when the rotor position error containing the 6 th harmonic is used as the input of the phase-locked loop. Therefore, the frequency self-adaptive comb filter suppresses the 5 th harmonic and the 7 th harmonic of back electromotive force, further suppresses the 6 th harmonic pulsation of the rotor position angle estimated by the phase-locked loop, and improves the rotor position estimation precision.
As shown in fig. 1 and fig. 2, in step S102 of this embodiment, the filter used for filtering the estimated value of the counter electromotive force is a frequency adaptive comb filter, the center frequency of the frequency adaptive comb filter is the same as the motor rotor speed of the permanent magnet synchronous motor, and the motor rotor speed of the permanent magnet synchronous motor is fed back to the frequency adaptive comb filter in real time through a phase-locked loop to implement frequency adaptation. In this embodiment, the transfer function G(s) of the frequency adaptive comb filter has the following functional expression:
Figure BDA0004006999220000066
in the above-mentioned method, the step of,
Figure BDA0004006999220000067
is the fundamental component of back EMF in a stationary two-phase coordinate system, z αβ Is the estimated value of the back electromotive force under the static two-phase coordinate system, s is an integral operator, e is a constant, j is an imaginary unit,>
Figure BDA0004006999220000068
is an estimated value of the rotor rotation speed period, and comprises: />
Figure BDA0004006999220000069
Wherein->
Figure BDA00040069992200000610
The rotor speed of the frequency adaptive comb filter is fed back to the phase-locked loop in real time. Based on the above functional expression of the transfer function G(s) of the frequency adaptive comb filter, the implementation structure of the frequency adaptive comb filter according to this embodiment is shown in fig. 3, where the frequency adaptive comb filter includes:
discrete delay module for respectively inputting estimated value z of alpha-axis back electromotive force α Estimated value z of beta-axis back electromotive force β Delay time
Figure BDA00040069992200000611
And discrete delay sampling is carried out by adopting a linear interpolation calculation method to obtain an estimated value z of alpha-axis back electromotive force after delay αd Estimated value z of beta-axis back electromotive force βd ；
An angle rotation module for estimating z of the alpha-axis back electromotive force after delay ad Estimated value z of beta-axis back electromotive force βd Multiplying a Park inverse transformation constant coefficient matrix to carry out Park inverse transformation to obtain back electromotive force estimated value
Figure BDA0004006999220000071
And->
Figure BDA0004006999220000072
A summation module for adding the estimated value z of the alpha-axis back electromotive force α Its back electromotive force estimated value obtained by angle rotation module
Figure BDA0004006999220000073
Adding, and multiplying by 1/2 to obtain back EMF fundamental wave +.>
Figure BDA0004006999220000074
Estimate z of the beta-axis back EMF to be input β And its back electromotive force estimated value +.>
Figure BDA0004006999220000075
Adding, and multiplying by 1/2 to obtain back electromotive force fundamental wave of beta axis +.>
Figure BDA0004006999220000076
The existing position harmonic suppression method is complex, and is generally used for filtering 5 th harmonic and 7 th harmonic of back electromotive force through parallel connection of multiple layers of same structures, and has a plurality of adjustable parameters; the frequency adaptive comb filter used in the embodiment has a single-layer structure, the parameters of the filter are all fixed constants, and the frequency adaptive comb filter is simple in structure and easy to apply practically.
In this embodiment, the function expression for discrete delay sampling by using the linear interpolation calculation method is:
z αd [k-N]＝f r ·z α [k-N c ]+(1-f r )·z α [k-N f ]
z βd [k-N]＝f r ·z β [k-N c ]+(1-f r )·z β [k-N f ]
in the above, z αd [k-N]、z βd [k-N]As the back electromotive force estimated value z α And back electromotive force estimation value z β The value, z, that has undergone linear interpolation processing at the kth-N sampling instants α [k-N c ]、z β [k-N c ]Back emf estimate z for the alpha axis α And a back emf estimate z of the beta axis β In the k-N c The value of each sampling instant, z α [k-N f ]、z β [k-N f ]Back emf estimate z for the alpha axis α And a back emf estimate z of the beta axis β In the k-N f The value of each sampling moment, N is the discrete sampling length, can be the decimal, f r ＝N-N f ，N f ＝floor(N)，N c ＝ceil(N)＝N f +1; wherein:
Figure BDA0004006999220000077
in the above formula, N is a discrete sampling length,
Figure BDA0004006999220000078
for the estimated value of the rotor rotation speed period, T s For the sampling period +.>
Figure BDA0004006999220000079
The rotor speed of the frequency adaptive comb filter is fed back to the phase-locked loop in real time.
In the existing vector control method, all control algorithms are realized in a discretization mode, and the high-order frequency domain transfer function of the traditional filtering method needs precise discretization, so that the digital realization of the methods is complex, and the calculation burden is increased. The frequency adaptive comb filter used in the embodiment is easy to realize digital discretization processing through a delay module and linear interpolation calculation in combination, and is simple and efficient.
Wherein, the function expression of the Park inverse transformation constant coefficient matrix is:
Figure BDA0004006999220000081
in the above, P -1 (pi/6) represents a Park inverse transform constant coefficient matrix.
FIG. 4 is a schematic diagram of a discrete implementation of the discrete delay module in this embodiment, and the equivalent of storing the original sampled signal after the delay discretization corresponds to
Figure BDA0004006999220000082
The discrete sample length of the delay is calculated as:
Figure BDA0004006999220000083
in the above formula, N is a discrete sampling length,
Figure BDA0004006999220000084
for the estimated value of the rotor rotation speed period, T s For the sampling period +.>
Figure BDA0004006999220000085
The rotor speed of the frequency adaptive comb filter is fed back to the phase-locked loop in real time. Because the discrete sampling length must be an integer, the lagrangian linear interpolation technology is adopted to process the discrete sampling length N, so that the time delay sampling precision is improved, and the specific calculation formula is as follows:
z αd [k-N]＝f r ·z α [k-N c ]+(1-f r )·z α [k-N f ]
z βd [k-N]＝f r ·z β [k-N c ]+(1-f r )·z β [k-N f ]
in the above, z αd [k-N]、z βd [k-N]As the back electromotive force estimated value z α And back electromotive force estimation value z β The value, z, that has undergone linear interpolation processing at the kth-N sampling instants α [k-N c ]、z β [k-N c ]Back emf estimate z for the alpha axis α And a back emf estimate z of the beta axis β In the k-N c The value of each sampling instant, z α [k-N f ]、z β [k-N f ]Back emf estimate z for the alpha axis α And a back emf estimate z of the beta axis β In the k-N f The value of each sampling moment, N is the discrete sampling length, f r ＝N-N f ，N f ＝floor(N)，N c ＝ceil(N)＝N f +1. Fig. 5 is a bode diagram corresponding to a transfer function of the frequency adaptive comb filter in the present embodiment, and it can be seen from the diagram that the frequency adaptive comb filter has no amplitude and no phase attenuation at a fundamental frequency, has notch characteristics at 5 th and 7 th harmonic frequencies, and can effectively filter out harmonic components in back electromotive force.
In step S103, the estimation of the rotor position angle and the speed by the phase-locked loop based on the fundamental component of the back electromotive force adopts various methods of estimating the rotor position angle and the speed as needed. For example, as an alternative implementation, as shown in fig. 1 and fig. 2, step S103 in this embodiment includes:
s301, performing per unit processing on fundamental wave components of back electromotive force under a stationary two-phase coordinate system to obtain α The axial back electromotive force fundamental wave component per unit value and the beta-axial back electromotive force fundamental wave component per unit value are expressed as the following functions:
Figure BDA0004006999220000086
Figure BDA0004006999220000087
in the above-mentioned method, the step of,
Figure BDA0004006999220000088
is the per unit value of the alpha-axis back electromotive force fundamental component, < >>
Figure BDA0004006999220000089
Is the fundamental component of the alpha-axis back EMF, +.>
Figure BDA00040069992200000810
Is the per unit value of the beta-axis back electromotive force fundamental component, < >>
Figure BDA0004006999220000091
A fundamental component of beta-axis back emf;
s302, multiplying the per unit value of the alpha-axis back electromotive force fundamental wave component by the cosine value of the rotor position angle currently output to be used as a first product term, multiplying the per unit value of the beta-axis back electromotive force fundamental wave component by the sine value of the rotor position angle currently output to be used as a second product term, multiplying the two product terms by-1 respectively, and adding to obtain a rotor position angle error, wherein the function expression is as follows:
Figure BDA0004006999220000092
in the above, epsilon is the rotor position angle error, theta e For the actual rotor position angle,
Figure BDA0004006999220000093
for the estimated rotor position angle (currently output rotor position angle);
s303, obtaining a speed estimated value of the rotor through a PI regulator by using the rotor position angle error, wherein the function expression is as follows:
Figure BDA0004006999220000094
in the above-mentioned method, the step of,
Figure BDA0004006999220000095
for estimating the speed of the rotor, k p Adjusting gain, k for proportional PI regulators i The gain is regulated for the integral of the PI regulator, t is time, epsilon is rotor position angle error; and integrating the speed estimation value of the rotor to obtain an estimated rotor position angle, wherein the function expression is as follows:
Figure BDA0004006999220000096
in the above-mentioned method, the step of,
Figure BDA0004006999220000097
is the estimated rotor position angle.
In summary, the method for suppressing harmonic disturbance of counter electromotive force of the permanent magnet synchronous motor in this embodiment obtains equivalent counter electromotive force information through the sliding mode observer, then filters 5 th harmonic and 7 th harmonic by using the frequency adaptive comb filter to obtain fundamental waves of equivalent counter electromotive force, and finally obtains the estimated value of the rotor position and the rotational speed of the motor through the phase-locked loop, thereby improving the rotor position and the rotational speed precision, and maintaining good dynamic performance. Compared with the prior art, the method for suppressing the harmonic disturbance of the counter electromotive force of the permanent magnet synchronous motor adopts the comb filter to filter the counter electromotive force harmonic, has a simple structure and is easy to discretize, and the counter electromotive force fundamental wave obtained after filtering is used for estimating the position and the rotating speed of the rotor, so that the accuracy of observing the position and the rotating speed of the rotor of the permanent magnet synchronous motor without a position sensor control method is greatly improved. Compared with the traditional method, the method for suppressing the harmonic disturbance of the back electromotive force of the permanent magnet synchronous motor is simpler to realize and easy to discretize, and the control efficiency and performance of the permanent magnet synchronous motor are further improved by suppressing the harmonic of the rotor position.
In addition, the embodiment also provides a system of the method for suppressing the back electromotive force harmonic disturbance of the permanent magnet synchronous motor, which comprises a microprocessor and a memory which are connected with each other, wherein the microprocessor is programmed or configured to execute the method for suppressing the back electromotive force harmonic disturbance of the permanent magnet synchronous motor.
Furthermore, the present embodiment also provides a computer readable storage medium having a computer program stored therein, the computer program being configured or programmed by a microprocessor to perform the method of back emf harmonic disturbance rejection of a permanent magnet synchronous motor.
It will be appreciated by those skilled in the art that embodiments of the present application may be provided as a method, system, or computer program product. Accordingly, the present application may take the form of an entirely hardware embodiment, an entirely software embodiment, or an embodiment combining software and hardware aspects. Furthermore, the present application may take the form of a computer program product embodied on one or more computer-readable storage media (including, but not limited to, disk storage, CD-ROM, optical storage, and the like) having computer-usable program code embodied therein. The present application is described with reference to flowchart illustrations and/or block diagrams of methods, apparatus (systems) and computer program products according to embodiments of the application. It will be understood that each flow and/or block of the flowchart illustrations and/or block diagrams, and combinations of flows and/or blocks in the flowchart illustrations and/or block diagrams, can be implemented by computer program instructions. These computer program instructions may be provided to a processor of a general purpose computer, special purpose computer, embedded processor, or other programmable data processing apparatus to produce a machine, such that the instructions, which execute via the processor of the computer or other programmable data processing apparatus, create means for implementing the functions specified in the flowchart flow or flows and/or block diagram block or blocks. These computer program instructions may also be stored in a computer-readable memory that can direct a computer or other programmable data processing apparatus to function in a particular manner, such that the instructions stored in the computer-readable memory produce an article of manufacture including instruction means which implement the function specified in the flowchart flow or flows and/or block diagram block or blocks. These computer program instructions may also be loaded onto a computer or other programmable data processing apparatus to cause a series of operational steps to be performed on the computer or other programmable apparatus to produce a computer implemented process such that the instructions which execute on the computer or other programmable apparatus provide steps for implementing the functions specified in the flowchart flow or flows and/or block diagram block or blocks.
The above description is only a preferred embodiment of the present invention, and the protection scope of the present invention is not limited to the above examples, and all technical solutions belonging to the concept of the present invention belong to the protection scope of the present invention. It should be noted that modifications and adaptations to the present invention may occur to one skilled in the art without departing from the principles of the present invention and are intended to be within the scope of the present invention.

Claims (10)

1. A method for back emf harmonic disturbance rejection of a permanent magnet synchronous motor, comprising:
s101, acquiring an estimated value of a back electromotive force by adopting a sliding mode algorithm based on the working voltage and current of a permanent magnet synchronous motor, wherein the sliding mode algorithm adopts a preset sliding mode switching function to reduce high-frequency buffeting of back electromotive force estimation;
s102, filtering the estimated value of the back electromotive force to obtain a fundamental component of the back electromotive force;
s103, estimation of rotor position angle and speed is performed by a phase-locked loop based on the fundamental component of back electromotive force.
2. The method of suppression of back emf harmonic disturbances of a permanent magnet synchronous motor according to claim 1, wherein step S101 comprises:
s201, three-phase current of the permanent magnet synchronous motor obtained through current sensor sampling is subjected to coordinate transformation to obtain two-phase current under a static two-phase coordinate system;
s202, taking two-phase current and voltage under a static two-phase coordinate system as input quantities, constructing a sliding mode observer based on a back electromotive force equation, selecting a preset sliding mode switching function, wherein the selected sliding mode surface S is the difference value between the two-phase current under the static two-phase coordinate system and the observed value thereof, and obtaining an estimated value of the back electromotive force under the static two-phase coordinate system when the sliding mode converges.
3. The method of claim 1, wherein filtering the estimated value of the counter electromotive force in step S102 means filtering out 5 th and 7 th harmonic components in the estimated value of the counter electromotive force to obtain a fundamental component of the counter electromotive force.
4. A method for suppressing harmonic disturbance of counter electromotive force of a permanent magnet synchronous motor according to claim 3, wherein the filter used for filtering the estimated value of counter electromotive force in step S102 is a frequency adaptive comb filter, the center frequency of the frequency adaptive comb filter is the same as the motor rotor speed of the permanent magnet synchronous motor, and the motor rotor speed of the permanent magnet synchronous motor is fed back to the frequency adaptive comb filter in real time through a phase-locked loop to realize frequency adaptation.
5. The method for suppressing harmonic disturbance of back emf of a permanent magnet synchronous motor according to claim 4, wherein the transfer function G(s) of said frequency adaptive comb filter has a functional expression as follows:
Figure FDA0004006999210000011
in the above-mentioned method, the step of,
Figure FDA0004006999210000012
is the fundamental component of back EMF in a stationary two-phase coordinate system, z αβ Is the estimated value of the back electromotive force under the static two-phase coordinate system, s is an integral operator, e is a constant, j is an imaginary unit,>
Figure FDA0004006999210000013
is an estimated value of the rotor rotation speed period and has：
Figure FDA0004006999210000014
Wherein->
Figure FDA0004006999210000015
The rotor speed of the frequency adaptive comb filter is fed back to the phase-locked loop in real time.
6. The method of back emf harmonic disturbance rejection of a permanent magnet synchronous motor according to claim 4, wherein said frequency adaptive comb filter comprises:
discrete delay module for respectively inputting estimated value z of alpha-axis back electromotive force α Estimated value z of beta-axis back electromotive force β Delay time
Figure FDA0004006999210000021
And discrete delay sampling is carried out by adopting a linear interpolation calculation method to obtain an estimated value z of alpha-axis back electromotive force after delay αd Estimated value z of beta-axis back electromotive force βd ；
An angle rotation module for estimating z of the alpha-axis back electromotive force after delay αd Estimated value z of beta-axis back electromotive force βd Multiplying a Park inverse transformation constant coefficient matrix to carry out Park inverse transformation to obtain a back electromotive force estimated value
Figure FDA0004006999210000022
And->
Figure FDA0004006999210000023
A summation module for adding the estimated value z of the alpha-axis back electromotive force α Its back electromotive force estimated value obtained by angle rotation module
Figure FDA0004006999210000024
Adding, and multiplying by 1/2 to obtain back EMF fundamental wave +.>
Figure FDA0004006999210000025
Estimate z of the beta-axis back EMF to be input β And its back electromotive force estimated value +.>
Figure FDA0004006999210000026
Adding, and multiplying by 1/2 to obtain back electromotive force fundamental wave of beta axis +.>
Figure FDA0004006999210000027
7. The method for suppressing harmonic disturbance of back electromotive force of a permanent magnet synchronous motor according to claim 6, wherein the function expression of discrete delay sampling by adopting a linear interpolation calculation method is as follows:
z αd [k-N]＝f r ·z α [k-N c ]+(1-f r )·z α [k-N f ]
z βd [k-N]＝f r ·z β [k-N c ]+(1-f r )·z β [k-N f ]
in the above, z αd [-N]、z βd [-N]As the back electromotive force estimated value z α And back electromotive force estimation value z β The value, z, that has undergone linear interpolation processing at the kth-N sampling instants α [-N c ]、z β [-N c ]Back emf estimate z for the alpha axis α And a back emf estimate z of the beta axis β In the k-N c The value of each sampling instant, z α [-N f ]、z β [-N f ]Back emf estimate z for the alpha axis α And a back emf estimate z of the beta axis β In the k-N f The value of each sampling moment, N is the discrete sampling length, can be the decimal, f r ＝-N f ，N f ＝floor()，N c ＝eil(N)＝ f +1; wherein:
Figure FDA0004006999210000028
in the above formula, N is a discrete sampling length,
Figure FDA0004006999210000029
for the estimated value of the rotor rotation speed period, T s For the sampling period +.>
Figure FDA00040069992100000210
The rotor speed of the frequency adaptive comb filter is fed back to the phase-locked loop in real time.
8. The method of suppression of back emf harmonic disturbances of a permanent magnet synchronous motor according to claim 1, wherein step S103 comprises:
s301, carrying out per unit processing on fundamental wave components of back electromotive force under a static two-phase coordinate system to respectively obtain a fundamental wave component per unit value of alpha-axis back electromotive force and a fundamental wave component per unit value of beta-axis back electromotive force;
s302, multiplying the per unit value of the alpha-axis back electromotive force fundamental wave component by the cosine value of the rotor position angle currently output to be used as a first product term, multiplying the per unit value of the beta-axis back electromotive force fundamental wave component by the sine value of the rotor position angle currently output to be used as a second product term, multiplying the two product terms by-1 respectively, and adding to obtain a rotor position angle error;
s303, the rotor position angle error is passed through a PI regulator to obtain a rotor speed estimated value, and the rotor speed estimated value is subjected to integral calculation to obtain an estimated rotor position angle.
9. A system for back emf harmonic disturbance rejection of a permanent magnet synchronous motor comprising a microprocessor and a memory interconnected, wherein the microprocessor is programmed or configured to perform the method of back emf harmonic disturbance rejection of a permanent magnet synchronous motor according to any one of claims 1 to 8.
10. A computer readable storage medium having a computer program stored therein, wherein the computer program is programmed or configured by a microprocessor to perform a method of back emf harmonic disturbance rejection of a permanent magnet synchronous motor according to any one of claims 1 to 8.
CN202211634322.2A 2022-12-19 2022-12-19 Method, system and medium for suppressing harmonic disturbance of counter electromotive force of permanent magnet synchronous motor Pending CN116054656A (en)

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN117277898A (en) * 2023-11-22 2023-12-22 泉州装备制造研究所 Permanent magnet synchronous motor prediction current control method considering harmonic disturbance

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN117277898A (en) * 2023-11-22 2023-12-22 泉州装备制造研究所 Permanent magnet synchronous motor prediction current control method considering harmonic disturbance
CN117277898B (en) * 2023-11-22 2024-02-06 泉州装备制造研究所 Permanent magnet synchronous motor prediction current control method considering harmonic disturbance

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