CN116015344A - Unmanned aerial vehicle anti-control anti-interception communication waveform design method, device and system - Google Patents
Unmanned aerial vehicle anti-control anti-interception communication waveform design method, device and system Download PDFInfo
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Abstract
The invention discloses a method, a device and a system for designing an anti-control anti-interception communication waveform of an unmanned aerial vehicle, which belong to the field of unmanned aerial vehicle communication and comprise the following steps: and setting a rate matching module parameter, a random modulation module parameter, a code hopping spread spectrum module parameter, a random phase rotation module parameter, a random time hopping module parameter, a frequency hopping module parameter and an array switching module parameter in a random process to generate an anti-control anti-interception communication waveform. The non-stationary anti-detection communication waveform has the characteristics of strong anti-detection capability, anti-parameter measurement capability, anti-signal identification capability, demodulation resistance and Doppler resistance, and the service information rate and the spread spectrum rate are flexibly set.
Description
Technical Field
The invention relates to the field of unmanned aerial vehicle communication, in particular to an unmanned aerial vehicle anti-control anti-interception communication waveform design method, device and system.
Background
Whether the unmanned aerial vehicle communication has excellent anti-control and anti-interception capability becomes a key factor for improving the survivability of the unmanned aerial vehicle platform. However, the higher the communication rate, the larger the capacity and the farther the distance means the larger the intercepted risk, and how to improve the interception resistance of the unmanned aerial vehicle communication as much as possible when guaranteeing high-rate and long-distance transmission is a difficult problem to be solved.
Disclosure of Invention
The invention aims to overcome the defects of the prior art and provide a method, a device and a system for designing an unmanned aerial vehicle anti-control anti-interception communication waveform, which have the characteristics of non-stable anti-detection communication waveform, strong anti-detection capability, anti-parameter measurement capability, signal identification capability, demodulation capability and Doppler capability, and flexible service information rate and spread spectrum rate setting.
The invention aims at realizing the following scheme:
an unmanned aerial vehicle anti-control anti-interception communication waveform design method comprises the following steps:
and setting a rate matching module parameter, a random modulation module parameter, a code hopping spread spectrum module parameter, a random phase rotation module parameter, a random time hopping module parameter, a frequency hopping module parameter and an array switching module parameter in a random process to generate an anti-control anti-interception communication waveform.
Further, after generating the anti-control anti-interception communication waveform, the method further comprises the steps of: and receiving the anti-control anti-interception communication waveform at a receiving end.
Further, the introducing random process sets a rate matching module parameter, a random modulation module parameter, a code hopping spread spectrum module parameter, a random phase rotation module parameter, a random time hopping module parameter, a frequency hopping module parameter and an array switching module parameter, and the method comprises the following sub-steps:
S1, generating service information for transmitting user data in a time slot, obtaining a code sequence C through a CRC (cyclic redundancy check) coding module, a Turbo coding module and a rate matching module, wherein the rate matching module is controlled by a random code rate generating module to obtain code sequences with different code rates, and the random code rate generating module generates a real value l of ith code residence time through a chaotic random process i Obtaining a quantized value g by a quantizer i The code rate controller is directed to randomly select the currently used code rate from the code rate set;
s2, the code sequence C is subjected to a random modulation module to obtain a modulation sequence M, the random modulation module is controlled by a random modulation generation module, and the random modulation generation module generates a real value o of the ith modulation residence time by the chaotic random process in the step S1 i Obtaining quantized value p by quantizer i The modulation controller is guided to randomly select the currently used modulation type from the modulation set;
s3, the modulation sequence M is subjected to a code hopping spread spectrum module to obtain a code hopping spread spectrum sequence S, the code hopping spread spectrum module is controlled by a code hopping generation module, and the code hopping generation module generates a real value y of the ith code hopping residence time by the chaotic random process in the step S1 i Obtaining quantized value z by quantizer i The method comprises the steps that a code hopping controller is guided to randomly select a spreading code sequence with spreading gain and spreading rate which are currently used from a code hopping sequence set, the local oscillation frequencies of the spreading rate and the service information rate are non-homologous, and different spreading codes have good cross-correlation properties;
S4, the code hopping spread spectrum sequence S obtains a service information baseband signal D through a random phase rotation module, the random phase rotation module is controlled by a random phase generation module, and the random phase generation module generates a real value h of the ith phase residence time through a chaotic random process in the step S1 i As offset phase, namely:
D i ＝S i ×exp(-j complex 2πh i )
wherein D is i And S is i Service baseband signal and code hopping spread spectrum signal, j, respectively, of ith phase dwell time complex Is an imaginary number mark;
s5, using initial values of chaotic random processes used by a random code rate generation module, a random modulation generation module, a code hopping generation module and a random phase generation module in a time slot as communication waveform characteristic parameter indication information so as to guide a receiving end to process a service baseband signal;
s6, after the synchronous information and the communication waveform characteristic parameter indication information are combined through a formatting module, a control baseband signal is obtained through a random modulation module, a code hopping spread spectrum module and a random phase rotation module by adopting a fixed pattern in sequence, wherein the spread spectrum rate and the local oscillation frequency of the control information rate are homologous;
s7, after the service baseband signal and the control baseband signal pass through a framing module, a time slot radiation signal F is obtained, radiation time jitter is generated through a random time hopping module, the random time hopping module is controlled by a time hopping generating module, and the time hopping generating module generates a real value h of the ith time hopping residence time through a chaotic random process in the step S1 i Obtaining quantized value c by quantizer i The controller randomly selects the currently used radiation starting time from the time hopping set;
s8, the baseband signal TH output by the random time hopping module is randomly selected by the array switching module, the array element combination conducted in the one-dimensional linear array switching residence time is conducted, then the delay amplitude adding weighting module in the conducting array element channel respectively carries out delay and amplitude adding weighting processing on the baseband signal TH, so that the main beam of the synthesized signal is aligned to the receiver, the array switching module is controlled by the array selection module, and the array selection module generates a real value v of the ith array switching residence time in the chaotic random process in the step S1 i Obtaining a quantized value r by a quantizer i To direct the array element selection controller to randomly select the currently used array from the optimal array setCombining array elements;
s9, signal vector U= [ U ] of the multipath signals processed by the delay amplitude-phase addition weighting module 1 ,U 2 ,…,U N ]Through up-conversion processing and digital-to-analog conversion processing, finally, the up-converted carrier frequency is generated by a frequency hopping module, and the frequency hopping module generates a real value x of the ith frequency residence time by the chaotic random process in the step S1 i Obtaining quantized value p by quantizer i To direct the frequency hopping controller to randomly select the currently used frequency from the set of frequencies.
Further, the receiving the anti-control anti-interception communication waveform at the receiving end comprises the following substeps:
s10, the receiving end carries out down-conversion treatment on the received signal, and then carries out treatment on the received signal through an analog-to-digital conversion module and a band-pass filter to obtain a digital intermediate frequency signal IF, namely:
where IF (k) is the kth sample time intermediate frequency signal, ψ (k) and n (k) are the kth sample time baseband signal and Gaussian noise,
for receiving intermediate frequency, θ is phase deviation, and T is sampling time;
s11, the capturing module obtains a local synchronous baseband signal according to known synchronous information and a baseband processing mode of a fixed pattern, gathers samples according to known frequency, simultaneously carries out parallel down-conversion processing on an intermediate frequency signal, correlates with the local synchronous baseband signal and carries out threshold detection, and the frequency and the time corresponding to the maximum correlation peak value which crosses the threshold are estimated as the frequency hopping frequency and the starting time adopted by the current time slot; then the local synchronous baseband signal is utilized to eliminate the information phase of the receiving synchronous segment, and the length L after the information phase is eliminated is obtained 0 Is a reception synchronization signal y of:
In the middle of
For Doppler frequency, n' (k) is baseband Gaussian noise, y (k) is a k-th sampling time signal of y, and x is conjugate operation; sequentially carrying out phase difference, weighting and cumulative average operation on the sampling signals at the front and rear moments on y to finish the minimum mean square unbiased estimation of delta f>
Namely:
in the middle of
For Doppler unbiased estimation, w (·) is a weighted window function, R (·) is a time domain autocorrelation function, arg (·) is a phase-taking operation; finally, using Doppler unbiased estimation +.>
Canceling out the influence of Doppler linear phase and estimating the phase deviation +.>
Namely:
s12, the tracking module performs Doppler unbiased estimation according to the output of the capturing module
And phase deviation->
Generating a local initial carrier, respectively regenerating a control channel and a service channel local spread spectrum signal after random phase rotation based on a known fixed pattern and a baseband processing mode of communication waveform characteristic parameter indication information provided by the control channel, and sequentially adjusting a receiver carrier local oscillation frequency, a spread spectrum code rate local oscillation frequency and an information rate local oscillation frequency by taking a carrier phase difference, a carrier frequency difference, a code phase difference and a modulation symbol phase difference of a local signal and a received signal as inputs in a mode of carrier loop, code loop and modulation symbol loop linkage to complete carrier phase synchronization, symbol synchronization and modulation symbol synchronization of the received control signal and the service signal;
S13, the synchronous control baseband signal output by the tracking module is sequentially subjected to a random phase rotation decoding module, a code hopping despreading module and a baseband demodulation module according to a known fixed pattern processing mode to obtain waveform characteristic indicating parameters;
s14, the synchronized service baseband signal output by the tracking module is processed according to the communication waveform characteristic parameter indicating information analyzed by the control baseband signal, and service information is recovered through the de-random phase rotation module, the code hopping de-spreading module, the baseband demodulation module, the Turbo decoding module and the CRC decoding module in sequence.
Further, in step S8, the optimal array set satisfies the following constraint condition:
s81, minimizing array side lobe level constraint conditions:
{a sidelobe |P sidelobe ≤Th sidelobe }
S＝{θ||θ min ≤θ≤θ d -Δθ/2∪θ d +Δθ/2≤θ≤θ max }
p in the formula sidelobe Average sidelobe level Th of conducting array set after amplitude addition weighting sidelobe For the average side lobe threshold, F (θ) is the antenna gain with direction θ, S is the side lobe angle range, S num For the number of samples of the angle range of the side lobe, theta min For the lower limit of the angle range of the side lobe, theta max For the upper limit of the range of the side lobe angle, delta theta is 3dB main lobe width, f n (θ) is the n-th array element pattern, θ d For the communication target direction, a sidelobe To meet P sidelobe ≤Th sidelobe Array set of constraints and a= [ a ] 1 ,a 2 ,...,a n ]，a n The n-th array element is conducted with a switch, the value is 0 or 1, lambda is the carrier wavelength, and d is the distance between adjacent array elements of the linear array;
minimizing target directional distortion constraints:
{a target |Λ target ≤Th target }
Λ target ＝∑|D i F(θ d )-D i |
middle lambda target Th is the target direction distortion degree target For the target direction distortion threshold, D i An ith modulation constellation point, a, for a traffic information modulated signal D target To satisfy lambda target ≤Th target Array set of constraints and a target ＝[a 1 ,a 2 ,...,a n ]，a n Turned on for the nth array elementThe value of the switch is 0 or 1;
maximizing non-target directional distortion constraints:
{a non-target |Λ non-target ≥Th non-target }
middle lambda non-target Th is non-target direction distortion degree non-target A is a non-target direction distortion threshold non-target To satisfy lambda non-target ≥Th non-target Array set of constraints and a non-target ＝[a 1 ,a 2 ,...,a n ]，a n For the n-th array element to turn on the switch, the value is 0 or 1, step is the angle step length, M mod Is the modulation order;
s82, setting an average sidelobe threshold Th sidelobe Target direction distortion threshold Th target And a non-target direction distortion threshold Th non-target Solving the mathematical model under the three constraint conditions in the step S81 to obtain an optimal array set, and randomly selecting array combinations from the optimal array set according to different array switching residence time to perform radiation switching of multiple array elements.
Further, in step S12, the calculation methods of the carrier phase difference, carrier frequency difference, code phase difference and modulation symbol phase difference of the local signal and the received signal are as follows:
Carrier phase difference and carrier frequency difference: firstly regenerating a control channel or service channel local spread spectrum signal psi (k) subjected to random phase rotation in the ith coherent integration time, and then utilizing Doppler unbiased estimation
And phase deviation->
Initializing local carrier wave, down-converting received signal and then combining with local carrier wavePerforming coherent integration on the spread spectrum signal psi (k), and then performing phase extraction operation to obtain a carrier phase residual, namely:
t in coh Is the coherent integration time, T is the sampling time, T is the conjugate operation, τ i For the phase deviation of the local spread spectrum signal and the received signal code in the ith coherent integration time, phi i For the carrier phase residual in the ith coherent integration time,
receiver carrier local oscillator frequency for the ith coherent integration time and initialized to +.>
Receiver carrier initial phase for the ith coherent integration time and initialized to +.>
mod (a, b) is a modulo b operation;
the carrier frequency residual error is obtained by carrying out differential operation on two adjacent coherent integration time carrier phase residual errors, namely:
finally utilizing the infinite length unit impulse response digital filter pair phi i And F i And (3) filtering to finish updating the carrier local oscillation frequency, namely:
ω 1 ＝1.275×B loop
in c 0 ，c 1 ，c 2 ，e 0 ，e 1 And e 2 Digital filter weight coefficient, omega of infinite length unit impulse response respectively 1 Is oscillation frequency, K is loop gain, B loop Is loop bandwidth;
receiver carrier initial phase within the ith coherent integration time
The updating is as follows:
code phase bias:
first regenerating the channel in the ith coherent integration timeThe control channel or traffic channel local spread spectrum signal ψ (k) after random phase rotation is then subject to the lead and lag code correlation interval t dll Operation to obtain the advanced local spread signal ψ (k+t) dll ) And lagging the local spread spectrum signal ψ (k-t dll ) Then uses the down-converted received baseband signal and advanced local spread spectrum signal psi (k+t) dll ) And lagging the local spread spectrum signal ψ (k-t dll ) After the coherent integration processing, the code phase deviation operation is completed, namely:
in the middle of
And->
The baseband signal and the advanced local spread spectrum signal ψ (k+t) are received in the ith coherent integration time, respectively dll ) And lagging the local spread spectrum signal ψ (k-t dll ) T is the integral result of (1) i code ＝1/f i code For the local spreading chip duration, f, in the ith coherent integration time i code For the i-th coherent integration time spreading code local oscillation frequency,/and>
and
receiving noise and leading book for ith coherent integration time respectivelyGround spread signal ψ (k+t) dll ) And lagging the local spread spectrum signal ψ (k-t dll ) Is a result of integration of (a);
finally, using infinite length unit impulse response digital filter pair tau i Filtering to update local oscillation frequency of spread spectrum code
ω 2 ＝1.886×B loop
Wherein K is loop gain, B loop For loop bandwidth omega 2 Is the oscillation frequency;
modulation symbol phase deviation: first, a modulation symbol correlation interval t is set mll Respectively extracting [ iT ] coh -t mll ,(i+1)T coh -t mll ]，[iT coh ,(i+1)T coh ]，[iT coh +t mll ,(i+1)T coh +t mll ]And the received signals in the integral time are subjected to coherent integration processing with the local spread spectrum signals after the down-conversion of the three types of signals, namely:
in the method, in the process of the invention,
for the correlation integration time [ iT ] coh ,(i+1)T coh ]Integration of the internal received signal with the local signal, < >>
For the correlation integration time [ iT ] coh ,(i+1)T coh ]Receiving an integration result of noise and a local signal; />
For the correlation integration time [ iT ] c o h -t mll ,(i+1)T coh -t mll ]Integration of the internal received signal with the local signal, < >>
For the correlation integration time [ iT ] coh ,(i+1)T coh ]Receiving an integration result of noise and a local signal; />
For the correlation integration time [ iT ] coh +t mll ,(i+1)T coh +t mll ]Integration of the internal received signal with the local signal, < >>
For the correlation integration time [ iT ] coh ,(i+1)T coh ]Receiving an integration result of noise and a local signal; t (T) i mod ＝1/f i mod For the local modulation symbol duration, f, in the ith coherent integration time i mod The local oscillation frequency is the information rate in the ith coherent integration time;
then utilize adjacent
Polarity change by->
And->
Calculating modulation symbol phase deviation +.>
Namely:
finally, utilizing infinite length unit impulse response digital filter pair 
Filtering to update local oscillation frequency of information rate
ω 2 ＝1.886×B loop
Wherein K is loop gain, B loop For loop bandwidth omega 2 Is the oscillation frequency.
Further, in step S1, the method includes the sub-steps of: the range of the code rate set is {0.1,0.2,0.3,0.4,0.5,0.6,0.7,0.8}, the chaotic random process adopts Logistic mapping, and the quantizer adopts a nonlinear quantization equation:
in the formula, the quantized value g i =j, q is the number of samples of the code rate set;
in step S2, the sub-steps are included: the modulation set is {2PSK,4PSK,8PSK,16PSK }, and the nonlinear quantization equation in the step S1 is adopted to satisfy the quantization value p i =j, j=0, 1..q-1, q is the number of modulation set samples;
in step S3, the sub-steps are included: the non-linear quantization equation in the step S1 is adopted to satisfy the quantization value z i =j, j=0, 1..q-1, q is the number of samples of the set of hopping sequences;
in step S7, the sub-steps are included: the non-linear quantization equation in the step S1 is adopted to satisfy the quantization value c i ＝j，j＝0,1...,q-1，qFor the number of time-hopping set samples, the time-hopping residence time is one time slot time length;
in step S8, the sub-steps are included: the non-linear quantization equation in the step S1 is adopted to satisfy the quantization value r i =j, j=0, 1..q-1, q is the optimal array set number of samples;
In step S9, the sub-steps are included: the non-linear quantization equation in the step S1 is adopted to satisfy the quantization value p i =j, j=0, 1..q-1, q is the number of samples of the frequency set and the frequency dwell time is one slot time length.
The unmanned aerial vehicle anti-control anti-interception communication waveform generation device comprises a CRC coding module, a Turbo coding module, a rate matching module, a random modulation module, a code hopping spread spectrum module, a random phase rotation module, a random code rate generation module, a random modulation generation module, a code hopping generation module, a random phase generation module, a framing module, a random time hopping module, a time hopping generation module, a frequency hopping module, an up-conversion module, a digital-to-analog conversion module, an array switching module, an array element selection module, a delay amplitude addition weight module and a multi-array element antenna, wherein the modules operate based on the method.
The unmanned aerial vehicle anti-control anti-interception communication waveform receiving device comprises a down-conversion module, an analog-to-digital conversion module, a band-pass filter, a capturing module, a tracking module, a de-random rotation module, a code hopping de-spreading module, a baseband demodulation module, a Turbo decoding module and a CRC decoding module, wherein the modules operate based on the method.
An unmanned aerial vehicle anti-control anti-interception communication system comprises the waveform generation device and the waveform receiving device.
The beneficial effects of the invention include:
(1) Resistance to detection: according to the technical scheme, an optimal array set is constructed according to the criteria of minimizing the array side lobe level, the criteria of minimizing the target direction distortion and the criteria of maximizing the non-target direction distortion, the array switching module is controlled by a random process, the conducting array elements are randomly selected from the optimal array set, the delayed amplitude addition weighting processing is carried out, the directional beam pointing to the receiver on the my side is formed through the electromagnetic wave dry-emission effect, the low side lobe gain is formed in the non-cooperative direction, the airspace exposure energy is reduced, and the detection performance of the third-party detection platform is reduced.
(2) Signal recognition resistance and demodulation resistance: according to the technical scheme, an optimal array set is constructed according to the minimum array side lobe level criterion, the minimum target direction distortion criterion and the maximum non-target direction distortion criterion, the array switching module is controlled by a random process, the conducting array elements are randomly selected from the optimal array set, delay amplitude addition weighting processing is carried out, and under the condition that the low distortion degree of the directional signals of the receiver is ensured, the non-cooperative direction distortion degree is maximized, so that the signal adaptability and the signal identification performance of the third party detection platform are reduced. Even if the third party detection platform and the my receiver are in the same direction, the random modulation module and the random phase rotation module are controlled through a random process so as to maximize the constellation pattern and uncertainty of the radiation signal, reduce the signal identification performance of the third party detection platform and increase the difficulty of correctly demodulating the detection signal.
(3) Resistance to parameter measurement: according to the technical scheme, the rate matching module, the random modulation module, the code hopping spread spectrum module, the random time hopping module and the frequency hopping module are controlled by a random process, so that the modulation symbol rate, the spread spectrum code rate, the spread spectrum period, the radiation period and the carrier frequency are dynamically changed, the cyclic stationarity is not possessed, and the third party detection platform cannot realize stable measurement of signal parameters through long-term observation.
(4) The service information rate and the spread spectrum rate are flexibly set and have strong Doppler resistance: according to the technical scheme, the local oscillators of the service information rate and the spread spectrum rate are non-homologous, the service information rate and the spread spectrum rate can be set independently, and the flexibility of parameter design is improved. Because the Doppler influence of the relative motion on the two local oscillation frequencies is different, the service information and the phase jump of the spread spectrum code have an uncertain relation, so that the code element and the modulation symbol are not synchronous under the coherent integration time, and the receiver cannot demodulate correctly. The invention adopts the mode of linkage of carrier loop, code loop and modulation symbol loop to calculate carrier phase difference, carrier frequency difference, code phase difference and modulation symbol phase difference between local signal and received signal, and after processing by an infinite length unit impact response digital filter, the carrier local oscillation frequency, spread spectrum code rate local oscillation frequency and information rate local oscillation frequency of the receiver are sequentially adjusted to complete carrier phase synchronization, symbol synchronization and modulation symbol synchronization of service signals, thereby enhancing Doppler resistance of the receiver.
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In order to more clearly illustrate the embodiments of the invention or the technical solutions of the prior art, the drawings which are used in the description of the embodiments or the prior art will be briefly described, it being obvious that the drawings in the description below are only some embodiments of the invention, and that other drawings can be obtained according to these drawings without inventive faculty for a person skilled in the art.
Fig. 1 is a schematic diagram of an anti-control anti-interception communication waveform time slot of an unmanned aerial vehicle according to an embodiment of the present invention;
fig. 2a is a first schematic diagram of an anti-control anti-interception communication waveform transmitting and receiving process of the unmanned aerial vehicle according to an embodiment of the present invention;
fig. 2b is a second schematic diagram of an anti-control anti-interception communication waveform transmitting and receiving process of the unmanned aerial vehicle according to an embodiment of the present invention;
fig. 3a is a schematic diagram of a preferred array set embodiment of the present invention, and a first schematic diagram of a receiver and a third party detection platform receiving signal modulation constellations;
FIG. 3b is a schematic diagram of a second preferred embodiment of the present invention, wherein the second preferred embodiment of the present invention is a receiver and third party detection platform for receiving signal modulation constellations;
FIG. 3c is a third schematic diagram of the preferred array set embodiment of the present invention and the third party detection platform and the receiver thereof receiving signal modulation constellations;
FIG. 3d is a fourth schematic diagram of the preferred array set embodiment of the present invention and the receiver and third party detection platform receiving signal modulation constellations;
FIG. 3e is a fifth schematic diagram of the preferred array set embodiment of the present invention and the receiver and third party detection platform receiving signal modulation constellations;
FIG. 3f is a sixth schematic diagram of a preferred array set embodiment of the present invention and a third party detection platform and a receiver thereof for receiving signal modulation constellations;
FIG. 4a is a first comparison diagram of the Doppler resistance of the tracking module according to the embodiment of the present invention and a conventional Garder method tracking module;
FIG. 4b is a second comparative schematic diagram of the Doppler resistance of the tracking module according to the embodiment of the present invention and the conventional Garder method tracking module.
Detailed Description
All of the features disclosed in all of the embodiments of this specification, or all of the steps in any method or process disclosed implicitly, except for the mutually exclusive features and/or steps, may be combined and/or expanded and substituted in any way.
In view of the problems in the background, the inventors of the present invention further recognized that: at present, the anti-control anti-interception technology mainly adopts a mixed spread spectrum system in unmanned aerial vehicle communication application, integrates spread spectrum/frequency hopping/time hopping technology, adopts a cascade coding modulation scheme of channel coding and high frequency spectrum efficiency modulation, and has the effects that:
1) The multi-frequency point and high-speed frequency hopping mode is adopted, and the third party detection platform generates bandwidth mismatch due to the fact that the rapid frequency hopping cannot be tracked;
2) By adopting a spread spectrum modulation mode, the signal power spectrum is widened and submerged under noise, so that the signal detection becomes very difficult;
3) High gain channel coding is adopted, so that error correction code gain is brought;
4) The transmission time is changed (transmission jitter is introduced, which is equivalent to time hopping), and the third party detection platform cannot align with the communication signal due to the randomness of the transmission time, so that time mismatch loss is also introduced.
In summary, the present unmanned aerial vehicle communication mainly uses frequency hopping technology, spread spectrum technology and time hopping technology, and signal parameters in a radiation section are single, so that a communication signal presents stable regular distribution characteristics in a time-frequency domain and a modulation constellation, and is easy to be measured and identified by a third party detection platform parameter. Therefore, after creative thinking, the technical scheme of the invention starts from the thought of inhibiting and eliminating the attackeable statistical characteristics existing in the traditional communication signals, and aims to construct an unmanned aerial vehicle anti-control anti-interception communication waveform, so that the unmanned aerial vehicle anti-control anti-interception communication waveform has time-varying characteristics of a time domain, a frequency domain and a modulation domain, a wireless communication signal transmitted in space has strong unpredictable and non-reconfigurable characteristics, obvious non-stationary statistical characteristics and non-various state experience properties, and a third party detection platform cannot acquire communication parameters by observing the communication signals for a long time so as to acquire control parameters, thereby realizing physical layer safety communication in a real sense.
In the further inventive concept, aiming at the technical problems of easy detection, parameter measurement and modulation recognition of communication signals, the invention provides an unmanned aerial vehicle anti-control anti-interception communication waveform design scheme, wherein a random process is introduced to set a rate matching module parameter, a code hopping spread spectrum module parameter, a random time hopping module parameter and a frequency hopping module parameter so as to inhibit and eliminate detectable and attackeable statistical characteristics existing in traditional communication signals, so that the performances of signal adaptability, parameter measurement and the like of a third party detection platform are reduced, and meanwhile, a random phase rotation module parameter, a random modulation module parameter and an array switching module parameter are introduced to set a random process, so that a directional beam is formed in the direction of a receiver on the my side, a non-cooperative direction forms a low gain sidelobe and an original baseband signal constellation is severely distorted, and the performances of signal detection and modulation recognition of the third party detection platform are reduced.
Specifically, as shown in fig. 1, the unmanned plane anti-control anti-interception communication waveform provided by the technical scheme of the invention is burst communication by taking time slots as units. The slot radiation segment message mainly includes control information and service information. The control information is used for transmitting the synchronous information and the communication waveform characteristic parameter indication information, and the service information is used for guiding the processing mode of the service information. The service information is used for transmitting user messages. The control information adopts a fixed pattern to sequentially pass through a random modulation module, a code hopping spread spectrum module and a random phase rotation module to obtain a control baseband signal, wherein the spread spectrum rate and the local oscillation frequency of the control information rate are homologous. The service information adopts a random pattern of communication waveform characteristic parameter indication information to sequentially pass through a random modulation module, a code hopping spread spectrum module and a random phase rotation module to obtain a service baseband signal, wherein the spread spectrum rate and the local oscillation frequency of the control information rate are non-homologous.
As shown in fig. 2a to fig. 2b, the unmanned aerial vehicle anti-control anti-interception communication waveform design scheme provided by the technical scheme of the invention actually comprises a design method, a system structure and a corresponding application device, wherein the system structure mainly comprises a CRC coding module, a Turbo coding module, a rate matching module, a random modulation module, a code hopping spread spectrum module, a random phase rotation module, a random code rate generation module, a random modulation generation module, a code hopping generation module, a random phase generation module, a formatting module, a framing module, a random time hopping module, a time hopping generation module, a frequency hopping module, an up-conversion module, a digital-to-analog conversion module, an array switching module, an array element selection module, a delay amplitude adding weight module, a multi-array element antenna, a down-conversion module, an analog-to-digital conversion module, a band-pass filter, a capturing module, a tracking module, a random rotation module, a code hopping despreading module, a baseband demodulation module, a Turbo decoding module and a CRC decoding module, wherein the specific implementation steps of the design method are as follows:
1) Firstly, generating service information for transmitting user information in a time slot, obtaining a code sequence C through a CRC (cyclic redundancy check) coding module, a Turbo coding module and a rate matching module, wherein the rate matching module is controlled by a random code rate generating module to obtain code sequences with different code rates, and the random code rate generating module generates a real value l of ith code residence time through a chaotic random process i Obtaining g through a quantizer i The code rate controller is directed to randomly select the currently used coding code rate from a code rate set, the range of the code rate set is {0.1,0.2,0.3,0.4,0.5,0.6,0.7,0.8}, the chaotic random process adopts Logistic mapping, and the quantizer adopts a nonlinear quantization equation:
in the formula, the quantized value g i And =j, q is the number of samples of the code rate set.
2) The code sequence C is subjected to a random modulation module to obtain a modulation sequence M, the random modulation module is controlled by a random modulation generation module, and the random modulation generation module generates a real value o of the ith modulation residence time by the chaotic random process in the step 1) i Obtaining p through quantizer i The modulation controller is guided to randomly select the currently used modulation type from the modulation set, the modulation set is {2PSK,4PSK,8PSK,16PSK }, the nonlinear quantization equation in the step 1) is adopted, and the quantization value p is satisfied i =j, j=0, 1..q-1, q is the number of modulation set samples.
3) The modulation sequence M is subjected to a code hopping spread spectrum module to obtain a code hopping spread spectrum sequence S, the code hopping spread spectrum module is controlled by a code hopping generation module, and the code hopping generation module generates a real value y of the ith code hopping residence time by the chaotic random process in the step 1) i Obtaining z through quantizer i The method comprises the steps that a code hopping controller is guided to randomly select a spreading code sequence with spreading gain and spreading rate which are currently used from a code hopping sequence set, local oscillation frequencies of the spreading rate and service information rate are non-homologous, different spreading codes have good cross-correlation properties, a nonlinear quantization equation in the step 1) is adopted, and a quantization value z is satisfied i =j, j=0, 1..q-1, q is the number of samples of the set of hopping sequences.
4) The code hopping spread spectrum sequence S obtains a service information baseband signal D through a random phase rotation module, the random phase rotation module is controlled by a random phase generation module, wherein the random phase generation module generates a real value h of the i-th phase residence time through the chaotic random process in the step 1) i As offset phase, namely:
D i ＝S i ×exp(-j complex 2πh i )
wherein D is i And S is i Service baseband signal and code hopping spread spectrum signal, j, respectively, of ith phase dwell time complex Is an imaginary number identification.
5) And using initial values of the chaotic random process used by the random code rate generation module, the random modulation generation module, the code hopping generation module and the random phase generation module in the time slot as communication waveform characteristic parameter indication information so as to guide a receiving end to process a service baseband signal.
6) After the synchronous information and the communication waveform characteristic parameter indication information are combined by a formatting module, a control baseband signal is obtained by adopting a fixed pattern and sequentially passing through a random modulation module, a code hopping spread spectrum module and a random phase rotation module, wherein the spread spectrum rate and the local oscillation frequency of the control information rate are homologous.
7) The service baseband signal and the control baseband signal are subjected to a framing module to obtain a time slot radiation signal F, a random time hopping module generates radiation time jitter, the random time hopping module is controlled by a time hopping generating module, and the time hopping generating module generates a real value h of the ith time hopping residence time by the chaotic random process in the step 1) i C is obtained through a quantizer i To direct the time hopping controller to randomly select the currently used radiation starting time from the time hopping set, and the nonlinear quantization equation in the step 1) is adopted to meet the quantization value c i =j, j=0, 1..q-1, q is the number of samples of the set of hops, and the time-hopping residence time is one slot time length.
8) The baseband signal TH output by the random time hopping module is randomly selected by the array switching module to conduct array element combination in the one-dimensional linear array switching residence time, then a delay amplitude adding weighting module in a conducting array element channel respectively conducts delay and amplitude adding weighting processing on the baseband signal TH, so that a main beam of a synthesized signal is aligned to the my receiver, the array switching module is controlled by the array selection module, wherein the array selection module generates a real value v of the ith array switching residence time by the chaotic random process in the step 1) i Obtaining r through quantizer i The array element selection controller is guided to randomly select the currently used array element combination from the optimal array set, and the nonlinear quantization equation in the step 1) is adopted to meet the quantization value r i =j, j=0, 1..q-1, q is the optimal array set sample number.
9) Signal vector processed by multipath signal through delay amplitude-phase weighting module U＝[U 1 ,U 2 ,…,U N ]Through up-conversion treatment and digital-to-analog conversion treatment, finally, the up-converted carrier frequency is generated by a frequency hopping module which generates a real value x of the ith frequency residence time by the chaotic random process in the step 1) through radiation of a multi-array element antenna to a channel, wherein N is the number of array elements of the multi-array element antenna i Obtaining p through quantizer i The frequency hopping controller is guided to randomly select the currently used frequency from the frequency set, and the nonlinear quantization equation in the step 1) is adopted to meet the quantization value p i =j, j=0, 1..q-1, q is the number of samples of the frequency set and the frequency dwell time is one slot time length.
10 The receiving end firstly carries out down-conversion treatment on the received signal, and then carries out treatment on the received signal through an analog-to-digital conversion module and a band-pass filter to obtain a digital intermediate frequency signal IF, namely:
where IF (k) is the kth sample time intermediate frequency signal, ψ (k) and n (k) are the kth sample time baseband signal and Gaussian noise,
for receiving the intermediate frequency, θ is the phase deviation, and T is the sampling time.
11 The capturing module obtains a local synchronous baseband signal according to the known synchronous information and the baseband processing mode of the fixed pattern, simultaneously carries out parallel down-conversion processing on the intermediate frequency signal according to all the known frequency set samples, correlates with the local synchronous baseband signal and carries out threshold detection, and the frequency and the time corresponding to the maximum correlation peak value which passes the threshold are estimated as the frequency hopping frequency and the starting time adopted by the current time slot. Then the local synchronous baseband signal is utilized to eliminate the information phase of the receiving synchronous segment, and the length L after the information phase is eliminated is obtained 0 Is a reception synchronization signal y of:
in the middle of
For Doppler frequency, n' (k) is baseband Gaussian noise, y (k) is the kth sampling time signal of y, and x is conjugate operation. Sequentially carrying out phase difference, weighting and cumulative average operation on the sampling signals at the front and rear moments on y to finish the minimum mean square unbiased estimation of delta f>
Namely:
in the middle of
For Doppler unbiased estimation, w (·) is a weighted window function, R (·) is a time-domain autocorrelation function, and arg (·) is a phase-taking operation. Finally, using Doppler unbiased estimation +.>
Canceling out the influence of Doppler linear phase and estimating the phase deviation +.>
Namely:
12 A tracking module for estimating Doppler unbiased according to the output of the capturing module
And phase deviation->
And generating a local initial carrier, respectively regenerating a control channel and a service channel local spread spectrum signal after random phase rotation based on a known fixed pattern and a baseband processing mode of communication waveform characteristic parameter indication information provided by the control channel, and sequentially adjusting a receiver carrier local oscillation frequency, a spread spectrum code rate local oscillation frequency and an information rate local oscillation frequency by taking a carrier phase difference, a carrier frequency difference, a code phase difference and a modulation symbol phase difference of a local signal and a received signal as inputs in a mode of carrier loop, code loop and modulation symbol loop linkage to complete carrier phase synchronization, symbol synchronization and modulation symbol synchronization of the received control signal and the service signal.
13 The synchronous control baseband signal output by the tracking module, and the waveform characteristic indicating parameters are obtained by sequentially carrying out a de-random phase rotation module, a code hopping de-spreading module and a baseband demodulation module according to a known fixed pattern processing mode.
14 The synchronous service baseband signal output by the tracking module is processed according to the communication waveform characteristic parameter indicating information analyzed by the control baseband signal, and service information is recovered through the de-random phase rotation module, the code hopping de-spreading module, the baseband demodulation module, the Turbo decoding module and the CRC decoding module in sequence.
In the foregoing embodiment, in a further implementation manner, the optimal array set in step 8) satisfies the following constraint condition:
a) Minimizing array sidelobe levels
{a sidelobe |P sidelobe ≤Th sidelobe }
S＝{θ|θ min ≤θ≤θ d -Δθ/2∪θ d +Δθ/2≤θ≤θ max }
P in the formula sidelobe Average sidelobe level Th of conducting array set after amplitude addition weighting sidelobe For the average side lobe threshold, F (θ) is the antenna gain with direction θ, S is the side lobe angle range, S num For the number of samples of the angle range of the side lobe, theta min For the lower limit of the angle range of the side lobe, theta max For the upper limit of the range of the side lobe angle, delta theta is 3dB main lobe width, f n (θ) is the n-th array element pattern, θ d For the communication target direction, a sidelobe To meet P sidelobe ≤Th sidelobe Array set of constraints and a= [ a ] 1 ,a 2 ,...,a n ]，a n The n-th array element is conducted to form a switch, the value is 0 or 1, lambda is the carrier wavelength, and d is the distance between adjacent array elements of the linear array.
b) Minimizing target directional distortion
{a target |Λ target ≤Th target }
Λ target ＝∑|D i F(θ d )-D i |
Middle lambda target Th is the target direction distortion degree target For the target direction distortion threshold, D i An ith modulation constellation point, a, for a traffic information modulated signal D target To satisfy lambda target ≤Th target Array set of constraints and a target ＝[a 1 ,a 2 ,...,a n ]，a n The switch is turned on for the nth array element, and the value is 0 or 1;
c) Maximizing non-target directional distortion
{a n o n-target |Λ non-target ≥Th non-target }
Middle lambda non-target Th is non-target direction distortion degree non-target A is a non-target direction distortion threshold non-target To satisfy lambda non-target ≥Th non-target Array set of constraints and a non-target ＝[a 1 ,a 2 ,...,a n ]，a n For the n-th array element to turn on the switch, the value is 0 or 1, step is the angle step length, M mod Is the modulation order.
Sequentially setting average side lobe threshold Th sidelobe Target direction distortion threshold Th target And a non-target direction distortion threshold Th non-target Solving the mathematical model under the three constraints to obtain an optimal array set, and randomly selecting array combinations from the optimal array set according to different array switching residence time to perform radiation switching of multiple array elements.
In the foregoing embodiment, in a further implementation manner, in step 12), the calculation method of the carrier phase difference, the carrier frequency difference, the code phase difference, and the modulation symbol phase difference of the local signal and the received signal of the tracking module is as follows:
a) Carrier phase difference and carrier frequency difference
Firstly regenerating a control channel or service channel local spread spectrum signal psi (k) subjected to random phase rotation in the ith coherent integration time, and then utilizing Doppler unbiased estimation
And phase deviation->
Initializing local carrier wave, down-converting received signal and spreading with local carrier waveThe signal psi (k) is subjected to coherent integration and then phase extraction operation, and a carrier phase residual is obtained, namely:
t in coh Is the coherent integration time, T is the sampling time, T is the conjugate operation, τ i For the phase deviation of the local spread spectrum signal and the received signal code in the ith coherent integration time, phi i For the carrier phase residual in the ith coherent integration time,
receiver carrier local oscillator frequency for the ith coherent integration time and initialized to +.>
Receiver carrier initial phase for the ith coherent integration time and initialized to +.>
mod (a, b) is a modulo b operation.
The carrier frequency residual error is obtained by carrying out differential operation on two adjacent coherent integration time carrier phase residual errors, namely:
finally utilizing the infinite length unit impulse response digital filter pair phi i And F i And (3) filtering to finish updating the carrier local oscillation frequency, namely:
ω 1 ＝1.275×B loop
in c 0 ，c 1 ，c 2 ，e 0 ，e 1 And e 2 Digital filter weight coefficient, omega of infinite length unit impulse response respectively 1 Is oscillation frequency, K is loop gain, B loop Is the loop bandwidth.
Receiver carrier initial phase within the ith coherent integration time
Updated to
b) Code phase deviation
First regenerating the random phase in the ith coherent integration timeThe rotated control channel or traffic channel local spread spectrum signal ψ (k) is then subject to lead and lag code correlation intervals t dll Operation to obtain the advanced local spread signal ψ (k+t) dll ) And lagging the local spread spectrum signal ψ (k-t dll ) Then uses the down-converted received baseband signal and advanced local spread spectrum signal psi (k+t) dll ) And lagging the local spread spectrum signal ψ (k-t dll ) After the coherent integration processing, the code phase deviation operation is completed, namely:
in the middle of
And->
The baseband signal and the advanced local spread spectrum signal ψ (k+t) are received in the ith coherent integration time, respectively dll ) And lagging the local spread spectrum signal ψ (k-t dll ) T is the integral result of (1) i code ＝1/f i code For the local spreading chip duration, f, in the ith coherent integration time i code For the i-th coherent integration time spreading code local oscillation frequency +.>
And->
Receiving noise and advancing local spread spectrum in the ith coherent integration time respectivelySignal ψ (k+t) dll ) And lagging the local spread spectrum signal ψ (k-t dll ) Is a result of integration of (a).
Finally, using infinite length unit impulse response digital filter pair tau i Filtering to update local oscillation frequency of spread spectrum code
ω 2 ＝1.886×B loop
Wherein K is loop gain, B loop For loop bandwidth omega 2 Is the oscillation frequency.
c) Modulation symbol phase deviation
First, a modulation symbol correlation interval t is set mll Respectively extracting [ iT ] coh -t mll ,(i+1)T coh -t mll ]，[iT coh ,(i+1)T coh ]，[iT coh +t mll ,(i+1)T coh +t mll ]And the received signals in the integral time are subjected to coherent integration processing with the local spread spectrum signals after the down-conversion of the three types of signals, namely:
in the method, in the process of the invention,
for the correlation integration time [ iT ] coh ,(i+1)T coh ]Integration of the internal received signal with the local signal, < >>
For the correlation integration time [ iT ] coh ,(i+1)T coh ]Receiving an integration result of noise and a local signal; />
For the correlation integration time [ iT ] coh -t mll ,(i+1)T coh -t mll ]Integration of the internal received signal with the local signal, < >>
For the correlation integration time [ iT ] coh ,(i+1)T coh ]Receiving an integration result of noise and a local signal; />
For the correlation integration time [ iT ] coh +t mll ,(i+1)T coh +t mll ]Integration of the internal received signal with the local signal, < >>
For the correlation integration time [ iT ] coh ,(i+1)T coh ]Receiving an integration result of noise and a local signal; t (T) i mod ＝1/f i mod For the local modulation symbol duration, f, in the ith coherent integration time i mod Is the information rate local oscillation frequency in the ith coherent integration time.
Then utilize adjacent
Polarity change by->
And->
Calculating modulation symbol phase deviation +.>
Namely: />
Finally, utilizing infinite length unit impulse response digital filter pair 
Filtering to update local oscillation frequency of information rate
ω 2 ＝1.886×B loop
Wherein K is loop gain, B loop For loop bandwidth omega 2 Is the oscillation frequency.
In addition to the above embodiment, the technical solution of the present invention further includes the following embodiments: the unmanned aerial vehicle anti-control anti-interception communication waveform generation device comprises a CRC coding module, a Turbo coding module, a rate matching module, a random modulation module, a code hopping spread spectrum module, a random phase rotation module, a random code rate generation module, a random modulation generation module, a code hopping generation module, a random phase generation module, a framing module, a random time hopping module, a time hopping generation module, a frequency hopping module, an up-conversion module, a digital-to-analog conversion module, an array switching module, an array element selection module, a delay amplitude addition weighting module and a multi-array element antenna, wherein the modules operate based on the method in the embodiment.
In addition to the above embodiment, the technical solution of the present invention further includes the following embodiments: the unmanned aerial vehicle anti-control anti-interception communication waveform receiving device comprises a down-conversion module, an analog-to-digital conversion module, a band-pass filter, a capturing module, a tracking module, a derandomizing rotation module, a code hopping despreading module, a baseband demodulation module, a Turbo decoding module and a CRC decoding module, wherein the modules operate based on the method in the embodiment.
In addition to the above embodiment, the technical solution of the present invention further includes the following embodiments: an unmanned aerial vehicle anti-control anti-interception communication system is provided, which comprises the waveform generation device and the waveform receiving device in the embodiment.
It should be noted that, within the scope of protection defined in the claims of the present invention, the above embodiments may be combined and/or expanded, and replaced in any manner that is logical from the above specific embodiments, such as the disclosed technical principles, the disclosed technical features or the implicitly disclosed technical features, etc.
As shown in fig. 3a to 3f, when the number N of antenna elements is 20, the average side lobe threshold is-15 dB, the target direction distortion threshold is 0.1, and the non-target direction distortion threshold is 0.5, fig. 3a shows 5 combinations in the optimal array set meeting the above constraint conditions according to the technical solution of the present invention, and fig. 3b shows antenna patterns of 5 optimal array combinations, as can be known from the figures, by using the electromagnetic wave dry-emission effect, a directional beam pointing to my receiver can be formed, and a low side lobe gain is formed in a non-cooperative direction, thereby reducing the airspace exposure energy and reducing the detection performance of the third party detection platform. Assuming that the modulation mode under the current modulation residence time is 8PSK, conducting array elements are randomly selected from the optimal array set and delay amplitude addition weight processing is performed, as shown in fig. 3c, the distortion degree of a modulation constellation received by a receiver on the side of a main lobe direction is low, as shown in fig. 3d, a third party detection platform dynamically changes the amplitude phase of a detection signal due to random switching of array combination in a non-cooperative direction, meanwhile, the phase of the detection signal is processed by a random phase rotation module, random rotation is also generated on the phase of the detection signal, serious distortion is caused, the third party detection platform cannot be effectively identified, as shown in fig. 3e, the third party detection platform in the direction of the receiver on the side of the main lobe direction is still subjected to phase rotation caused by random switching array combination although the influence of random switching is small, so that the adaptability and signal identification performance of the third party detection platform are reduced, and the receiver on the side of the main lobe direction can correctly receive and demodulate the detection signal due to the fact that the modulation constellation of the third party detection platform in the cooperative direction and the non-cooperative direction is distorted, and further demodulation of a follow-up signal is affected.
As shown in fig. 4a to 4B, the loop bandwidth B is equal to the loop gain k=1 loop =0.7 Hz, oscillation frequency ω 2 =0.892 Hz, modulation symbol correlation interval t mll Code correlation interval t=0.2 dll In the embodiment where the normalized carrier-to-noise ratio is 50db·hz, the local oscillation frequencies of the spread spectrum rate and the service information rate are non-homologous and are respectively set to 24Kbps and 5MHz, the initial doppler frequency jitter range is-5 KHz, and the doppler acceleration is 1KHz/s, as shown in fig. 4a, as the doppler frequency increases, the conventional Garder method tracking module cannot accurately track the doppler dynamic change, so that the doppler frequency tracking is unlocked; however, under the same condition, as shown in fig. 4b, the tracking module provided by the technical scheme of the invention can accurately track the dynamic change of the Doppler and has stronger Doppler resistance.
The units involved in the embodiments of the present invention may be implemented by software, or may be implemented by hardware, and the described units may also be provided in a processor. Wherein the names of the units do not constitute a limitation of the units themselves in some cases.
According to an aspect of embodiments of the present invention, there is provided a computer program product or computer program comprising computer instructions stored in a computer readable storage medium. The computer instructions are read from the computer-readable storage medium by a processor of a computer device, and executed by the processor, cause the computer device to perform the methods provided in the various alternative implementations described above.
As another aspect, the embodiment of the present invention also provides a computer-readable medium that may be contained in the electronic device described in the above embodiment; or may exist alone without being incorporated into the electronic device. The computer-readable medium carries one or more programs which, when executed by the electronic device, cause the electronic device to implement the methods described in the above embodiments.
The invention is not related in part to the same as or can be practiced with the prior art.
The foregoing technical solution is only one embodiment of the present invention, and various modifications and variations can be easily made by those skilled in the art based on the application methods and principles disclosed in the present invention, not limited to the methods described in the foregoing specific embodiments of the present invention, so that the foregoing description is only preferred and not in a limiting sense.
In addition to the foregoing examples, those skilled in the art will recognize from the foregoing disclosure that other embodiments can be made and in which various features of the embodiments can be interchanged or substituted, and that such modifications and changes can be made without departing from the spirit and scope of the invention as defined in the appended claims.
Claims (10)
1. The unmanned aerial vehicle anti-control anti-interception communication waveform design method is characterized by comprising the following steps of:
and setting a rate matching module parameter, a random modulation module parameter, a code hopping spread spectrum module parameter, a random phase rotation module parameter, a random time hopping module parameter, a frequency hopping module parameter and an array switching module parameter in a random process to generate an anti-control anti-interception communication waveform.
2. The unmanned aerial vehicle anti-control anti-intercept communication waveform design method of claim 1, further comprising the steps of, after generating the anti-control anti-intercept communication waveform: and receiving the anti-control anti-interception communication waveform at a receiving end.
3. The unmanned aerial vehicle anti-control anti-interception communication waveform design method according to any one of claims 1 or 2, wherein the introducing a random process sets a rate matching module parameter, a random modulation module parameter, a code hopping spread spectrum module parameter, a random phase rotation module parameter, a random time hopping module parameter, a frequency hopping module parameter and an array switching module parameter, comprising the following sub-steps:
s1, generating service information for transmitting user data in a time slot, obtaining a code sequence C through a CRC (cyclic redundancy check) coding module, a Turbo coding module and a rate matching module, wherein the rate matching module is controlled by a random code rate generating module to obtain code sequences with different code rates, and the random code rate generating module generates a real value l of ith code residence time through a chaotic random process i Obtaining a quantized value g by a quantizer i The code rate controller is directed to randomly select the currently used code rate from the code rate set;
s2, the code sequence C is subjected to a random modulation module to obtain a modulation sequence M, the random modulation module is controlled by a random modulation generation module, and the random modulation generation module generates a real value o of the ith modulation residence time by the chaotic random process in the step S1 i Obtaining quantized value p by quantizer i The modulation controller is guided to randomly select the currently used modulation type from the modulation set;
s3, the modulation sequence M is subjected to a code hopping spread spectrum module to obtain a code hopping spread spectrum sequence S, the code hopping spread spectrum module is controlled by a code hopping generation module, and the code hopping generation module generates a real value y of the ith code hopping residence time by the chaotic random process in the step S1 i Obtaining quantized value z by quantizer i The method comprises the steps that a code hopping controller is guided to randomly select a spreading code sequence with spreading gain and spreading rate which are currently used from a code hopping sequence set, the local oscillation frequencies of the spreading rate and the service information rate are non-homologous, and different spreading codes have good cross-correlation properties;
s4, the code hopping spread spectrum sequence S obtains a service information baseband signal D through a random phase rotation module, the random phase rotation module is controlled by a random phase generation module, and the random phase generation module generates a real value h of the ith phase residence time through a chaotic random process in the step S1 i As offset phase, namely:
D i ＝S i ×exp(-j complex 2πh i )
wherein D is i And S is i Respectively the firsti service baseband signals and code hopping spread spectrum signals of phase residence time j complex Is an imaginary number mark;
s5, using initial values of chaotic random processes used by a random code rate generation module, a random modulation generation module, a code hopping generation module and a random phase generation module in a time slot as communication waveform characteristic parameter indication information so as to guide a receiving end to process a service baseband signal;
s6, after the synchronous information and the communication waveform characteristic parameter indication information are combined through a formatting module, a control baseband signal is obtained through a random modulation module, a code hopping spread spectrum module and a random phase rotation module by adopting a fixed pattern in sequence, wherein the spread spectrum rate and the local oscillation frequency of the control information rate are homologous;
s7, after the service baseband signal and the control baseband signal pass through a framing module, a time slot radiation signal F is obtained, radiation time jitter is generated through a random time hopping module, the random time hopping module is controlled by a time hopping generating module, and the time hopping generating module generates a real value h of the ith time hopping residence time through a chaotic random process in the step S1 i Obtaining quantized value c by quantizer i The controller randomly selects the currently used radiation starting time from the time hopping set;
S8, the baseband signal TH output by the random time hopping module is randomly selected by the array switching module, the array element combination conducted in the one-dimensional linear array switching residence time is conducted, then the delay amplitude adding weighting module in the conducting array element channel respectively carries out delay and amplitude adding weighting processing on the baseband signal TH, so that the main beam of the synthesized signal is aligned to the receiver, the array switching module is controlled by the array selection module, and the array selection module generates a real value v of the ith array switching residence time in the chaotic random process in the step S1 i Obtaining a quantized value r by a quantizer i The array element selection controller is guided to randomly select the currently used array element combination from the optimal array set;
s9, signal vector U= [ U ] of the multipath signals processed by the delay amplitude-phase addition weighting module 1 ,U 2 ,…,U N ]Through up-conversion and D/A conversion, the antenna with multiple array elements is usedRadiating to a channel, wherein N is the number of array elements of the multi-array element antenna, the up-converted carrier frequency is generated by a frequency hopping module, and the frequency hopping module generates a real value x of the ith frequency residence time by the chaotic random process in the step S1 i Obtaining quantized value p by quantizer i To direct the frequency hopping controller to randomly select the currently used frequency from the set of frequencies.
4. The unmanned aerial vehicle anti-control anti-interception communication waveform design method according to claim 3, wherein the receiving the anti-control anti-interception communication waveform at the receiving end comprises the following sub-steps:
s10, the receiving end carries out down-conversion treatment on the received signal, and then carries out treatment on the received signal through an analog-to-digital conversion module and a band-pass filter to obtain a digital intermediate frequency signal IF, namely:
where IF (k) is the kth sample time intermediate frequency signal, ψ (k) and n (k) are the kth sample time baseband signal and Gaussian noise,
for receiving intermediate frequency, θ is phase deviation, and T is sampling time;
s11, the capturing module obtains a local synchronous baseband signal according to known synchronous information and a baseband processing mode of a fixed pattern, gathers samples according to known frequency, simultaneously carries out parallel down-conversion processing on an intermediate frequency signal, correlates with the local synchronous baseband signal and carries out threshold detection, and the frequency and the time corresponding to the maximum correlation peak value which crosses the threshold are estimated as the frequency hopping frequency and the starting time adopted by the current time slot; then the local synchronous baseband signal is utilized to eliminate the information phase of the receiving synchronous segment, and the length L after the information phase is eliminated is obtained 0 Is a reception synchronization signal y of:
In the middle of
For Doppler frequency, n' (k) is baseband Gaussian noise, y (k) is a k-th sampling time signal of y, and x is conjugate operation; sequentially carrying out phase difference, weighting and cumulative average operation on the sampling signals at the front and rear moments on y to finish the minimum mean square unbiased estimation of delta f>
Namely:
in the middle of
For Doppler unbiased estimation, w (·) is a weighted window function, R (·) is a time domain autocorrelation function, arg (·) is a phase-taking operation; finally, using Doppler unbiased estimation +.>
Offset the influence of Doppler linear phase and estimate the phase deviation
Namely:
s12, the tracking module performs Doppler unbiased estimation according to the output of the capturing module
And phase deviation->
Generating a local initial carrier, respectively regenerating a control channel and a service channel local spread spectrum signal after random phase rotation based on a known fixed pattern and a baseband processing mode of communication waveform characteristic parameter indication information provided by the control channel, and sequentially adjusting a receiver carrier local oscillation frequency, a spread spectrum code rate local oscillation frequency and an information rate local oscillation frequency by taking a carrier phase difference, a carrier frequency difference, a code phase difference and a modulation symbol phase difference of a local signal and a received signal as inputs in a mode of carrier loop, code loop and modulation symbol loop linkage to complete carrier phase synchronization, symbol synchronization and modulation symbol synchronization of the received control signal and the service signal;
S13, the synchronous control baseband signal output by the tracking module is sequentially subjected to a random phase rotation decoding module, a code hopping despreading module and a baseband demodulation module according to a known fixed pattern processing mode to obtain waveform characteristic indicating parameters;
s14, the synchronized service baseband signal output by the tracking module is processed according to the communication waveform characteristic parameter indicating information analyzed by the control baseband signal, and service information is recovered through the de-random phase rotation module, the code hopping de-spreading module, the baseband demodulation module, the Turbo decoding module and the CRC decoding module in sequence.
5. The unmanned aerial vehicle anti-control anti-interception communication waveform design method of claim 3, wherein in step S8, the optimal array set satisfies the following constraint conditions:
s81, minimizing array side lobe level constraint conditions:
{a sidelobe |P sidelobe ≤Th sidelobe }
S＝{θ|θ min ≤θ≤θ d -Δθ/2∪θ d +Δθ/2≤θ≤θ max }
p in the formula sidelobe Average sidelobe level Th of conducting array set after amplitude addition weighting sidelobe For the average side lobe threshold, F (θ) is the antenna gain with direction θ, S is the side lobe angle range, S num For the number of samples of the angle range of the side lobe, theta min For the lower limit of the angle range of the side lobe, theta max For the upper limit of the range of the side lobe angle, delta theta is 3dB main lobe width, f n (θ) is the n-th array element pattern, θ d For the communication target direction, a sidelobe To meet P sidelobe ≤Th sidelobe Array set of constraints and a= [ a ] 1 ,a 2 ,…,a n ]，a n The n-th array element is conducted with a switch, the value is 0 or 1, lambda is the carrier wavelength, and d is the distance between adjacent array elements of the linear array;
minimizing target directional distortion constraints:
{a target |Λ target ≤Th target }
Λ target ＝Σ|D i F(θ d )-D i |
middle lambda target Th is the target direction distortion degree target For the target direction distortion threshold, D i An ith modulation constellation point, a, for a traffic information modulated signal D target To satisfy lambda target ≤Th target Array set of constraints and a target ＝[a 1 ,a 2 ,…,a n ]，a n The switch is turned on for the nth array element, and the value is 0 or 1;
maximizing non-target directional distortion constraints:
{a non-target |Λ non-target ≥Th non-target }
middle lambda non-target Th is non-target direction distortion degree non-target A is a non-target direction distortion threshold non-target To satisfy lambda non-target ≥Th non-target Array set of constraints and a non-target ＝[a 1 ,a 2 ,...,a n ]，a n For the n-th array element to turn on the switch, the value is 0 or 1, step is the angle step length, M mod Is the modulation order;
s82, setting an average sidelobe threshold Th sidelobe Target direction distortion threshold Th target And a non-target direction distortion threshold Th non-target Solving the mathematical model under the three constraint conditions in the step S81 to obtain an optimal array set, and randomly selecting array combinations from the optimal array set according to different array switching residence time to perform radiation switching of multiple array elements.
6. The unmanned aerial vehicle anti-control anti-interception communication waveform design method of claim 4, wherein in step S12, the calculation methods of the carrier phase difference, carrier frequency difference, code phase difference and modulation symbol phase difference of the local signal and the received signal are as follows:
carrier phase difference and carrier frequency difference: first regenerating the control channel or service channel after random phase rotation in the ith coherent integration timeThe local spread spectrum signal ψ (k) is then estimated using doppler unbiased
And phase deviation->
Initializing a local carrier, performing coherent integration on a received signal after down-conversion processing and a local spread spectrum signal psi (k), and performing phase extraction operation to obtain a carrier phase residual, namely:
t in coh Is the coherent integration time, T is the sampling time, T is the conjugate operation, τ i For the phase deviation of the local spread spectrum signal and the received signal code in the ith coherent integration time, phi i For the carrier phase residual in the ith coherent integration time,
receiver carrier local oscillator frequency for the ith coherent integration time and initialized to +.>
Receiver carrier initial phase for the ith coherent integration time and initialized to +.>
mod (a, b) is a modulo b operation;
the carrier frequency residual error is obtained by carrying out differential operation on two adjacent coherent integration time carrier phase residual errors, namely:
Finally utilizing the infinite length unit impulse response digital filter pair phi i And F i And (3) filtering to finish updating the carrier local oscillation frequency, namely:
ω 1 ＝1.275×B loop
in c 0 ，c 1 ，c 2 ，e 0 ，e 1 And e 2 Digital filter weight coefficient, omega of infinite length unit impulse response respectively 1 Is oscillation frequency, K is loop gain, B loop Is loop bandwidth;
receiver carrier initial phase within the ith coherent integration time
Updated to
Code phase bias:
firstly regenerating a control channel or service channel local spread spectrum signal psi (k) subjected to random phase rotation in the ith coherent integration time, and then performing lead and lag code correlation interval t on the psi (k) dll Operation to obtain the advanced local spread signal ψ (k+t) dll ) And lagging the local spread spectrum signal ψ (k-t dll ) Then uses the down-converted received baseband signal and advanced local spread spectrum signal psi (k+t) dll ) And lagging the local spread spectrum signal ψ (k-t dll ) After the coherent integration processing, the code phase deviation operation is completed, namely:
in the middle of
And->
The baseband signal and the advanced local spread spectrum signal ψ (k+t) are received in the ith coherent integration time, respectively dll ) And lagging the local spread spectrum signal ψ (k-t dll ) T is the integral result of (1) i code ＝1/f i code For the local spreading chip duration, f, in the ith coherent integration time i code For the i-th coherent integration time spreading code local oscillation frequency,/and >
And->
The noise and the early local spread spectrum signal ψ (k+t) are received in the ith coherent integration time, respectively dll ) And lagging the local spread spectrum signal ψ (k-t dll ) Is a result of integration of (a);
finally, using infinite length unit impulse response digital filter pair tau i Filtering to update local oscillation frequency of spread spectrum code
ω 2 ＝1.886×B loop
Wherein K is loop gain, B loop For loop bandwidth omega 2 Is the oscillation frequency;
modulation symbol phase deviation: first, a modulation symbol correlation interval t is set mll Respectively extracting [ iT ] coh -t mll ,(i+1)T coh -t mll ]，[iT coh ,(i+1)T coh ]，[iT coh +t mll ,(i+1)T coh +t mll ]And the received signals in the integral time are subjected to coherent integration processing with the local spread spectrum signals after the down-conversion of the three types of signals, namely:
in the method, in the process of the invention,
for the correlation integration time [ iT ] coh ,(i+1)T coh ]The result of integrating the internal received signal with the local signal,
for the correlation integration time [ iT ] coh ,(i+1)T coh ]Receiving an integration result of noise and a local signal; />
For the correlation integration time [ iT ] coh -t mll ,(i+1)T coh -t mll ]Integration of the internal received signal with the local signal, < >>
For the correlation integration time [ iT ] coh ,(i+1)T coh ]Receiving an integration result of noise and a local signal; />
For the correlation integration time [ iT ] c o h +t mll ,(i+1)T coh +t mll ]Integration of the internal received signal with the local signal, < >>
For the correlation integration time [ iT ] coh ,(i+1)T coh ]Receiving an integration result of noise and a local signal; t (T) i mod ＝1/f i mod For the local modulation symbol duration, f, in the ith coherent integration time i mod The local oscillation frequency is the information rate in the ith coherent integration time; />
Then utilize adjacent
Polarity change by->
And->
Calculating modulation symbol phase deviation +.>
Namely:
finally, utilizing infinite length unit impulse response digital filter pair
Filtering to update local oscillation frequency of information rate
ω 2 ＝1.886×B loop
Wherein K is loop gain, B loop For loop bandwidth omega 2 Is the oscillation frequency.
7. The unmanned aerial vehicle anti-control anti-interception communication waveform design method of claim 3, wherein,
in step S1, the sub-steps are included: the range of the code rate set is {0.1,0.2,0.3,0.4,0.5,0.6,0.7,0.8}, the chaotic random process adopts Logistic mapping, and the quantizer adopts a nonlinear quantization equation:
in the formula, the quantized value g i =j, q is the number of samples of the code rate set;
in step S2, the sub-steps are included: the modulation set is {2PSK,4PSK,8PSK,16PSK }, and the nonlinear quantization equation in the step S1 is adopted to satisfy the quantization value p i =j, j=0, 1..q-1, q is the number of modulation set samples;
in step S3, the sub-steps are included: the non-linear quantization equation in the step S1 is adopted to satisfy the quantization value z i =j, j=0, 1..q-1, q is the number of samples of the set of hopping sequences;
in step S7, the sub-steps are included: the non-linear quantization equation in the step S1 is adopted to satisfy the quantization value c i =j, j=0, 1..q-1, q is the number of samples of the time-hopped set, and the time-hopped dwell time is one slot time length;
in step S8, the sub-steps are included: the non-linear quantization equation in the step S1 is adopted to satisfy the quantization value r i =j, j=0, 1..q-1, q is the optimal array set number of samples;
in step S9, the sub-steps are included: the non-linear quantization equation in the step S1 is adopted to satisfy the quantization value p i =j, j=0, 1..q-1, q is the number of samples of the frequency set and the frequency dwell time is one slot time length.
8. An unmanned aerial vehicle anti-control anti-interception communication waveform generation device is characterized by comprising a CRC coding module, a Turbo coding module, a rate matching module, a random modulation module, a code hopping spread spectrum module, a random phase rotation module, a random code rate generation module, a random modulation generation module, a code hopping generation module, a random phase generation module, a framing module, a random time hopping module, a time hopping generation module, a frequency hopping module, an up-conversion module, a digital-to-analog conversion module, an array switching module, an array element selection module, a delay amplitude addition weight module and a multi-array element antenna, wherein the modules operate based on the method as claimed in claim 3.
9. An unmanned aerial vehicle anti-control anti-interception communication waveform receiving device is characterized by comprising a down-conversion module, an analog-to-digital conversion module, a band-pass filter, a capturing module, a tracking module, a derandomizing rotation module, a code hopping despreading module, a baseband demodulation module, a Turbo decoding module and a CRC decoding module, wherein the modules operate based on the method as claimed in claim 4.
10. An unmanned aerial vehicle anti-control anti-interception communication system, comprising the waveform generation device and the waveform receiving device as claimed in claims 8 and 9.
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