CN115593250A

CN115593250A - Constant-power wireless charging system

Info

Publication number: CN115593250A
Application number: CN202211427861.9A
Authority: CN
Inventors: 林智声; 任耀华; 蔡劭钧; 麦沛然; 马许愿
Original assignee: University of Macau
Current assignee: University of Macau
Priority date: 2022-11-15
Filing date: 2022-11-15
Publication date: 2023-01-13

Abstract

The application provides a constant-power wireless charging system, and relates to the technical field of wireless charging. The method comprises the steps that under the condition that no additional conversion layer is needed, a compensation module and a wireless power transmission module are introduced into a resonance compensation circuit on a primary side, and a pulse density modulation control wave is connected to a second input end of a semi-active rectifier; in this way, the output current output by the resonance compensation circuit on the primary side to the wireless charging receiving device is unrelated to the mutual inductance between the wireless power transmission module and the resonance compensation circuit on the secondary side, and the value of the output current can be controlled by adjusting the parameter value of each component in the compensation module; and the duty ratio of the output alternating-current voltage output to the wireless charging receiving device by the resonance compensation circuit on the primary side is modulated in a pulse density modulation control wave period, so that the aim of adjusting an equivalent alternating-current load is fulfilled, wireless constant-power charging is realized, the system efficiency is obviously improved, and the complexity and the cost of the system are reduced.

Description

Constant-power wireless charging system

Technical Field

The application relates to the technical field of wireless charging, in particular to a constant-power wireless charging system.

Background

Inductive Power Transfer (IPT) technology uses Inductive coupling to wirelessly Transfer energy from a Power supply to a load through a magnetic field, and improves convenience, reliability and safety of Power Transfer by virtue of non-contact characteristics of the Inductive Power Transfer. Therefore, the inductive power transfer technology is suitable as a power transfer solution in the case of medical implants, portable devices, electric bicycles, and electric automobiles. In the IPT technical field, the research of wireless charging such as constant current, constant voltage and constant power is generally related; compared with the Constant Current (CC) charging or Constant Voltage (CV) charging which is dominant in the conventional battery charging scheme, constant Power (CP) charging can provide a faster charging rate, and can also alleviate the problem of battery aging.

Currently, some constant power wireless charging schemes propose that an additional direct current Converter (DC to DC Converter, abbreviated as DC-DC) can be added directly at the input and/or output of the inductive power transfer Converter to regulate the output current and voltage, thereby directly modulating the output power.

However, this significantly reduces system efficiency and increases system complexity and cost due to the need to add additional translation layers.

Disclosure of Invention

The present application aims to provide a constant power wireless charging system to solve the problems of the prior art that the system efficiency is reduced and the system complexity and cost are increased due to the need of adding an additional conversion layer.

In order to achieve the above purpose, the technical solutions adopted in the embodiments of the present application are as follows:

in a first aspect, an embodiment of the present application provides a constant-power wireless charging system, including: the wireless charging system comprises a wireless charging transmitting device, a wireless charging receiving device and a control device;

wherein, wireless transmitting device that charges includes: a full-bridge voltage source inverter, a primary side resonance compensation circuit; the wireless charging receiving device comprises: a resonance compensation circuit on the secondary side, a semi-active rectifier;

the resonance compensation circuit on the primary side includes: the wireless power transmission device comprises a compensation module and a wireless power transmission module;

the input end of the full-bridge voltage source inverter is used for accessing a direct current voltage source, and the full-bridge voltage source inverter is used for converting the direct current voltage source into high-frequency alternating current voltage;

the output end of the full-bridge voltage source inverter is connected with the input end of the compensation module, the output end of the compensation module is connected with the input end of the wireless power transmission module, the compensation module is used for compensating the high-frequency alternating voltage output by the output end of the full-bridge voltage source inverter so as to output the compensated high-frequency alternating voltage, and the wireless power transmission module is used for wirelessly transmitting the compensated high-frequency alternating voltage;

the output end of the wireless power transmission module is electromagnetically connected with the secondary side resonance compensation circuit, and the secondary side resonance compensation circuit is used for receiving the compensated high-frequency alternating-current voltage;

the output end of the resonance compensation circuit on the secondary side is connected with the first input end of the semi-active rectifier, the second input end of the semi-active rectifier is connected with the control device, the control device is used for outputting pulse density modulation control waves to the semi-active rectifier, the semi-active rectifier is used for rectifying the compensated high-frequency alternating voltage under the control of the pulse density modulation control waves to obtain direct current voltage, and the output end of the semi-active rectifier transmits the direct current voltage to a load to be charged.

Optionally, the compensation module comprises: the first inductor, the first capacitor and the second capacitor;

one end of the first inductor is connected with the output end of the full-bridge voltage source inverter, and the other end of the first inductor is respectively connected with one end of the first capacitor and one end of the second capacitor;

the other end of the first capacitor is respectively connected with the output end of the full-bridge voltage source inverter and the input end of the wireless power transmission module;

the other end of the second capacitor is connected with the the input end of the wireless power transmission module is connected.

Optionally, the wireless power transmission module includes: a transmitting coil;

one end of the transmitting coil is connected with the other end of the second capacitor, and the other end of the transmitting coil is connected with the other end of the first capacitor;

the transmitting coil is electromagnetically connected with the resonance compensation circuit on the secondary side.

Optionally, the resonance compensation circuit of the secondary side includes: a third capacitor, a receiving coil;

the receiving coil is electromagnetically connected with the transmitting coil;

one end of the third capacitor is connected with one end of the receiving coil, and the other end of the third capacitor is connected with the first input end of the semi-active rectifier;

and the other end of the receiving coil is connected with the first input end of the semi-active rectifier.

Optionally, the semi-active rectifier comprises: the first diode, the second diode, the first switch tube and the second switch tube;

one end of the first diode is connected with one end of the second diode and one end of the load to be charged respectively, and the other end of the first diode is connected with the other end of the third capacitor and the first end of the first switch tube respectively;

the second end of the first switching tube is respectively connected with the second end of the second switching tube and the other end of the load to be charged;

the first end of the second switch tube is respectively connected with the other end of the second diode and the other end of the receiving coil;

and the third end of the first switch tube and the third end of the second switch tube are both connected with the control device so as to access the pulse density modulation control wave, and the pulse density modulation control wave controls the on-off states of the first switch tube and the second switch tube.

Optionally, the system further comprises: a filter capacitor;

one end of the filter capacitor is connected with one end of the first diode, one end of the second diode and one end of the load to be charged respectively, and the other end of the filter capacitor is connected with the second end of the first switch tube, the second end of the second switch tube and the other end of the load to be charged respectively;

the filter capacitor is used for filtering the direct-current voltage output by the output end of the semi-active rectifier to obtain a filtered direct-current voltage, and transmitting the filtered direct-current voltage to a load to be charged.

Optionally, the full-bridge voltage source inverter comprises: a third switching tube, a fourth switching tube, a fifth switching tube and a sixth switching tube;

the first end of the third switching tube and the first end of the fourth switching tube are both used for being connected to the positive electrode of the direct-current voltage source, and the second end of the third switching tube is connected with the first end of the fifth switching tube and one end of the first inductor respectively;

the second end of the fourth switching tube is respectively connected with the first end of the sixth switching tube and the other end of the first capacitor;

the second end of the fifth switching tube and the second end of the sixth switching tube are both used for being connected to the negative electrode of the direct-current voltage source;

the third end of the third switching tube and the third end of the sixth switching tube are both used for accessing a first switching sequence waveform, the third end of the fourth switching tube and the third end of the fifth switching tube are both used for accessing a second switching sequence waveform, and the third switching tube, the fourth switching tube, the fifth switching tube and the sixth switching tube are used for performing high-frequency conversion on the direct-current voltage source under the control of the first switching sequence waveform and the second switching sequence waveform to obtain high-frequency alternating-current voltage.

Optionally, the resonant compensation circuit on the primary side and the resonant compensation circuit on the secondary side form a loosely coupled transformer having an Inductor-Capacitor-Series (LCC-S) compensation topology.

Optionally, when the resonance compensation circuit on the primary side and the resonance compensation circuit on the secondary side form a T equivalent model of a loose coupling transformer with an LC-series LCC-S compensation topology, the resonance frequency satisfies

Under the condition, the output current of the resonance compensation circuit on the secondary side is

Wherein, ω is _R Is the resonant frequency, L _M Is mutual inductance, L _lp Is primary side leakage inductance, C _P1” Is a first capacitor C _P1 One of two capacitors in parallel, C _P2 Is a second capacitor, I _S To output an alternating current i _S Phasor of the fundamental component of (g) _m,IPT Is transconductance, V _P For inputting an alternating voltage v _P Phasor of the fundamental component of (a).

Optionally, when the phase angle between the input ac voltage and the input ac current in the loosely coupled transformer is greater than or equal to a minimum phase angle, the full-bridge voltage source inverter operates under a zero-voltage switching characteristic condition, wherein the minimum phase angle is formulated by

Calculating to obtain;

wherein f is _R Is a resonant frequency, C _oss,u And C _oss,l The parasitic output capacitors of the third switching tube and the sixth switching tube are respectively; vin is a DC voltage source connected to the third and sixth switching tubes, I _p,rms Is I _P Root mean square value of (d).

Optionally, the semi-active rectifier is further configured to adjust an equivalent ac load by a control factor of the semi-active rectifier.

Optionally, the turn ratio of the transmit coil to the receive coil is 1:1.

Optionally, the pulse density modulation control wave is generated by the control device according to a ratio between an actual output power and a rated power of the load to be charged.

Optionally, the control device comprises: a sensor circuit, a microcontroller unit and a zero-crossing detector;

the output end of the sensor circuit is connected with the first input end of the microcontroller unit, and the sensor circuit is used for collecting output current and output voltage and outputting the collected output current and output voltage to the microcontroller unit;

the output end of the zero-crossing detector is connected with the second input end of the microcontroller unit, and the zero-crossing detector is used for acquiring the output alternating current of the resonance compensation circuit on the secondary side, generating a clock signal according to the output alternating current and outputting the clock signal to the microcontroller unit;

the output end of the microcontroller unit is connected with the second input end of the semi-active rectifier, and the microcontroller unit is used for generating the pulse density modulation control wave according to the output current, the output voltage and the clock signal, and outputting the pulse density modulation control wave to a first switch tube and a second switch tube in the semi-active rectifier, so that the first switch tube and the second switch tube rectify the compensated high-frequency alternating-current voltage under the control of the pulse density modulation control wave to obtain direct-current voltage.

Optionally, the microcontroller unit comprises: the device comprises a multiplier, a divider, a proportional-integral controller and a modulator;

the output end of the sensor circuit is connected with the input end of the multiplier, the output end of the multiplier is connected with the input end of the divider, and the multiplier is used for receiving the output current and the output voltage acquired by the sensor circuit, calculating to obtain the current output power according to the output current and the output voltage, and transmitting the current output power to the divider;

the output end of the divider is connected with the input end of the proportional-integral controller, and the divider is used for calculating to obtain a first coefficient according to the current output power and a preset reference rated output power and transmitting the first coefficient to the proportional-integral controller;

the output end of the proportional-integral controller is connected with the first input end of the modulator, and the proportional-integral controller is used for correcting the first coefficient to obtain a control factor and transmitting the control factor to the modulator;

the second input end of the modulator is connected with the output end of the zero-crossing detector, and the modulator is used for receiving the clock signal output by the output end of the zero-crossing detector and generating the pulse density modulation control wave according to the clock signal and the control factor.

The beneficial effect of this application is:

the embodiment of the application provides a constant-power wireless charging system, which is mainly characterized in that an additional compensation module and a wireless power transmission module are introduced into a primary side resonance compensation circuit, and a pulse density modulation control wave is connected to a second input end of a semi-active rectifier; thus, the output current I output by the primary side resonance compensation circuit to the wireless charging receiving device can be enabled _S Independent of mutual inductance M between the wireless power transmission module and the secondary side resonance compensation circuit, so that the output current I is _S The value of the compensation module can be controlled by adjusting the parameter value of each component in the compensation module; and in a pulse density modulated control wave period T _PDM Output alternating voltage v output from resonance compensation circuit internally modulating primary side to wireless charging receiving device _S So as to adjust the equivalent alternating current load R at the side of the wireless charging receiving device _eq To do (1)The method (1); and then the output power of the wireless charging transmitting device to the wireless charging receiving device is constant power, namely, the wireless constant power charging is realized under the condition that additional conversion layers such as a DC-DC converter, an auxiliary SCC circuit or a wireless feedback communication circuit are not needed, the system efficiency is obviously improved, and the system complexity and the cost are reduced.

Drawings

In order to more clearly illustrate the technical solutions of the embodiments of the present application, the drawings that are required to be used in the embodiments will be briefly described below, it should be understood that the following drawings only illustrate some embodiments of the present application and therefore should not be considered as limiting the scope, and for those skilled in the art, other related drawings can be obtained from the drawings without inventive effort.

Fig. 1 is a schematic diagram of a charging curve of constant current charging and constant power charging in the prior art;

fig. 2 is a schematic structural diagram of a frame of a constant-power wireless charging system according to an embodiment of the present disclosure;

fig. 3 is a first schematic circuit diagram of a constant-power wireless charging system according to an embodiment of the present disclosure;

fig. 4 is a second schematic circuit structure diagram of a constant-power wireless charging system according to an embodiment of the present disclosure;

fig. 5 is a schematic diagram of an operating waveform, a first switching sequence waveform and a second switching sequence waveform of a full-bridge voltage source inverter according to an embodiment of the present application;

fig. 6 is a schematic diagram of a T equivalent model of an IPT converter with an LCC-S compensation topology provided by an embodiment of the present application;

fig. 7 is a schematic diagram illustrating a relationship between a first inductance, an equivalent ac load, and an angle difference according to an embodiment of the present application;

fig. 8 is a schematic waveform diagram of pulse density modulation control waves of a first switching tube and a second switching tube in a semi-active rectifier according to an embodiment of the present disclosure;

fig. 9 is a schematic diagram of an M-equivalent circuit model of an IPT converter of an LCC-S compensation topology provided by an embodiment of the present application;

FIG. 10 is a schematic diagram of curves of different internal resistances of the battery corresponding to Po/Po, rate and η/η Optimal under various control factors α according to an embodiment of the present disclosure;

fig. 11 is a schematic diagram of control factors α corresponding to different internal resistances RL of the battery and ratios (Req/Req, opt) of equivalent load/optimal equivalent load with and without semi-active rectifier control according to an embodiment of the present application;

fig. 12 is a schematic structural diagram of a proposed control device according to an embodiment of the present application;

fig. 13 is a schematic waveform diagram illustrating the charging start of the constant power charger according to the embodiment of the present application;

fig. 14 is a schematic diagram of waveforms in charging of a constant power charger according to an embodiment of the present application;

fig. 15 is a schematic waveform diagram of the constant power charger provided by the embodiment of the present application when charging is finished;

fig. 16 is a diagram illustrating DC-to-DC transmission efficiency and output power during charging according to an embodiment of the present disclosure;

fig. 17 shows loads R with different output voltages and output currents according to an embodiment of the present application _L A schematic of the measured values of (a);

FIG. 18 is a schematic diagram of operating electrical points in alignment and misalignment provided by an embodiment of the present application;

fig. 19 is a schematic diagram of DC-to-DC transmission efficiency and output power measured with alignment and misalignment during charging according to an embodiment of the present disclosure.

Icon: 100-power wireless charging system; 101-a wireless charging transmitting device; 102-a wireless charging receiving means; 103-a control device; 104-full bridge voltage source inverter; 105-a resonance compensation circuit on the primary side; 106-resonance compensation circuit on the secondary side; 107-semi-active rectifier; 108-a compensation module; 109-wireless power transfer module.

Detailed Description

In order to make the purpose, technical solutions and advantages of the embodiments of the present application clearer, the technical solutions in the embodiments of the present application will be clearly and completely described below with reference to the drawings in the embodiments of the present application, and it should be understood that the drawings in the present application are for illustrative and descriptive purposes only and are not used to limit the scope of protection of the present application. Additionally, it should be understood that the schematic drawings are not necessarily drawn to scale.

In addition, the described embodiments are only a part of the embodiments of the present application, and not all of the embodiments. The components of the embodiments of the present application, generally described and illustrated in the figures herein, can be arranged and designed in a wide variety of different configurations. Thus, the following detailed description of the embodiments of the present application, presented in the accompanying drawings, is not intended to limit the scope of the claimed application, but is merely representative of selected embodiments of the application. All other embodiments, which can be derived by a person skilled in the art from the embodiments of the present application without making any creative effort, shall fall within the protection scope of the present application.

It should be noted that in the embodiments of the present application, the term "comprising" is used to indicate the presence of the features stated hereinafter, but does not exclude the addition of further features.

First, before the technical solutions provided in the present application are explained in detail, the related background related to the present application will be briefly explained.

In the IPT technology field, constant Current and Constant Voltage (CC-CV) charging curves are commonly used for wireless charging research. Generally, constant current charging dominates the charging phase, the battery is charged with constant current until the charging voltage reaches a predetermined threshold, and then will switch to a constant voltage charging phase. Referring to fig. 1, wherein fig. 1 (a) is a schematic diagram of a charging curve of constant current charging in the prior art, it can be seen from fig. 1 (a) that when the battery voltage is low, the output power is low due to the constant current charging, and the charger provides the maximum output power only during the transition period from the constant current charging to the constant voltage charging phase, i.e., immediately before the end of the constant current charging phase. To obtain a better charge rate, the constant current charging phase typically employs a higher charging current, but at the expense of a faster battery degradation rate.

Therefore, in order to utilize the maximum Power transfer capability of the charging system, and thus increase the charging rate, the output voltage and current may be modulated throughout the charging process to achieve a consistent output Power process, referred to as Constant Power (CP) charging. FIG. 1 ((b) is a schematic view of a charging curve for constant power charging in the prior art, as shown in FIG. 1 (b), constant output power can be achieved by decreasing output current as battery voltage increases, the charger can achieve both faster charging rate with the same design and reduced overall size and material cost by replacing constant current charging with constant power charging, possibly due to factors such as component characteristics and cooling system, etc. furthermore, constant power charging is a fast charging method that helps alleviate battery aging issues.

Typically, in order to achieve load independent transfer characteristics, IPT converters are usually designed to operate at some fixed frequency to match the compensation network. Unfortunately, even if a constant current or constant voltage output can be achieved, the output power of the IPT converter varies with load variations, resulting in failure to achieve a constant power charging curve.

Currently, some direct solutions for wireless constant power charging implementations are simply to add an additional DC-DC power converter at the input and/or output of the IPT converter to regulate the output current and voltage, thereby directly modulating the output power.

However, this can significantly reduce system efficiency and increase system complexity and cost due to the additional translation layer.

Based on the above problems, the present application provides a constant power wireless charging system without adding an additional DC-DC converter or a switch control capacitor circuit to implement constant power charging, which saves cost and reduces control complexity.

The structure of the constant-power wireless charging system provided in the present application will be described in detail by a plurality of embodiments as follows.

Fig. 2 is a schematic structural diagram of a frame of a constant-power wireless charging system according to an embodiment of the present disclosure; as shown in fig. 2, the constant power wireless charging system 100 includes: a wireless charging transmitting device 101, a wireless charging receiving device 102 and a control device 103.

Wherein, wireless charging emitter 101 includes: a full-bridge Voltage Source Inverter (VSI) 104, a primary side resonance compensation circuit 105, and a wireless charging receiving apparatus 102 including: a secondary side resonance compensation circuit 106, and a Semi-Active Rectifier (SAR) 107.

It should be noted that the primary side resonance compensation circuit 105 and the secondary side resonance compensation circuit 106 in the constant power wireless charging system proposed in the present application need to be considered as a whole to derive the whole design solution, and the whole design solution is only applicable to the LCC-S compensation topology.

In the present embodiment, the resonance compensation circuit 105 on the primary side and the semi-active rectifier 107 in the wireless charging receiver are core modules of the present application.

With continued reference to fig. 2, the primary-side resonance compensation circuit 105 includes: compensation module 108 and wireless the power transmission module 109. Wherein, the reactive power caused by the wireless power transmission module 109 can be compensated by the compensation module 108, and the compensated ac form electric energy can be wirelessly transmitted by the wireless power transmission module 109.

The power of the constant-power wireless charging system 100 is provided by a dc voltage source Vin, that is, the input end of the full-bridge voltage source inverter 104 is connected to the dc voltage source Vin, the full-bridge voltage source inverter 104 operates at a fixed frequency, and the dc voltage source Vin can be converted into a high-frequency ac voltage by the full-bridge voltage source inverter 104.

The output end of the full-bridge voltage source inverter 104 is connected with the input end of the compensation module 108, the output end of the compensation module 108 is connected with the input end of the wireless power transmission module 109, the compensation module 108 is used for compensating the high-frequency alternating-current voltage output by the output end of the full-bridge voltage source inverter 104 to output the compensated high-frequency alternating-current voltage, and the wireless power transmission module 109 is used for wirelessly transmitting the compensated high-frequency alternating-current voltage, namely, the high-frequency alternating-current voltage compensated by the wireless power transmission module 109 and the resonance compensation circuit 106 on the secondary side is transmitted to the wireless charging receiving device 102 side.

The output end of the wireless power transmission module 109 is electromagnetically connected to the secondary-side resonance compensation circuit 106, and the secondary-side resonance compensation circuit 106 is configured to receive the compensated high-frequency ac voltage.

Meanwhile, the output end of the resonance compensation circuit 106 on the secondary side is connected to the first input end of the semi-active rectifier 107, the second input end of the semi-active rectifier 107 is connected to the control device 103, the control device 103 is configured to output a Pulse Density Modulation (Pulse Density Modulation, PDM for short) control wave to the semi-active rectifier 107, the semi-active rectifier 107 is configured to rectify the compensated high-frequency alternating current under the control of the Pulse Density Modulation control wave PDM to obtain a direct current voltage, and the output end of the semi-active rectifier 107 transmits the direct current voltage to a load to be charged.

Illustratively, the load to be charged may refer to a battery on an electric vehicle, for example.

The operation principle of the constant power wireless charging system will be described as follows.

It should be understood that, in the constant-power wireless charging system 100, the output power of the wireless charging transmitting device 101 to the wireless charging receiving device is Po = I _S ² *R _eq Wherein, I _S Output current I to the wireless charging receiver 102 for the primary side resonance compensation circuit 105 _S ，R _eq Is an equivalent ac load on the wireless charging receiving device 102 side. If the output power Po is required to be a constant value, the output current I is required to be maintained _S And R _eq Are all a constant value.

With continued reference to fig. 2, in the present embodiment, by introducing an additional compensation module 108 and a wireless power transmission module 109 in the primary side resonance compensation circuit 105, the reactive power caused by the primary side resonance compensation circuit 105 can be compensated for by the compensation module 108,and compensation by the wireless power transmission module 109 the power in the form of alternating current is transmitted wirelessly. In this way, the primary side resonance compensation circuit 105 outputs the output current I to the wireless charge receiving device 102 _S Independent of the mutual inductance M between the wireless power transmission module 109 and the resonance compensation circuit 106 on the secondary side, and outputs a current I _S The value of (d) can also be controlled by adjusting the parameter values of the components in the compensation module 108, that is, different output currents I can be configured by the parameter values of the components in the compensation module 108 _S The method is suitable for the scenes of developing products with different output specifications based on the same LCT/coil and platform.

Meanwhile, in the present embodiment, a pulse density modulation control wave is connected to the second input terminal of the semi-active rectifier 107, and the rectification process of the semi-active rectifier 107 is controlled by the pulse density modulation control wave, so as to control the period T of the pulse density modulation control wave _PDM Internally modulating an output ac voltage v output from the primary side resonance compensation circuit 105 to the wireless charging receiver 102 _S So as to adjust the equivalent ac load R on the wireless charging receiving device 102 side _eq The purpose of (1) is to always keep the equivalent alternating current load Req at a constant value.

Therefore, the constant-power wireless charging system provided by the application realizes wireless constant-power charging without an additional DC-DC converter, an auxiliary SCC circuit or a wireless feedback communication circuit, obviously improves the system efficiency and reduces the system complexity and cost.

In summary, the present application provides a constant-power wireless charging system, in this scheme, an additional compensation module and a wireless power transmission module are mainly introduced into a primary-side resonance compensation circuit, and a pulse density modulation control wave is connected to a second input end of a semi-active rectifier; thus, the output current I output by the resonance compensation circuit on the primary side to the wireless charging receiving device can be enabled _S Independent of mutual inductance M between the wireless power transmission module and the secondary side resonance compensation circuit, so as to output current I _S Can be given a value ofThe control is realized by adjusting the parameter values of all components in the compensation module; and in a pulse density modulated control wave period T _PDM Output alternating voltage v output from resonance compensation circuit internally modulating primary side to wireless charging receiving device _S So as to adjust the equivalent AC load R at the side of the wireless charging receiving device _eq The object of (a); and then the output power of the wireless charging transmitting device to the wireless charging receiving device is constant power, namely, the wireless constant power charging is realized under the condition that additional conversion layers such as a DC-DC converter, an auxiliary SCC circuit or a wireless feedback communication circuit are not needed, the system efficiency is obviously improved, and the system complexity and the cost are reduced.

The following embodiments will be used to specifically explain the circuit structure of each module in the constant-power wireless charging system in fig. 2.

Optionally, as shown with reference to fig. 3, the compensation module 108 includes: first inductance L _P1 A first capacitor C _P1 A second capacitor C _P2 。

First inductance L _P1 Is connected with the output end of the full-bridge voltage source inverter, and a first inductor L _P1 Respectively connected with the first capacitor C _P1 One terminal of (C), a second capacitor C _P2 Is connected with one end of the connecting rod;

a first capacitor C _P1 The other end of the voltage source is connected with the output end of the full-bridge voltage source inverter and the input end of the wireless power transmission module 109 respectively;

a second capacitor C _P2 And the other end thereof is connected to an input terminal of the wireless power transmission module 109.

Optionally, as shown with continued reference to fig. 3, the wireless power transmission module includes: transmitting coil L _P ；

Transmitting coil L _P And a second capacitor C _P2 Is connected to the other end of the transmitting coil L _P And the other end of the first capacitor C _P1 The other end of the first and second connecting rods is connected;

transmitting coil L _P Is electromagnetically connected with the resonance compensation circuit on the secondary side.

Optionally, with continued reference to FIG. 3The resonance compensation circuit 106 on the secondary side includes: third capacitor C _S And a receiving coil L _S ；

Receiving coil L _S And a transmitting coil L _P Electromagnetic connection;

optionally, a transmitting coil L _P And a receiving coil L _S Is 1:1, i.e. a loosely coupled transformer with LCC-S compensation topology consisting of a primary side resonance compensation circuit and a secondary side resonance compensation circuit consists of symmetrical coils (i.e. turns ratio =1, n = 1), wherein the transmitting coil L _P And a receiving coil L _S M, transmitting coil L _P And a receiving coil L _S May be determined by a coupling coefficient

And (4) showing.

It is noted that a transmitting coil L is shown in fig. 3 _P Resistance R of _P And receiving coil L _S Resistance R of _S Resistance R _P And a resistor R _S It is the coil loss of the loosely coupled transformer that can be expressed as two components.

With continued reference to FIG. 3, the third capacitor C _S One end of (2) and a receiving coil L _S One end of the third capacitor is connected with the first input end of the semi-active rectifier;

receiving coil L _S And the other end of the second diode is connected with the first input end of the semi-active rectifier.

Optionally, with continued reference to fig. 3, the semi-active rectifier comprises: the diode comprises a first diode D1, a second diode D2, a first switch tube S1 and a second switch tube S2; i.e. a semi-active rectifier, is composed of two diodes D1 and D2 and two MOSFET switches S1 and S2, wherein the two MOSFET switches S1 and S2 are accompanied by two oppositely arranged diodes D3 and D4.

One end of the first diode D1 is connected to one end of the second diode D2 and one end of the load to be charged, and the other end of the first diode D1 is connected to the third capacitor C _S Is connected with the first end of the first switch tube S1;

the second end of the first switch tube S1 is connected to the second end of the second switch tube S2 and the other end of the load to be charged;

a first end of the second switch tube S2 is connected to the other end of the second diode D2 and the receiving coil L respectively _S The other end of the first and second connecting rods is connected;

the third end of the first switch tube S1 and the third end of the second switch tube S2 are both connected with a control device so as to access the pulse density modulation control wave, and the pulse density modulation control wave controls the on-off states of the first switch tube S1 and the second switch tube S2.

Optionally, referring to fig. 4, in order to filter the dc power output from the half-active rectifier, the constant-power wireless charging system 100 further includes: filter capacitor C _f ；

Filter capacitor C _f One end of the first diode D1, one end of the second diode D2, and one end of the load to be charged are connected to the filter capacitor C _f The other end of the first switch tube S1, the second end of the second switch tube S2, and the load R to be charged _L The other end of the first and second connecting rods is connected; wherein R is _L The load to be charged may act as a battery.

Filter capacitor C _f The DC filter is used for filtering the DC output by the output end of the semi-active rectifier to obtain filtered DC voltage, and transmitting the filtered DC voltage to a load R to be charged _L 。

The circuit configuration of the full-bridge voltage source inverter in fig. 2 will be explained in the following embodiments.

Optionally, with continued reference to fig. 3 or fig. 4, the full-bridge voltage source inverter comprises: a third switching tube S3, a fourth switching tube S4, a fifth switching tube S5 and a sixth switching tube S6; i.e. the full bridge voltage source inverter is a full bridge VSI consisting of four MOSFET switches S3 to S6 to convert the dc voltage source into a high frequency ac form through these four MOSFET switches.

The first end of the third switch tube S3 and the first end of the fourth switch tube S4 are both used for connecting to the positive electrode Vin of the direct-current voltage sourceA second end of the third switch tube S3, a first end of the fifth switch tube S5 and the first inductor L _P1 Is connected with one end of the connecting rod;

the second end of the fourth switching tube S4 is respectively connected to the first end of the sixth switching tube S6 and the first capacitor C _P1 The other end of the first and second connecting rods is connected;

the second end of the fifth switching tube S5 and the second end of the sixth switching tube S6 are both used for connecting to the negative electrode Vin-of the dc voltage source.

Fig. 5 is a schematic diagram of a working waveform, a first switching sequence waveform, and a second switching sequence waveform of a full-bridge voltage source inverter, and referring to fig. 5, a third terminal of a third switching tube S3 and a third terminal of a sixth switching tube S6 are both used for accessing the first switching sequence waveform, a third terminal of a fourth switching tube S4 and a third terminal of a fifth switching tube S5 are both used for accessing the second switching sequence waveform, and the third switching tube S3, the fourth switching tube S4, the fifth switching tube S5, and the sixth switching tube S6 are used for performing high-frequency conversion on a direct-current voltage source under the control of the first switching sequence waveform and the second switching sequence waveform to obtain a high-frequency alternating-current voltage.

The operation principle of the constant-power wireless charging system provided by the present application will be described below with reference to the circuit structures of the constant-power wireless charging system shown in fig. 2 to 4.

First, the meaning of each parameter in the circuit configuration of the constant-power wireless charging system will be explained. During the wireless power transmission, v _P And i _P Respectively representing input AC voltage and input AC current, v _S And i _S Respectively representing output AC voltage and output AC current, V _o And I _o Respectively representing the dc charging voltage and the dc charging current of the battery.

Since battery charging is a slow process compared to the duty cycle of the LCC-SIPT converter, the battery can be modeled as a resistor R _L Wherein R is _L ＝V _o /I _o 。

Several advantages of the constant power wireless charging system provided by the present application will be described in detail below, wherein (1) the primary side of the wireless charging transmitter isOutput current I output by the output end of the resonance compensation circuit _S Independent of the load; (2) By varying the first inductance L in the primary side resonance compensation circuit _P1 A first capacitor C _P1 Without changing the transmitting coil L in the primary side resonance compensation circuit _P Can change the characteristics of the output capability; (3) The zero-voltage switching characteristic can be obtained, and the purpose of reducing the system power consumption is achieved.

Advantages (1) to (2) of the constant-power wireless charging system will be described first.

In order to clarify the parameter design process of each component in the constant-power wireless charging system, the constant-power wireless charging system is analyzed by using a T equivalent circuit model of the LCC-SIPT converter shown in fig. 6.

FIG. 6 is a schematic diagram of a T equivalent model of an IPT converter with LCC-S compensation topology by assuming the resonant circuit of FIG. 6 at a resonant angular frequency ω _R In operation, the equivalent circuit model using the Fundamental Harmonic Approximation (FHA) is accurate enough and can simplify the analysis process. Wherein the first capacitor C _P1 Is divided into two parallel capacitors C _P1' And C _P1” In addition, the semi-active rectifier and the load R to be charged _L Can be modeled as an equivalent resistance R _eq 。

Wherein, the variable V _P 、I _P 、V _S And I _S Are each v _P 、i _P 、v _S 、i _S Of the fundamental component of (a), and I _LC Is at C _P1' And C _P1” The phasor of the fundamental component of the current flowing in between. V _P 、V _S 、I _P And I _S Respectively has a root mean square value of

And

in this embodiment, the parasitic resistance and coil loss of the LCC-S compensation element are ignored for simplicity of the analysis process. In addition, the LCC-SIPT converter uses a symmetric coil, which makes component values reflected from the secondary side remain unchanged in the T model.

The loosely coupled transformer LCT is modeled in the circuit as three elements, the primary side leakage inductance L _lp Secondary side leakage inductance L _ls And mutual inductance L _M The relationship therebetween can be expressed as the following formula (1) to formula (2):

L _M ＝n·M＝M (1)

L _P ＝L _lp +L _M ＝L _S ＝L _ls +L _M (2)

wherein the application of high order compensation like LCC-S to an IPT converter can be modeled as a combination of several resonant circuits and the load independent transfer characteristics can be derived step by step. For the LC resonant circuit in FIG. 6, the inductance L _P1 And a capacitor C _P1' The following formula (3) should be satisfied:

the transconductance of the LC resonant circuit can be expressed as the following equation (4):

the LC resonant circuit then acts as a current source I for a pi-type resonant circuit _LC . Its corresponding current gain

Can be expressed as the following equation (5):

if the condition shown in the following formula (6) is satisfied:

due to the second capacitance C _P2 Designed as leakage inductance L _lp Thus, the current gain G _ii Is set as a unit number (G) _ii = 1), which means that the transconductance of the entire LCC-SIPT converter can be expressed as the following equation (7):

thus, neglecting coil losses and converter losses, the output current I is independent of the load _S Shown by the following formula (8):

furthermore, unlike the SS compensation topology, in which the output current I is different from the output current I, it can be obtained from equation (8) _S Independent of mutual inductance M, and I _S The value can be adjusted by adjusting the first capacitance L _P1 Is controlled by the value of (c). This shows that the constant-power wireless charging system can keep the design of the coil unchanged by changing the first inductor L _P1 And a first capacitor C _P1 To configure different output currents I _S . Therefore, the constant-power wireless charging system can be suitable for the scenes of developing products with different output specifications based on the same LCT/coil and platform.

Next, the advantage (3) of the constant-power wireless charging system will be described.

The switching sequence and the operating waveform of each switching tube in the full-bridge voltage source inverter 104 are shown in fig. 5. For realizing Zero voltage switching characteristics (ZVS), wherein v is _P And i _P The phase angle theta between the first and second switching tubes should satisfy a certain amount to make the third switching tube S ₃ And a first step ofSix switch tubes S ₆ Parasitic output capacitance C of _oss (or a fourth switching tube S ₄ &Fifth switch tube S ₅ ) Fully charged/discharged before the MOSFET turns on, the minimum value of which is shown in equation (9):

wherein f is _R Is the resonant frequency of the circuit, C _oss,u And C _oss,l Respectively represent S ₃ &S ₆ (or S) ₄ &S ₅ ) Wherein the switch tube S is assumed ₃ To S ₆ All adopt the same model (namely C) _oss,u ＝C _oss,l ) I.e. C _oss,u And C _oss,l Are the same herein (i.e., C) _oss,u ＝C _oss,l ＝C _oss )。

Combining equations (9) and V _P,rms And I _P,rms Expression of (a), theta _min It can be further deduced as shown in the following equation (10):

θ _min ＝arccos(1-π ² f _R |Z _in |C _oss ) (10)

this is related to the input impedance | Z _in And | is related. In addition, the phase angle θ is determined by the load condition (i.e., input impedance Z) _in ) And (4) determining. To calculate the input impedance Z _in First, leakage inductance L of primary side _ls Should be formed by the third capacitor C _S Compensation, and is shown in equation (11) below:

then, based on the above analysis of the T equivalent circuit of the constant-power wireless charging system, the input impedance can be calculated as shown in the following equation (12):

i.e. it can be obtained that the input impedance is inductive. Therefore, its input phase angle can be derived as shown in the following equation (13):

wherein, the formula (13) represents the input phase angle θ and the first inductance L _P1 And an equivalent AC load R _eq And (4) correlating. Referring to fig. 7, fig. 7 shows the first inductance L after visualization _P1 Equivalent AC load R _eq And the angular difference Δ θ = θ - θ _min In which Δ θ should be above the zero plane to achieve the ZVS condition, (assuming C is higher than the zero plane) _OSS =1000 pF) indicating that points above the zero plane are desirable system parameters such that θ is>θ _min And ensures a zero voltage switching characteristic ZVS at the full bridge voltage source inverter VSI. Furthermore, due to the equivalent AC load R _eq The regulation to constant can be done by controlling SAR at the secondary side to maintain the angular difference Δ θ, which makes it possible to guarantee that the full-bridge voltage source inverter VSI operates under ZPA conditions throughout the charging process.

How to use the semi-active rectifier SAR in the constant power wireless charging system for the equivalent ac load R will be described as follows _eq And control strategies for optimal equivalent ac load Req, opt and CP charging.

(A) Semi-active rectifier SAR for regulating equivalent alternating current load

Wherein, the semi-active rectifier SAR is controlled by the pwm control wave, fig. 8 is a waveform diagram of the pwm control wave, and as shown in fig. 8, two MOSFET switch tubes S in the semi-active rectifier SAR can be controlled by the pwm control wave ₁ And S ₂ Simultaneously turned on or off a plurality of times to modulate the control wave period T at a pulse density _PDM Inner modulation v _S The duty cycle of (c).

Compared with the traditional pulse modulation width, the method can always meet the zero-voltage switching characteristic ZVS condition without an additional control circuit, because i _S And v _S Are aligned in unison. Specifically, T _Pi Is a MOSFET switching tube S ₁ And S ₂ Period of operation in the passive state, T _Ai Is a MOSFET switching tube S ₁ And S ₂ Operating in the active state for a period of time, where i is within the pulse density modulated control wave period [1,q ]]Sequence numbers within the range. The ratio of the total passive state time to the time of one pulse density modulation control wave period is the control factor α, which is specifically expressed by the following equation (14):

since the control factor α affects the average output voltage of the semi-active rectifier SAR, it can be specifically expressed as the following formula (15):

V _o,avg ＝V _o ·α (15)

then, the average output power in one period of the pulse density modulation control wave can be specifically expressed as shown in the following formula (16):

combining equation (16) with the expression of IS, rms, the equivalent ac load Req can be derived as shown in equation (17) below:

that is, as is clear from the formula (17), the equivalent ac load R can be adjusted by controlling the control factor α of the semi-active rectifier SAR _eq 。

(B) CP charging control strategy with optimal transmission efficiency

(1) Conditions for optimum transmission efficiency

Fig. 9 is an M-equivalent model of IPT with LCC-S compensation topology, where fig. 9 (a) is the model before secondary reflection and fig. 9 (b) is the model after reflection. For the transmission efficiency analysis, a simplified analysis is mainly performed by using an M equivalent circuit model of the LCC-S IPT converter as shown in fig. 9, where coil loss is considered, fig. 9 (a) is a model before secondary side reflection, and fig. 9 (b) is a model after reflection. Applying kirchhoff's voltage law to the M model, the following relationships (18) to (20) can be obtained:

in this embodiment, the total transmission efficiency can be regarded as the product of the transmission efficiencies of the respective stages. Therefore, starting from the secondary side, the transmission efficiency can be expressed as shown in the following equation (21):

to calculate the transmission efficiency of the primary side, first, the slave voltage source-j ω MIS can be replaced with an equivalent impedance ZRcv, which is the equivalent impedance reflected from the secondary side to the primary side, as shown in equation (22):

in conjunction with equations (20) and (22), Z Rcv can be further derived as shown in equation (23) below:

then, the transmission efficiency of the primary side can be obtained as shown in the following equation (24):

therefore, the overall transmission efficiency is as shown in the following equation (25):

by analyzing the formula (25), it can be known that the transmission efficiency is affected by the equivalent ac load Req. Therefore, the total transmission efficiency can be maximized under the optimal equivalent ac load Req, opt which can be calculated

Thus obtaining the product. The optimal equivalent alternating current load is obtained by calculation as shown in the following formula (26):

(2) Rated output power design and constant power output control strategy

As described in the above embodiments, the proposed LCC-SIPT based wireless charger has an output current independent of the load, and the semi-active rectifier SAR can regulate the equivalent alternating load Req. In order to ensure that the constant-power wireless charging system provides constant power output while providing optimal transmission efficiency, the constant-power wireless charging system should be designed to output constant power under optimal equivalent alternating current loads Req, opt in the whole charging process.

Since the optimal ac equivalent load Req, opt is determined by the inherent characteristics of the LCT, the ac equivalent load Req can be maintained at its optimal value by the modulation of the semi-active rectifier SAR, the rated output power can be determined by designing the output current independent of the load, and the present embodiment designs the average output power to be close to the rated output power, which is expressed by the following equation (27):

FIG. 10 shows different cell internal resistances R at various control factors α (α =1, α =0.9375, α = 0.8750), po/Po, rated and η/η Optimal _L As shown in FIG. 10, when the ratio is

Equal to 1 and always kept at 1 (Po = Po, rated = constant), the proposed charger can achieve both CP output and optimal transmission efficiency by applying only the PDM-based SAR control. In addition, since the adjusted value of the equivalent alternating current load Req is only greater than the value before adjustment, in order to ensure that the system operates with the optimal transmission efficiency during the entire charging process, the minimum equivalent alternating current load of the charger should be set as the optimal equivalent alternating current load. The equivalent ac load Req range can be expressed as shown in the following equation (28):

FIG. 11 shows the internal resistances R of different batteries _L Corresponding control factor alpha and ratio of Req/Req, opt with and without SAR control, fig. 11 shows the internal resistance R of different cells to maintain the optimum equivalent ac load Req, opt _L The lower corresponding monotone decreasing curve of the control factor alpha. The dashed line and the monotonically increasing curve show the values of Req/Req, opt with and without SAR control, respectively. The curve visually displays the operating points of the equivalent alternating load regulation control strategy in different states.

According to the above analysis, a control scheme of an IPT converter for realizing wireless CP charging and optimal transfer efficiency based on LCC-S is proposed, and its control diagram is shown in fig. 12.

Optionally, the control device in fig. 1 includes: a sensor circuit, a microcontroller unit and a zero crossing detector.

the output end of the microcontroller unit is connected with the second input end of the semi-active rectifier, and the microcontroller unit is used for generating a pulse density modulation control wave according to the output current, the output voltage and a clock signal, and outputting the pulse density modulation control wave to a first switch tube and a second switch tube in the semi-active rectifier, so that the first switch tube and the second switch tube rectify the compensated high-frequency alternating voltage under the control of the pulse density modulation control wave to obtain the direct-current voltage.

the output end of the sensor circuit is connected with the input end of the multiplier, the output end of the multiplier is connected with the input end of the divider, and the multiplier is used for receiving the output current and the output voltage collected by the sensor circuit, calculating to obtain the current output power according to the output current and the output voltage, and transmitting the current output power to the divider;

the output end of the divider is connected with the input end of the proportional-integral controller, and the divider is used for calculating to obtain a first coefficient according to the current output power and the preset reference rated output power and transmitting the first coefficient to the proportional-integral controller;

and the second input end of the modulator is connected with the output end of the zero-crossing detector, and the modulator is used for receiving the clock signal output by the output end of the zero-crossing detector and generating a pulse density modulation control wave according to the clock signal and the control factor.

In the present embodiment, first, the sensor circuit senses the charging information, i.e., the output voltage Vo and the output current Io are collected by the sensor circuit and input to a microcontroller unit (MCU). Then, a multiplier is used to calculate the detected current output power Po, and Po is divided by its reference rated output power Po, and the rated result is a first coefficient β, which is expressed by the following equation (29):

a Proportional-Integral (PI) controller is used to approximate the required control factor alpha by finally correcting the first coefficient beta to approximately 1. Finally, the MCU generates a PDM sequence to control the switching tube S by inputting the control factor α into a sigma-delta (Σ Δ) modulator ₁ And S ₂ 。

In addition, the PDM period T _PDM It should be set in advance to determine the accuracy of the control factor alpha. Using secondary side current i _S The zero crossing detector as an input may generate a clock signal for synchronizing the power signal (i.e., v;) _S And i _S ) And a control signal. With the control scheme in fig. 12, the proposed wireless constant power charger only needs to perform data acquisition and modulation on the secondary side to achieve direct and safe output regulation, which means that communication between the primary side and the secondary side can be eliminated.

In addition, in order to verify the validity and feasibility of the proposed constant-power wireless charging system, a test is performed on the constant-power wireless charging system, wherein table 1 below shows parameters of each component in the constant-power wireless charging system. The dc power supply used in this test was KIKUSUI PWX1500H.

TABLE 1 parameters of components in constant-power wireless charging system

In terms of the range of rated output power and battery charging voltage, the equivalent dc load range for simulating battery charging in the voltage range of 74 to 109V is about 42 to 91 Ω. Therefore, an electronic load (ITECHIT 6833A) was applied to simulate battery packs having different states of charge. A YOKOGAWADL850E oscilloscope was used to capture the waveforms of the experimental voltage and current, and a YOKOGAWAPX8000 precision power oscilloscope was used to measure the dc input power and the dc output power.

In this test, the constant-power wireless charging system is tested under the alignment condition and the dislocation condition, respectively.

(1) Experimental results under aligned conditions

The test was first performed under aligned conditions. Fig. 13 is a waveform diagram of the constant power charger starting charging, fig. 14 is a waveform diagram of the constant power charger during charging, and fig. 15 is a waveform diagram of the constant power charger ending charging.

Wherein the AC input voltage v at the beginning, middle and end of the constant-power charging process _P And current i _P And an AC output voltage v _S And current i _S Fig. 13-15, where the corresponding dc output power curve is located at the bottom of each waveform diagram, indicating that the system can output constant power as desired.

Operating waveform v as amplified in fig. 13 _P And i _P Shown, primary side voltage v _P And current i _P The phase angle therebetween is around 20 °, satisfying a minimum required phase angle of about 7 ° to achieve the zero voltage switching characteristic ZVS of the switches in the full-bridge voltage source inverter VSI. Then, from the amplified operating waveforms vS and iS in fig. 14, it iS concluded that vS and iS are in phase, and no large voltage spike appears on the voltage vS curve, which shows the ZVS characteristic inherent in SAR. Furthermore, it is worth mentioning that the current ripple problem becomes significant as the control factor α decreases, as can be seen from FIGS. 14 and 15 hereinThe dc output power curve of (a) is reflected and verified. The direct current charging current with superimposed alternating current frequency does not accelerate the degradation of the lithium ion battery. Thus, current ripple is a tolerable problem for lithium ion battery charging in terms of battery life.

FIG. 16 is a schematic diagram of the DC-to-DC transmission efficiency η and the output power Po during the charging process, as shown in FIG. 16, the measured output power points correspond to different internal resistances R of the battery _L The curve marked with an asterisk (i.e. "). Obviously, the output power remains approximately 132.5W throughout the constant power charging process. Furthermore, the entire charging process can maintain the DC-to-DC transfer efficiency in the range of 87% to 90.25% while achieving the constant output power Po, which verifies that the proposed constant power charger and the control scheme thereof can achieve the constant output power and maximize the transfer efficiency at the same time. Furthermore, the lower part of the graph shows the DC-to-DC transfer efficiency η measured under conditions of α =1 with different R _L The relationship between (i.e., ") and (ii) the optimum point is marked with a diamond, which means that the constant power charging system begins charging at the optimum efficiency point, the analysis in the above section of the embodiments is consistent. Furthermore, it is worth noting that as the control factor α decreases, the transfer efficiency η decreases slightly, which may be due to the increase of the total time the MOSFET switch transistors S1 and S2 in the semi-active rectifier are operated in the active state, resulting in the increase of the power consumption consumed thereby.

FIG. 17 shows the load R with different output voltage Vo and output current Io during charging _L Fig. 17 shows that the dc output current Io decreases with increasing dc output voltage Vo during charging, which is consistent with a constant power charging curve. Fig. 18 is a schematic diagram of the operating electrical points in the aligned and misaligned condition, and the control factor α for maintaining constant power output for different internal cell resistances Req is shown in fig. 18, which is represented as a monotonically decreasing curve marked by a circle (i.e., "o"). As the charging process advances, the control factor a of the SAR decreases. By adding R _L Increasing from 42 Ω to 92 Ω, the control factor α changes from 1 to 0.75. Compared with the simulation results shown in FIG. 11, although the experimental results have a certain deviationTheir trends are nearly identical. Furthermore, the semi-active rectifier SAR is adjusted according to the approximation of the actual output power Po to the reference output power Po, rated, which indicates that the control strategy can inherently cope with such slight deviations.

(2) Experimental results in case of misalignment

Because wireless charging positioning error, the dislocation can lead to the leakage inductance to increase. A purposefully designed LCT can mitigate and minimize this effect on the variation of the coupling coefficient k. Furthermore, since the CP output can be achieved by equivalent ac load regulation, the proposed charging system can still provide a constant output power within a reasonable misalignment range.

Fig. 19 is a schematic diagram of the DC-to-DC transfer efficiency η and the output power Po measured in the case of alignment and misalignment during charging, and in order to verify this, tests were performed under the misalignment condition (k = 0.571) and the alignment condition (k = 0.615), with specific reference to fig. 19. In addition, the corresponding changes in their control factors are shown in figure 18, with the asterisked (i.e. ") curves. Throughout the charging process, the output power Po in the misaligned condition can remain almost the same, about 133.1W, as in the aligned condition, while the charging system still achieves its optimal transfer efficiency, verifying the effectiveness and feasibility of the proposed constant power charger even in the misaligned condition. The slight decrease in transmission efficiency η may be mainly due to the reduction in k caused by misalignment and a smaller control factor α is required to obtain the optimal equivalent ac load Req, opt compared to the aligned case.

In summary, the present application provides a single-stage IPT converter based on an LCC-S, which is used for realizing wireless constant power charging and optimal transmission efficiency in the whole charging process. In order to achieve two aims of constant power output and efficiency optimization at the same time, a semi-active rectifier SAR adjustment which is based on pulse density modulation control waves and does not need communication is adopted, and a single control factor is adjusted only on a secondary side. Compared with the prior art, the IPT converter adopts LCC-S compensation to enable the output characteristic independent of the load to be independent of mutual inductance, and provides freedom for system parameter design. The proposed wireless constant power charger does not require an additional DC-DC converter or an auxiliary SCC circuit, nor a wireless feedback communication circuit, saving costs and reducing control complexity. Furthermore, this solution achieves a zero-voltage switching characteristic ZVS simultaneously in the full-bridge inverter and the semi-active rectifier SAR.

It will be understood by those skilled in the art that the foregoing is only a preferred embodiment of the present invention, and is not intended to limit the invention, and that any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included within the scope of the present invention.

Claims

1. A constant-power wireless charging system, comprising: the wireless charging system comprises a wireless charging transmitting device, a wireless charging receiving device and a control device;

the primary side resonance compensation circuit includes: the wireless power transmission device comprises a compensation module and a wireless power transmission module;

2. The system of claim 1, wherein the compensation module comprises: the first inductor, the first capacitor and the second capacitor;

the other end of the first capacitor is connected with the output end of the full-bridge voltage source inverter and the input end of the wireless power transmission module respectively;

the other end of the second capacitor is connected with the input end of the wireless power transmission module.

3. The system of claim 2, wherein the wireless power transfer module comprises: a transmitting coil;

4. The system of claim 3, wherein the secondary-side resonance compensation circuit comprises: a third capacitor, a receiving coil;

the receiving coil is electromagnetically connected with the transmitting coil;

5. The system of claim 4, wherein the semi-active rectifier comprises: the first diode, the second diode, the first switch tube and the second switch tube;

6. The system of claim 5, further comprising: a filter capacitor;

7. The system of claim 6, wherein the full bridge voltage source inverter comprises: a third switching tube, a fourth switching tube, a fifth switching tube and a sixth switching tube;

8. The system of claim 7, wherein the primary side resonant compensation circuit and the secondary side resonant compensation circuit comprise a loosely coupled transformer having an LC-series LCC-S compensation topology.

9. The system of claim 8, wherein the resonant frequency in the T-equivalent model of the loosely coupled transformer with LC-series LCC-S compensation topology is full when the primary side resonant compensation circuit and the secondary side resonant compensation circuit form a full resonant frequencyFoot

Wherein, ω is _R Is the resonant frequency, L _M Is mutual inductance, L _lp Is primary side leakage inductance, C _P1” Is a first capacitor C _P1 One of two capacitors in parallel, C _P2 Is a second capacitor, I _S To output an alternating current i _S Phasor of the fundamental component of (g) _m,IPT Is transconductance, V _P For input of an alternating voltage v _P Phasor of the fundamental component of (a).

10. The system of claim 8, wherein the full-bridge voltage source inverter operates at a zero voltage switching characteristic when the phase angle between the input ac voltage and the input ac current in the loosely coupled transformer is greater than or equal to a minimum phase angle, wherein the minimum phase angle is formulated for operation at zero voltage switching characteristic

Calculating to obtain;

11. The system of claim 8, wherein the semi-active rectifier is further configured to regulate an equivalent ac load by a control factor of the semi-active rectifier.

12. The system according to claim 1, wherein the pulse density modulated control wave is generated by the control means according to a ratio between an actual output power and a rated power of the load to be charged.

13. The system of claim 1, wherein the control device comprises: a sensor circuit, a microcontroller unit and a zero-crossing detector;

the output end of the microcontroller unit is connected with the second input end of the semi-active rectifier, and the microcontroller unit is configured to generate the pulse density modulation control wave according to the output current, the output voltage, and the clock signal, and output the pulse density modulation control wave to a first switching tube and a second switching tube in the semi-active rectifier, so that the first switching tube and the second switching tube rectify the compensated high-frequency alternating-current voltage under the control of the pulse density modulation control wave to obtain a direct-current voltage.

14. The system of claim 13, wherein the microcontroller unit comprises: the device comprises a multiplier, a divider, a proportional-integral controller and a modulator;

and the second input end of the modulator is connected with the output end of the zero-crossing detector, and the modulator is used for receiving the clock signal output by the output end of the zero-crossing detector and generating the pulse density modulation control wave according to the clock signal and the control factor.