CN115441171A - Two-radiation-zero coplanar waveguide dual-polarization trapped wave crossed dipole antenna - Google Patents

Two-radiation-zero coplanar waveguide dual-polarization trapped wave crossed dipole antenna Download PDF

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CN115441171A
CN115441171A CN202211016170.XA CN202211016170A CN115441171A CN 115441171 A CN115441171 A CN 115441171A CN 202211016170 A CN202211016170 A CN 202211016170A CN 115441171 A CN115441171 A CN 115441171A
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coplanar waveguide
antenna
radiation
dipole antenna
notch
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李敏
许崇彩
杨敏
李海霞
尹多多
刘昊
沃陈陈
高景行
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Suqian College
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Suqian College
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/36—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith
    • H01Q1/38—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith formed by a conductive layer on an insulating support
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/50—Structural association of antennas with earthing switches, lead-in devices or lightning protectors
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q15/00—Devices for reflection, refraction, diffraction or polarisation of waves radiated from an antenna, e.g. quasi-optical devices
    • H01Q15/0006—Devices acting selectively as reflecting surface, as diffracting or as refracting device, e.g. frequency filtering or angular spatial filtering devices
    • H01Q15/0053—Selective devices used as spatial filter or angular sidelobe filter
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q19/00—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic
    • H01Q19/10—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic using reflecting surfaces
    • H01Q19/104—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic using reflecting surfaces using a substantially flat reflector for deflecting the radiated beam, e.g. periscopic antennas
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/20—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements characterised by the operating wavebands
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q7/00—Loop antennas with a substantially uniform current distribution around the loop and having a directional radiation pattern in a plane perpendicular to the plane of the loop
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/16—Resonant antennas with feed intermediate between the extremities of the antenna, e.g. centre-fed dipole

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Abstract

The invention relates to a two-radiation-zero coplanar waveguide dual-polarization trapped wave crossed dipole antenna which comprises a dielectric substrate, a metal reflecting plate and two coaxial cables. The invention firstly designs a broadband dual-polarized crossed dipole antenna covering 1.7-3.6GHz by utilizing a coplanar waveguide feed structure, then designs a coplanar waveguide DMS-DGS mixed type second-order band-stop filter, and prints the miniaturized coplanar waveguide filter with high selectivity and the antenna on the same dielectric substrate, thereby realizing a 2.9-3.1GHz stop band. Under the condition that VSWR is less than 1.5, compared with an original broadband antenna covering a frequency band of 1.7-3.6GHz, the proposed notch antenna works in two working frequency bands of 1.7-2.7GHz and 3.4-3.6GHz, and the notch frequency band of 2.9-3.1GHz is introduced at the same time, so that the notch antenna can be applied to a 2G/3G/4G/5G base station requiring a notch function.

Description

Two-radiation-zero coplanar waveguide dual-polarization trapped wave crossed dipole antenna
Technical Field
The invention relates to the technical field of wireless communication, in particular to a coplanar waveguide dual-polarized trapped wave crossed dipole antenna applicable to a 2G/3G/4G/5G base station.
Background
The broadband dual-polarized antenna with the working frequency band of 1.71-2.69GHz is widely applied to wireless base station communication for the application of a 2G/3G/4G communication system. With the maturity of the 5G technology, the 3.4-3.6GHz band is applied to the large-scale MIMO technology, which not only ensures signal coverage, but also ensures channel capacity, and becomes the first band for operators to deploy 5G networks. For operators, a 5G network of 3.4-3.6GHz is built by multiplexing 2G/3G/4G sites, and the method is more cost-effective than large-scale construction of micro sites and has wider user coverage. Therefore, for 2G/3G/4G/5G base station application, it is necessary to research and design a broadband dual-polarized antenna covering 2G/3G/4G/5G frequency band (1.7-3.6 GHz).
According to the international telecommunication union, the 2.9-3.1GHz frequency band between 2G/3G/4G (1.71-2.69 GHz) and 5G (3.4-3.6 GHz) is used for radio navigation and positioning. Therefore, for the 2G/3G/4G/5G base station, the research and design of the dual-polarized antenna with the notch frequency band of 2.9-3.1GHz is commercially valuable. In order to suppress the interfering signals from these narrow bands, it is an effective way to introduce notch characteristics in the wideband antenna without affecting other technical parameters.
The patch antenna, the magnetoelectric dipole antenna, the crossed dipole antenna and other broadband dual-polarized antennas can be applied to the base station. The cross dipole antenna has the advantages of wide frequency band, dual polarization, stable radiation direction, small volume, convenient manufacture and the like, and is widely applied to a wireless communication system. Document [1] (yellow sea, liu yi, goer s. A broadband dual-polarized base station antenna with anti-interference capability [ J ]. IEEE antenna and wireless transmission prompter, 2016 1-1) realizes a stop band of 2.27-2.53GHz by arranging a C-shaped stub beside a feeder line. Document [2] (chenyili, chukuxin, a new base station filter antenna [ C ]. IEEE-APS wireless communication antenna and propagation topic conference (APWC). IEEE,2019, 063-065) -document [3] (broadband dual-polarized base station antenna with a second-order notch characteristic [ C ]. IEEE MTT-S International Wireless Seminar (IWS): IEEE, 2019) implements a second-order notch characteristic by placing a U-shaped stub near the feed line and etching a notched resonant ring on the main radiator. The difference between a document [4] (Chen Y L, chu Q x. A compact dual-band slot dual-polarized antenna for a base station [ C ]. International conference on microwave and millimeter wave technology (icmtt). 2019) and a document [3] is that the document [4] additionally adds two open resonant rings on a main radiator, so that a first-order notch characteristic and a second-order notch characteristic can be realized. Document [5] (fuss, cao Z, full X, et al. A broadband dual-polarized notch antenna for 2/3/4/5G base stations [ J ]. IEEE antenna and radio transmission bulletin, 2020, 19 (1): 69-73) proposes to introduce a cross-dumbbell-shaped parasitic element on the radiator to produce a notch of 2.9-3.1GHz. However, the documents [1] to [5] all implement the notch characteristic by adding an additional feeding structure or a parasitic element below the crossed dipole antenna, which increases the installation difficulty of the antenna and the manufacturing cost of the antenna.
In document [6] (waiver, litde, etc.. Extra circuit-free dual polarized notch antenna for 2.4/5GHz WLAN application [ J ]. IEEE Access,2019, PP (99): 1-1), notch function is realized by introducing C-shaped open resonant ring in broadband dipole antenna without adding extra filter circuit. The minimum gain of the notch band is suppressed from 8dBi to-6 dBi compared to the broadband dipole antenna as reference. Although this antenna is easy to install and can achieve good rejection, the notch band for VSWR >2 is too wide to be suitable for 2G/3G/4G/5G base station applications.
Disclosure of Invention
Aiming at the difficulties of poor trap effect, difficult processing and the like in the background technology, the invention researches a two-radiation-zero coplanar waveguide dual-polarization trap crossed dipole antenna which is suitable for 2G/3G/4G/5G base station application. Under the condition that VSWR is less than 1.5, the proposed notch antenna works in two working frequency bands of 1.7-2.7GHz and 3.4-3.6GHz, and the notch frequency band of 2.9-3.1GHz is introduced at the same time, so that the proposed notch antenna has two radiation zeros, and the difficulty that the notch effect is poor is solved. Meanwhile, only one dielectric substrate is needed for antenna processing, and the antenna is simple to process.
In order to achieve the purpose, the invention adopts the following technical scheme:
a two-radiation-zero coplanar waveguide dual-polarization trapped wave crossed dipole antenna comprises a dielectric substrate, a metal reflecting plate and two coaxial cables, wherein the crossed dipole antenna is printed on the upper layer and the lower layer of the substrate and is fed by the two coaxial cables, and the metal reflecting plate is positioned below the dielectric substrate; the crossed dipole antenna comprises an upper metal layer and a lower metal layer, each layer is provided with two pairs of annular dipole antennas, the inside of each annular dipole arm is provided with a rectangular metal layer, and the edge of each antenna arm between crossed dipoles is designed into an exponential function shape; two sections of coplanar waveguide transmission lines are arranged in the upper metal layer, a hexagonal capacitive open-circuit branch is arranged along each section of coplanar waveguide transmission line, and an M-shaped DMS coplanar waveguide filter is embedded in each coplanar waveguide transmission line; in the lower metal layer, a double-turn hexagonal DGS resonator, a hexagonal capacitive open-circuit branch and a DGS resonator are arranged right below each hexagonal capacitive open-circuit branch corresponding to the upper metal layer to form a DGS band elimination filter unit.
Preferably, the dielectric substrate adopts Rogers-4350, and the dielectric constant of the dielectric substrate is epsilon r ＝3.66，The length of the substrate is Ls =55mm, the thickness is Hs =1mm, and the thickness of the metal layer of the substrate is Hm = 0.035mm.
Preferably, the metal reflecting plate is square and is placed 35mm below the dielectric substrate, so as to realize a directional radiation mode of about 65 degrees.
Preferably, the upper and lower metal layers are connected to each other by 21 short-circuit metalized via holes, wherein 20 short-circuit metalized via holes are used for connecting the upper and lower layer dipole antennas, and 1 short-circuit metalized via hole is used for connecting the coplanar waveguide transmission line.
More preferably, to avoid the overlap, one of the coplanar waveguide transmission lines is modified so that a portion of the line is printed on the bottom layer of the substrate, and then the top and bottom of the coplanar waveguide transmission line are connected by a short-circuited metalized via.
Further, the expression of the exponential function is Y (x) = Ce kx + B, where k is a constant coefficient of the exponential function and B is a parameter of the exponential function.
Further, the outer conductor of the coaxial cable is connected to one arm of the dipole at the bottom of the substrate, the inner conductor of the coaxial cable is connected to one end of the coplanar waveguide transmission line through the substrate, and the other end of the coplanar waveguide transmission line is connected to the other arm of the dipole.
Preferably, the DMS is 12.34mm in length.
Compared with the prior art, the invention has the following beneficial effects:
the invention firstly designs a broadband dual-polarized crossed dipole antenna covering 1.7-3.6GHz by utilizing a coplanar waveguide feed structure, then designs a coplanar waveguide DMS-DGS mixed type second-order band-stop filter, and prints the miniaturized highly-selective coplanar waveguide filter and the antenna on the same dielectric substrate, thereby realizing a 2.9-3.1GHz stop band. Under the condition that VSWR is less than 1.5, compared with the original broadband antenna covering the frequency band of 1.7-3.6GHz, the notch antenna provided works in two working frequency bands of 1.7-2.7GHz and 3.4-3.6GHz, and simultaneously introduces the notch frequency band of 2.9-3.1GHz. The proposed notch antenna has an average gain of about 8.3dBi at low band and about 7.1dBi at high band. Within the stop band, the antenna gains are all lower than 0dBi, and the minimum gain is-14.5 dBi. Therefore, when VSWR is less than 1.5, the dual-polarized notch antenna provided by the invention has working frequency bands of 1.7-2.7GHz and 3.4-3.6GHz and a notch frequency band of 2.9-3.1GHz, and can be applied to 2G/3G/4G/5G base stations needing notch functions.
Drawings
Fig. 1-1 is an overall structural view of a notch antenna of the present invention.
Fig. 1-2 are overall test charts of notch antennas of the present invention.
Fig. 2-1 is a top view of an upper metal layer of the present invention.
Fig. 2-2 is a top view of a lower metal layer of the present invention.
Fig. 3 is two broadband antenna diagrams, wherein (a) the reference broadband antenna and (b) the coplanar waveguide fed broadband antenna proposed by the present invention.
Fig. 4 is a graph of simulation results for two antennas.
FIG. 5 is a graph of single-turn DGS and two-turn DGS resonators and their resonant frequencies versus W3.
Fig. 6 is a coplanar waveguide DGS filter cell.
Fig. 7 shows the electromagnetic and circuit simulation results of the coplanar waveguide filter unit.
Fig. 8 shows three DMS resonators.
FIG. 9 shows the resonant frequencies and F of three DMS resonators 2 The relationship (2) of (c).
Figure 10 shows the electromagnetic and circuit simulation results of a coplanar waveguide DMS filter.
Fig. 11 is a coplanar waveguide hybrid second-order band-stop filter according to the present invention.
Fig. 12 is an equivalent circuit of the coplanar waveguide hybrid second-order band-stop filter according to the present invention.
Fig. 13 shows the electromagnetic and circuit simulation results of the coplanar waveguide DMS filter according to the present invention.
FIG. 14 is a simulation of | S11| for a DGS notch, DMS notch and hybrid filter.
Fig. 15 is a coplanar waveguide broadband antenna and notch antenna.
Fig. 16 is a simulation result of both the coplanar waveguide broadband antenna and the notch antenna.
Detailed Description
To further illustrate the technical means and effects of the present invention adopted to achieve the predetermined objects, the following detailed description of the embodiments, structures, features and effects according to the present invention will be made with reference to the accompanying drawings and preferred embodiments.
As shown in fig. 1-1 and 1-2, the overall structure of the present invention comprises a dielectric substrate, a metal reflector and two coaxial cables. The crossed dipole antenna is printed on the upper layer and the lower layer of the substrate and is fed by two coaxial cables. The dielectric substrate adopts Rogers-4350 with a dielectric constant of epsilon r =3.66, length of substrate L s =55mm, thickness H s =1mm, the thickness of the metal layer of the substrate is H m = 0.035mm. Due to the thickness H of the metal layer of the substrate m The performance of the proposed coplanar waveguide filter is affected and therefore H needs to be considered when simulating and designing the filter m The value of (c). A square metal reflecting plate is placed 35mm below the dielectric substrate to realize a directional radiation mode of about 65 degrees.
As shown in fig. 2-1 and 2-2, the parameter values for the top view of the upper metal layer and the top view of the lower metal layer are given, respectively. The upper and lower metal layers are connected with each other through 21 short circuit metallized through holes, wherein 20 short circuit metallized through holes are used for connecting the upper and lower layer dipole antennas, and 1 short circuit metallized through hole is used for connecting the coplanar waveguide transmission line. Each layer is provided with two pairs of annular dipole antennas, and the inside of each annular dipole arm is a rectangular metal layer. Meanwhile, the designed antenna adopts an exponential antenna arm to realize broadband characteristics. The edges of the antenna arms between the crossed dipoles are designed to be exponential in shape. The expression of the exponential function is Y (x) = Ce kx + B, where k is a constant coefficient of the exponential function. The outer conductor of the coaxial cable is connected to one arm of the dipole at the bottom of the substrate and the inner conductor of the coaxial cable is connected to one end of the coplanar waveguide transmission line through the substrate. The other end of the coplanar waveguide transmission line is connected to the other arm of the dipole. To is coming toAvoiding overlapping, modifying one of the coplanar waveguide transmission lines to print a part of the line on the bottom layer of the substrate, and then connecting the upper part and the bottom part of the coplanar waveguide transmission line through a short circuit metallized via hole.
As shown in fig. 2-1, an M-shaped DMS coplanar waveguide filter is embedded in the coplanar waveguide microstrip line in the upper metal layer. Length F of DMS 2 =12.34mm, the groove width and the metal line width are both set to W n . Along each segment of the coplanar waveguide transmission line, there is a hexagonal capacitive open stub. As shown in fig. 2-2, there is a double turn hexagonal DGS resonator in the lower metal layer directly below each hexagonal capacitive open stub. A DGS band stop filter unit can be formed by a hexagonal capacitive open-circuit branch and a defected ground structure DGS resonator.
Experimental data:
1. coplanar waveguide broadband antenna
The invention uses a broadband cross dipole antenna based on an improved direct feed structure as a reference antenna, which is proposed by a document [8] (Zusan, lide, and the like. An ultra-wideband dual-polarized antenna [ J ] with three resonance modes for a 2G/3G/4G/5G communication system, IEEE Access, 2019. In fig. 3 (b), the feed structure of the antenna proposed by the present invention is changed from a ring structure to a coplanar waveguide structure with all-metal printed inner arm. Fig. 4 shows simulation results of two antennas, and the proposed broadband antenna using the coplanar waveguide feed structure can also achieve wide bandwidth (1.7-3.6 GHz) and high isolation (> 28 dB). Although the two antennas have the same characteristics, the antenna proposed by the present invention can integrate a filter into a coplanar waveguide feed transmission line. Therefore, the coplanar waveguide broadband antenna provided by the invention is the basis for integrating a filter inside the antenna to realize the filtering characteristic.
2. Coplanar waveguide second-order hybrid filter
In order to integrate the band-stop filter into the proposed coplanar waveguide fed broadband antenna, the band-stop filter needs to be designed in a miniaturized manner. In order to realize the notch characteristic in the 2.9-3.1GHz band, the band-stop filter needs to have a high rectangular coefficient. Therefore, the invention designs a miniaturized coplanar waveguide hybrid filter with two transmission zeros.
(1) DGS band-stop filter with coplanar waveguide
The experiment adopts a DGS band elimination filter, the working principle of a DGS band elimination filter unit is that the DGS band elimination filter unit consists of a double-turn Defected Ground Structure DGS (Defected Ground Structure) resonator and an open-circuit branch used as a compensation capacitor. FIG. 5 shows the resonant frequency of a single-turn DGS and a double-turn DGS resonator versus W 3 It can be seen that, the double-turn DGS resonator can adopt a smaller size to realize miniaturization while realizing the same resonance frequency. W 3 The size of (A) determines the resonant frequency of the DGS resonator, and the appropriate W is selected 3 Can be dimensioned to achieve the desired notch frequency.
In order to improve the rejection characteristics of the notch filter, as shown in fig. 6, a side length W is added to the transmission line above the defective ground structure 2 =1.62mm hexagonal coplanar waveguide capacitive open stub. The proposed coplanar waveguide filter unit was simulated using HFSS software, the resonance frequency f 0 = 2.96GHz cut-off frequency f c =2.889GHz, | S11| = -0.6, dB = -0.93, characteristic impedance Z 0 Set to 50 omega. C =11.07nf, L =0.2611nh, and R =1392 Ω can be calculated according to the formula (1-3), where C: capacitance, L: inductance, and R: represents resistance.
Figure RE-GDA0003923673530000081
Figure RE-GDA0003923673530000082
Figure RE-GDA0003923673530000083
The results of electromagnetic simulation and circuit simulation are shown in fig. 7, and the DGS filter unit has good suppression performance, but the rectangular coefficient is not high enough, and the selectivity is poor.
(2) Coplanar waveguide DMS band stop filter
The experiment used a DMS band reject filter, with the DMS resonators etched in the coplanar waveguide transmission line to achieve a 3.4-3.6GHz stop band. The DMS resonator has an overall length close to half the wavelength of the resonance frequency. Fig. 8 (a), (b) and (c) show U-, N-and M-slotted DMS resonators, respectively, which are bent once, twice and three times, respectively. Resonant frequencies of the three resonators are dependent on F 2 The variation of (c) is shown in fig. 9. It can be seen that the M-type DMS resonator can achieve a lower resonant frequency at the same size. When resonance is required to be performed near the frequency point of 3.1GHz, only the M-shaped DMS resonator can be embedded into the broadband antenna provided by the invention. Therefore, experimental comparison is made with a miniaturized design using an M-shaped DMS resonator as a notch filter.
Document [9 ]](Sam W Y, zakaaria Z, mutalib M A et al. Compact DMS three band stop filter [ C ] for use in U-slot of communication systems]International electronic design conference (ICED), IEEE, 2014) and literature [10](Zakraria Z, mutalib M A, ismail A et al. Compact Structure of band-pass Filter for Integrated Defect Microstrip Structure (DMS) for broadband applications [ C]. European antennas and transmission conferences. IEEE, 2014) gives an equivalent circuit of parallel RLC lumped elements of DMS resonators. Conventional stopband circuit parameters can be represented by equations (1) - (3), where f c Is the cut-off frequency of the band-stop filter, f 0 Is the resonant frequency, Z 0 Is the characteristic impedance. F is simulated in HFSS software 2 The electromagnetic simulation result of the coplanar waveguide M-shaped DMS filter with the thickness of =12.34mm shows that the resonant frequency f 0 =3.043GHz, cut-off frequency f c =2.975GHz, | S11| = -0.62db = -0.93, characteristic impedance Z 0 Set to 50 omega. From equations (1) - (3), C =11.57nF, L =0.2364nH, and R =1355 Ω can be calculated. The electromagnetic and circuit simulation results of the coplanar waveguide DMS filter are shown in fig. 10, and the two simulation results are matched to verify the correctness of the equivalent circuit. The DMS filter unit has good rejection performance, but is rectangularThe number is not high enough and the selectivity is poor.
(3) Coplanar waveguide mixed type second-order band-stop filter
To ensure a coplanar waveguide transmission line (F) in a broadband antenna 2 =12.34 mm), the invention proposes a coplanar waveguide DMS-DGS hybrid quadratic notch filter, as shown in fig. 11. The equivalent circuit corresponding to the DMS unit and the DGS unit can be understood as that the DMS unit and the DGS unit are connected in series through a section of coplanar waveguide transmission line, as shown in figure 12, and the results of electromagnetic simulation and circuit simulation are shown in figure 13, and the results of the electromagnetic simulation and the circuit simulation are identical. From electromagnetic and circuit simulation results, the filter can realize trap wave characteristics in the frequency range of 2.7-3.2GHz, has two transmission zeros and thus has high selectivity.
FIG. 14 presents simulation results of a DGS notch, DMS notch and proposed hybrid filter, facilitating better comparative analysis. It can be seen that the DGS filter or the DMS filter, which have only one transmission zero, have a low rectangular coefficient and poor selectivity. In the same length as the DMS unit (F) 2 =12.34 mm), the proposed hybrid filter achieves two transmission zeros, a wider stop band frequency range, a larger rectangular coefficient, and a higher notch selectivity. In conclusion, the proposed coplanar waveguide hybrid second-order band-stop filter meets the requirements of high selectivity and miniaturization.
3. Trapped wave antenna based on coplanar waveguide
Fig. 15 (a) and (b) are structural diagrams of a coplanar waveguide broadband antenna and a proposed coplanar waveguide notch antenna, respectively, as a reference. Compared with a broadband antenna, the notch antenna introduces a hybrid second-order band-stop filter under the condition of no change of other designs. The VSWR of these two antennas is shown in fig. 16 (a), and the results show that when the VSWR is less than 1.5, the broadband antenna operates in the frequency band of 1.7-3.6GHz, while the band-stop antenna proposed by the present invention operates in two operating frequency bands of 1.7-2.7GHz and 3.4-3.6GHz, and simultaneously introduces the notch frequency band of 2.9-3.1GHz. And the VSWR of the antenna is more than 10 in the notch frequency band of 2.9-3.1GHz. Fig. 16 (b) shows a comparison of the radiation gains of the two antennas in the main radiation direction, and the average gain of the notch antenna proposed by the present invention is about 8.3dBi in the low frequency band and about 7.1dBi in the high frequency band. Within the stop band, the antenna gains are all lower than 0dBi, and the minimum gain is-14.5 dBi. The result shows that the proposed antenna can realize two working frequency bands of 1.7-2.7GHz and 3.4-3.6GHz when VSWR is less than 1.5, and the notch frequency band is 2.9-3.1GHz.
In conclusion, the invention firstly designs a broadband dual-polarized crossed dipole antenna covering 1.7-3.6GHz by utilizing a coplanar waveguide feed structure. Then, a coplanar waveguide DMS-DGS mixed type second-order band-stop filter is designed, and the miniaturized coplanar waveguide filter with high selectivity and an antenna are printed on the same dielectric substrate, so that a 2.9-3.1GHz stop band is realized. Under the condition that VSWR is less than 1.5, compared with an original broadband antenna covering a frequency band of 1.7-3.6GHz, the notch antenna works at two working frequency bands of 1.7-2.7GHz and 3.4-3.6GHz, and simultaneously introduces the notch frequency band of 2.9-3.1GHz, the average gain of the notch antenna at a low frequency band is about 8.3dBi, and the average gain of a high frequency band is about 7.1dBi. Within the stop band, the antenna gains are all lower than 0dBi, and the minimum gain is-14.5 dBi. Therefore, when the VSWR is less than 1.5, the dual-polarized notch antenna provided by the invention has working frequency bands of 1.7-2.7GHz and 3.4-3.6GHz and a notch frequency band of 2.9-3.1GHz, and can be applied to a 2G/3G/4G/5G base station which needs a notch function.
Although the present invention has been described with reference to the preferred embodiments, it should be understood that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the invention as defined by the appended claims.

Claims (8)

1. The utility model provides a two radiation zero coplanar waveguide double polarization trapped wave crossed dipole antennas which characterized in that: the cross dipole antenna comprises a dielectric substrate, a metal reflecting plate and two coaxial cables, wherein the cross dipole antenna is printed on the upper layer and the lower layer of the substrate and is fed by the two coaxial cables, and the metal reflecting plate is positioned below the dielectric substrate; the crossed dipole antenna comprises an upper metal layer and a lower metal layer, each layer is provided with two pairs of annular dipole antennas, the inside of each annular dipole arm is provided with a rectangular metal layer, and the edge of each antenna arm between every two crossed dipoles is designed into an exponential function shape; two sections of coplanar waveguide transmission lines are arranged in the upper metal layer, a hexagonal capacitive open-circuit branch is arranged along each section of coplanar waveguide transmission line, and an M-shaped DMS coplanar waveguide filter is embedded in each coplanar waveguide transmission line; in the lower metal layer, a double-turn hexagonal DGS resonator, a hexagonal capacitive open-circuit branch and a DGS resonator are arranged right below each hexagonal capacitive open-circuit branch corresponding to the upper metal layer to form a DGS band elimination filter unit.
2. A two-radiation-null coplanar waveguide dual-polarization notch crossed dipole antenna as claimed in claim 1, wherein: the dielectric substrate adopts Rogers-4350, and the dielectric constant of the dielectric substrate is epsilon r =3.66, length of substrate L s =55mm and a thickness H s =1mm, the thickness of the metal layer of the substrate is H m ＝0.035mm。
3. A two-radiation-null coplanar waveguide dual-polarization notch crossed dipole antenna as claimed in claim 1, wherein: the metal reflecting plate is square and is placed 35mm below the dielectric substrate so as to realize a directional radiation mode of about 65 degrees.
4. A two-radiation-null coplanar waveguide dual-polarization notch crossed dipole antenna as claimed in claim 1, wherein: the upper and lower metal layers are connected with each other through 21 short circuit metallized through holes, wherein 20 short circuit metallized through holes are used for connecting the upper and lower layer dipole antennas, and 1 short circuit metallized through hole is used for connecting the coplanar waveguide transmission line.
5. The two-radiation-zero coplanar waveguide dual-polarization notch crossed dipole antenna as claimed in claim 4, wherein: to avoid overlap, a portion of the coplanar waveguide transmission line is printed on the bottom layer of the substrate, and then the top and bottom of the coplanar waveguide transmission line are connected by a short-circuited metalized via.
6. A two-radiation-null coplanar waveguide dual-polarization notch crossed dipole antenna as claimed in claim 1, wherein: the expression of the exponential function is Y (x) = Ce kx + B, where k is a constant coefficient of the exponential function and B is a parameter of the exponential function.
7. A two-radiation-null coplanar waveguide dual-polarization notch crossed dipole antenna as claimed in claim 1, wherein: the outer conductor of the coaxial cable is connected to one arm of the dipole at the bottom of the substrate, the inner conductor of the coaxial cable penetrates through the substrate to be connected with one end of the coplanar waveguide transmission line, and the other end of the coplanar waveguide transmission line is connected with the other arm of the dipole.
8. A two-radiation-null-point coplanar waveguide dual-polarized notch crossed dipole antenna as claimed in any one of claims 1 to 7, wherein: the DMS length is 12.34mm.
CN202211016170.XA 2022-08-24 2022-08-24 Two-radiation-zero coplanar waveguide dual-polarization trapped wave crossed dipole antenna Pending CN115441171A (en)

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Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN116093596A (en) * 2023-01-18 2023-05-09 珠海正和微芯科技有限公司 Millimeter wave broadband package antenna
CN116435772A (en) * 2023-06-15 2023-07-14 东集技术股份有限公司 Miniaturized low-profile dual polarized antenna, antenna assembly and PDA equipment

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN116093596A (en) * 2023-01-18 2023-05-09 珠海正和微芯科技有限公司 Millimeter wave broadband package antenna
CN116093596B (en) * 2023-01-18 2023-09-12 珠海正和微芯科技有限公司 Millimeter wave broadband package antenna
CN116435772A (en) * 2023-06-15 2023-07-14 东集技术股份有限公司 Miniaturized low-profile dual polarized antenna, antenna assembly and PDA equipment
CN116435772B (en) * 2023-06-15 2023-09-01 东集技术股份有限公司 Miniaturized low-profile dual polarized antenna, antenna assembly and PDA equipment

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