CN115435777A - Inertial measurement circuit, corresponding device and method - Google Patents

Abstract

Embodiments of the present disclosure relate to inertial measurement circuits, corresponding devices, and methods. In one embodiment, a circuit includes: an inertial measurement unit configured to oscillate via a drive signal provided by the drive circuitry; a lock-in amplifier configured to receive the sensing signal and a reference demodulation signal as a function of the drive signal from the inertial measurement unit and to provide the inertial measurement signal based on the sensing signal, wherein the reference demodulation signal is affected by the variable phase error; phase meter circuitry configured to receive the drive signal and the sense signal and to provide as a function of a phase difference between the drive signal and the sense signal; a phase correction signal for the reference demodulated signal and a correction node configured to apply the phase correction signal to the reference demodulated signal such that the phase error is maintained around a reference value in response to applying the phase correction signal to the reference demodulated signal.

Description

Inertial measurement circuit, corresponding device and method

Cross Reference to Related Applications

The application claims the benefit of italian application No.102021000014621 filed on 6/4/2021, which is hereby incorporated by reference.

Technical Field

The present description relates to Inertial Measurement Units (IMUs). The present description also relates to micro-electromechanical systems (MEMS) sensors (e.g., gyroscopes), which are examples of IMUs. Furthermore, the present description relates to automotive applications, in particular to navigation systems.

Background

Low offset drift performance is a desirable feature of Inertial Measurement Units (IMUs) such as MEMS gyroscopes.

MEMS gyroscopes are used today in a wide range of products, such as consumer products, including augmented reality/virtual reality (AR/VR) applications, automotive and transportation (navigation) and other high-end applications where offset stability is a concern.

Leakage of the "quadrature" offset caused by the phase error into the output channels of the gyroscope may adversely affect stability performance.

This problem can be solved by minimizing the quadrature value. This can be done electronically or electromechanically. In either case, the compensation is incomplete and the residual drift is caused by drift in the phase error.

As a general background (and as known to those skilled in the art), gyroscopes are used to detect angular (rotation) rates with high accuracy (low offset drift) that represents the desired feature.

As schematically shown in fig. 1, the main source of offset drift is believed to be the leakage of unwanted orthogonal motion qq into the sense channel.

This can be minimized by coherent demodulation. However, the demodulation phase is affected by an error Φ err, which varies with temperature (T) and results in an output offset Ω q.

Acting directly on the system to compensate for the quadrature q has certain benefits. However, the compensation is imperfect and the residual quadrature qqr remains, so that the residual error qqr Φ err (T) remains and its drift adversely affects the gyroscope performance.

Various methods have been proposed to solve this problem.

For example, one may attempt to minimize the quadrature component using current injection at the analog front end (e.g., Ω qr. Φ err (T) - Ω comp).

A disadvantage of this method is that it leaves residual quadrature error and hence offset. In fact, in the absence of a perfect match between the quadrature phase and its compensation phase, a residual error occurs that is proportional to the phase error (Φ err). In addition, the drift remains substantially uncompensated.

Another approach may include minimizing the quadrature component, e.g., (Ω q- Ω comp) Φ err (T), by using closed-loop electromechanical compensation.

The disadvantage of this method is that it compensates for the orthogonality at its source, leaving in any case a residual uncompensated term, and thus the associated drift.

Another method may include minimizing phase error drift by varying the phase in dependence on the linearized phase v. Temperature models, e.g., Ω q [ Φ err (T) - Φ model (T) ].

The disadvantage of this approach is that it relies on a linearized model, whereas the behavior of the actual system is non-linear (it varies from component to component) and does not take into account other effects (sensing softening, drive hardening, deformation) that contribute to the phase drift.

Thus, the problem of enhancing Zero Rate Level (ZRL) stability over the life of the system is still widely perceived.

Therefore, a solution is needed that facilitates higher ZRL stability without limiting noise and power consumption performance.

Disclosure of Invention

Embodiments provide an Inertial Measurement Unit (IMU). Further embodiments provide a micro-electro-mechanical system (MEMS) sensor (e.g., a gyroscope), which is an example of an IMU. Other embodiments relate to automotive applications, and more particularly to navigation systems. Embodiments may solve the above-described problems.

In accordance with one or more embodiments, a circuit may have the features set forth below.

MEMS (micro-electromechanical system) gyroscope based circuits may be examples of such circuits.

One or more embodiments relate to a corresponding apparatus.

Various types of consumer products, such as augmented reality/virtual reality (AR/VR) viewer viewing, navigation devices for the automotive industry including user circuitry utilizing output from an IMU may be examples of such devices.

One or more embodiments relate to a corresponding method.

The claims are an integral part of the technical teaching provided herein with respect to the embodiments.

Examples as discussed herein may provide one or more of the following advantages:

due to the underlying feedback mechanism, the phase error can remain close to zero at any time; therefore, no component of the quadrature value leaks into the signal path;

in certain embodiments, the solution as discussed herein may be combined with other compensation mechanisms and provide further improved overall compensation;

using a closed-loop method that on the one hand keeps the phase error close to zero and on the other hand does not take into account the variability of e.g. the gyroscope parameters, which makes the temperature ("T") dependency partly different from partly;

in some embodiments, for example, as applied to a gyroscope, no compensation is required at the gyroscope level; this has significant advantages over conventional techniques.

Examples discussed herein propose a closed loop compensation method to improve Zero Rate Output (ZRO) stability performance of Amplitude Modulation (AM) capacitive MEMS gyroscopes.

The examples discussed herein rely on direct measurement of the relative phase change between the quadrature signal modulated by the drive carrier frequency and the demodulation reference (and closed loop compensation).

In the examples discussed herein, the closed loop arrangement includes a phase meter, an open loop, a demodulation chain, which acts on the phase of the lock-in amplifier in the readout mode.

The examples discussed herein may be applied in MEMS gyroscopes, for use in the automotive field or in augmented reality/virtual reality (AR/VR) applications.

The examples discussed herein may be applied to other conventional gyroscopes and Inertial Measurement Units (IMUs), such as 6 xmimus, with enhanced capabilities to represent "mainstream" inertial MEMS products.

The examples discussed herein do not relate to substantial variations in the electromechanical design of such conventional products.

Examples discussed herein may include additional (analog) stages that take signals out of the drive/sense chain and square them. These may include, for example, dual paths implementing a Band Pass Filter (BPF) stage (with associated passive components) operating at the gyroscope operating frequency.

Other additional (analog) stages and related signals may be present along with the phase meter in the path towards the lock-in amplifier (LIA) reference.

In operation, examples discussed herein may produce an output tone associated with periodic activation of a compensation feature.

Furthermore, in certain examples, periodic variations in the sign of the voltage applied to the MEMS quadrature compensation electrode may be detected.

Drawings

One or more embodiments will now be described, by way of example only, with reference to the accompanying drawings, in which:

FIG. 1 shows a representation of a gyroscope output;

FIG. 2 shows a block diagram of a circuit according to an example discussed in this specification;

FIG. 3 shows a block diagram illustrating a circuit according to an embodiment; and

fig. 4 shows a time diagram illustrating a possible time behavior of a signal that may occur in an embodiment.

Corresponding numerals and symbols in the various drawings generally refer to corresponding parts unless otherwise indicated. The drawings are drawn for clarity of illustrating relevant aspects of the embodiments and are not necessarily to scale. The edges of a feature drawn in the drawings do not necessarily represent the end of the range of the feature.

Detailed Description

In the following description, numerous specific details are set forth in order to provide a thorough understanding of various examples of embodiments according to the description. Embodiments may be practiced without one or more of the specific details, or with other methods, components, materials, etc. In other instances, well-known structures, materials, or operations are not shown or described in detail to avoid obscuring aspects of the embodiments.

Reference to "one embodiment" or "an embodiment" within the framework of the specification is intended to indicate that a particular configuration, structure, or characteristic described in connection with the embodiment is included in at least one embodiment. Thus, phrases such as "in an embodiment," "in one embodiment," and the like that may be present in various points of the specification do not necessarily refer to the same embodiment with certainty. Furthermore, the particular configurations, structures, or characteristics may be combined in any suitable manner in one or more embodiments.

The headings/references used herein are provided for convenience only and thus do not define the scope of protection or the scope of the embodiments.

For the sake of brevity and simplicity, the same reference numerals may be used below to designate a certain circuit node or line and a signal (e.g., a voltage signal) appearing at the node or line.

By way of a summary of the instant detailed description, reference may be made to the document US2020/400434A1. This document discloses a MEMS gyroscope having a moving mass carried by a support structure to move in a drive direction and a first sense direction perpendicular to each other.

The structure of such a gyroscope will be assumed to be well known to those skilled in the art. For the sake of brevity, the detailed description is not repeated herein.

For the purposes herein, the principle of operation of such gyroscopes may be briefly reviewed: even if the support of the vibrating body is rotated, the vibrating body tends to continue vibrating in the same plane. Due to the coriolis effect, the vibrating body exerts a significant force on the support. The rate of rotation can be determined by measuring the displacement resulting from the force.

Reliable and inexpensive vibrating structure gyroscopes can be made with MEMS technology. These applications are found in mobile communication devices, electronic games, cameras and various other applications.

An overview of MEMS gyroscope technology is described in a.a.trusov: "Overview of MEMSs Gyroscopies: history, principieSof Operations, typeSof MeasurementS" microsystems laboratory, mechanical and Daerosporace engineering university of California, irvine, CA,92697, USA, may 10 days 2011 (see uch. Edu).

And, at s.facchi, l.guerinoni, l.g.falorni, a.donadel and c.valzasina: "Development of a complete model to estimate the zero rate level above temperature in a MEMS differential simulation" 2017IEEE international symposium on inert sensor SanDSystesmS (INERTIAL), 2017, pp.125-128, doi: a comprehensive model is proposed in 10.1109/isiss.2017.7935673 to analytically estimate the Zero Rate Level (ZRL) variation with temperature in a micromachined Coriolis Vibration Gyroscope (CVG) with associated electronics, with the purpose of providing robust guidance for the development of high stability MEMS Inertial Measurement Units (IMU).

The examples presented herein rely on compensation methods based on direct measurement of the phase error Φ err.

In particular, the illustrated method utilizes orthogonal channels to recover phase information.

Note that the quadrature channel signal is often (much) larger than the rate full-scale signal.

This is especially the case if embedded quadrature compensation electrodes are used to purposefully de-compensate the phase. These have been found to tend to increase the "chopping" or modulation in order to increase the resolution in the phase measurement.

Various implementation options may be considered.

In some examples, no compensation electrodes are provided and the measurement relies on quadrature signals (which inevitably) are present in the system output.

In other examples, by providing quadrature compensation electrodes, quadrature may be intentionally added (or modulated) to facilitate phase measurements.

In either case, the process may be applied in a continuous time mode, or only at certain times (each occurrence) to account for the temperature drift transients involved.

The basic principle therefore aims to continuously keep the term Φ err at zero (zero) via closed-loop feedback, so that the offset and its associated drift are zero.

In a conventional, uncompensated configuration as shown in fig. 2 (considered by way of example to be mounted on the vehicle V, as in the case of fig. 3), the circuit 10 considered herein comprises (in fact built around) a MEMS gyroscope 12 (of any known type suitable for the purposes herein).

During the drive mode, the gyroscope 12, represented here in an intentionally simplified manner, remains oscillating in the drive direction (which may be assumed horizontal in the figure) via a main phase loop including first electrodes 14a,14b driven by drive stages 16A (D +) and 16B (D-), which drive stages 16A (D +) and 16B (D-) are in turn coupled to a Variable Gain Amplifier (VGA) 18.

The second electrodes 20a,20b, which are sensitive to oscillation in the drive direction, generate signals that are applied to the differential stage 22, which differential stage 22 provides phase adjuster/gain stages 24, 26. The output from the gain stage closes a loop to the MEMS 12 via the variable gain amplifier 18.

The amplitude of the oscillation held by the main phase loop just discussed is precisely controlled by an Automatic Gain Control (AGC) negative feedback loop that includes a rectifier circuit 28 coupled to the output from the differential amplifier stage 22.

The output from rectifier circuit 28 is applied to a first input (e.g., inverting) of comparator 32 via low pass filter 30, and a second input (e.g., non-inverting) of comparator 32 is coupled to reference voltage Vref.

The output of comparator 32 in turn controls the gain of variable gain amplifier 18.

As shown, sensing is performed via an open loop chain including third electrodes 34a,34b, which is assumed vertical in the figure).

The signals generated by the third electrodes 34a,34b are applied to a full differential stage 36, which full differential stage 36 provides differential output signals S +, S "to a lock-in amplifier (LIA) 38, which lock-in amplifier 38 generates the desired output signal Vout.

The LIA is an amplifier that is able to extract a signal with a known carrier from a (rather) noisy environment. It may be implemented as a homodyne detector with cascaded adjustable low-pass filters.

As shown in fig. 2, the LIA38 uses a demodulation reference signal Φ opt obtained from the drive loop (e.g., from the output of the variable gain amplifier 18).

In FIG. 2, the phase of the signal is at T ₀ Is designated as Φ opt to highlight that its phase value is at a certain temperature T ₀ And thus may be exposed to temperature-dependent changes.

Those skilled in the art will appreciate that the conventional implementation shown in fig. 2 is merely exemplary.

For both the drive chain and the sense chain associated with the gyroscope 12, a number of alternative implementations are possible: indeed, the examples provided herein are substantially "transparent" to the particular implementation of the drive chain and/or sense chain.

Note that in principle, the "optimal" demodulation phase can be calibrated (find Φ opt under reference conditions). Due to both the mechanical (mainly MEMS gyroscope 12) and electronic components in the drive and sense chains, drift in Φ opt will adversely affect the output signal Vout in operation.

Fig. 3 illustrates the basic principle of the embodiment.

As noted above, corresponding numbers and symbols in different drawings generally refer to corresponding parts, unless otherwise indicated. Accordingly, components or elements that are similar to components or elements already discussed in connection with fig. 2 are indicated by similar numerals and symbols in fig. 3, and the corresponding description will not be repeated for the sake of brevity.

It should be understood that the particular parts or elements indicated by the same numerals or symbols in fig. 2 and 3 do not imply that such parts or elements must be implemented in the same manner in fig. 2 and 3.

Furthermore, in fig. 3 a user circuit arrangement is shown, denoted UC.

This may be any type of circuit configured to utilize an inertial measurement (gyroscope) output signal Vout in, for example, an augmented reality/virtual reality (AR/VR) viewer (e.g., a product of a navigation device in the automotive industry).

As shown in fig. 3, a phase meter 40 is provided to measure a phase difference Φ er between a signal Dsq indicating a "drive phase" and a signal Ssq indicating a "sense phase".

As shown in fig. 3, across the first electrodes 14a,14b (D +, D-), e.g. at the outputs of drivers 1698, 1693 b, the signal Dsq is sensed and passed to the phase meter 40 (after possible bandpass filtering at 42a, 42b) via an instrumentation amplifier (INA) 44 and a high gain (hiG) stage 46.

As also shown in fig. 3, the signal Ssq is sensed across third electrodes 34a,34b (S +, S-) and passed to the phase meter 40 (after possible bandpass filtering at 48a, 48b) via an instrumentation amplifier (INA) 50 and a high gain (hiG) stage 52.

The output from the phase meter 40 is a measure of the phase difference between the signals Dsq and Ssq (see Δ Φ dS in fig. 4), denoted Φ er in fig. 3.

At T ₀ The reference phase delay Φ er0 at (a) is added (signed, i.e., subtracted) to the output of the phase meter 40 at the trim node 54, and the result is used to correct for T at node 56 ₀ The phase of the signal Φ opt at.

The LIA phase adjuster thus closes the feedback on the phase of the reference demodulation wave entering the lock-in amplifier 38, so that Φ er (the phase difference between the signals Dsq and Ssq) is eventually kept close to Φ er0 (reference phase delay) by the feedback action.

Briefly, the exemplary circuit 10 of fig. 3 includes an inertial measurement unit 12 (e.g., a MEMS gyroscope), the inertial measurement unit 12 configured to oscillate via a drive signal, e.g., the signal D +, D-, generated by the drive circuit devices including the elements 14a,14b,16, 18, 20a,20b,22, 24, 26, 28, 30, and 32.

The lock-in amplifier 38 receives the sensing signal S +, S-from the inertia measurement unit 12 and the reference demodulation signal based on the driving signal D +, D-, which ultimately generates the signal Dsq.

The lock-in amplifier 38 is configured to generate the inertial measurement signal Vout based on the sensing signal S +, S-from the inertial measurement unit 12 and a reference demodulation signal, which is affected by the variable phase error Φ er.

The phase meter circuitry 40 is configured to receive the drive signal (D +, D- > > Dsq) and the sense signal (S +, S- > > Ssq) and generate a phase correction signal (via nodes 54, 56) for the reference demodulation signal of the lock-in amplifier 38 based on the phase difference Δ Φ ds between the drive signal Dsq and the sense signal Ssq (see fig. 4).

A correction node such as 56 is provided that is configured to apply such a phase correction signal to the reference demodulated signal of the lock-in amplifier 38.

The associated phase error Φ er is maintained near a (constant) reference value (i.e., Φ er 0) in response to a phase correction signal applied to the reference demodulation signal of lock-in amplifier 38.

A trim node 54 may be provided between the phase meter circuitry 40 and the correction node 56, the trim node 54 being configured at T ₀ The phase correction signal is trimmed to a reference value Φ er0.

The circuit 10 as shown in FIG. 3 includes sensing circuitry intermediate the inertial measurement unit 12 and the lock-in amplifier 38 that includes electrodes 34A,34B configured to generate the sense signal S +, S- > > Ssq.

As discussed below, while advantageous for reliability, such orthogonal electrodes (and pads) are not mandatory and may be omitted, for example, in the On/active/On (On/Act/On) option combined with electronic compensation, as discussed below.

As shown in FIG. 3, the sensing circuitry 34A,34B,36 may be configured to provide both the sense signals S +, S- > > Ssq to the lock-in amplifier 38 and the phase meter 40.

Advantageously, the phase meter 40 may be implemented in the digital domain (see fig. 4).

As shown in fig. 3, the phase meter circuit arrangement 40 is coupled to the input signal path for the drive signals D +, D-, dsq and the sense signals S +, S-, ssq.

As shown, these input signal paths include:

bandpass filter circuitry (e.g., filters 42a,42b,48a, 48b), and/or

The saturating circuit arrangement optionally comprises a cascade arrangement of an instrumentation amplifier such as 44 or 50 and a high gain stage such as 46 or 52.

The LIA38 with phase adjuster can be implemented in the analog or digital domain.

The pre-filtering (BPF) stages 42A,42B,48A,48B, INA amplifiers 44, 50 and hiG stages 46 and 52 are analog stages.

As shown, pre-filtering (BPF) and saturation (INA + hiG) are applied to the sensing and reference signals before (upstream of) the phase meter. This was found to improve the resolution of the phase measurement, reducing noise folding.

The architecture illustrated in FIG. 3 is suitable for use in a variety of ways.

In the On/Act/Off approach, in combination with electronic compensation, quadrature ("On" step) and measurement and correction of the optimal phase ("Act" step) can be intentionally added. In the final step, the intentional increase in orthogonality is removed and the orthogonality itself is compensated again (the "off" step).

This option in principle does not require dedicated orthogonal electrodes and pads.

In a stand-alone approach that continuously applies quadrature modulation, the orthogonality may be intentionally increased again, applying "chopping" at a frequency above the sensing bandwidth of the coriolis channel. The quadrature of the modulation can be used to detect the phase and apply a (continuous) correction.

Of course, this option involves orthogonal electrodes and pads.

Another On/action/Off (On/Act/Off) method in combination with electromechanical compensation may be similar to the first method discussed previously, where orthogonality is again increased and compensated using electromechanical methods.

This (more reliable) selection again involves orthogonal electrodes and pads.

Thus, examples discussed herein improve zero-rate output (zor) stability of Inertial Measurement Units (IMUs), such as MEMS gyroscopes, by substantially intervening at the hardware level.

However, the examples discussed herein are suitable for cooperation with appropriate software acting on the phase meter and phase adjuster (as shown in blocks 38 and 40 in fig. 3).

It should be understood that the apparatus illustrated herein may be applied to a multi-axis, multi-parameter IMU, the description being expressly simplified for purposes of explanation and understanding.

Without prejudice to the underlying principles, the details and the embodiments may vary, even significantly, with respect to what has been described by way of example only, without departing from the scope of protection. The scope of protection is determined by the appended claims.

Claims (13)

1. A circuit, comprising:

an inertial measurement unit configured to oscillate via a drive signal provided by a drive circuit arrangement;

a lock-in amplifier configured to:

receiving a sensing signal from the inertial measurement unit and a reference demodulation signal dependent on the drive signal;

providing an inertial measurement signal based on the sense signal, wherein the reference demodulation signal is subject to a variable phase error;

phase meter circuitry configured to:

receiving the drive signal and the sense signal; and

providing a phase correction signal for the reference demodulation signal in dependence on the phase difference between the drive signal and the sense signal; and

a correction node configured to apply the phase correction signal to the reference demodulation signal such that the phase error is maintained near a reference value in response to the phase correction signal being applied to the reference demodulation signal.

2. The circuit of claim 1, further comprising: a trim node disposed between the phase meter circuitry and the correction node, wherein the trim node is configured to trim the phase correction signal by a reference phase correction value.

3. The circuit of claim 1, further comprising: a sensing circuitry disposed between the inertial measurement unit and the lock-in amplifier, wherein the sensing circuitry is configured to provide the sensing signal.

4. The circuit of claim 3, wherein the sensing circuitry is configured to provide the sensing signal to both the lock-in amplifier and the phase meter circuitry.

5. The circuit of claim 1, wherein the phase meter circuitry is coupled to input signal paths for the drive signal and the sense signal, respectively, the input signal paths comprising:

a band-pass filter circuit means; and/or

A saturation circuit arrangement.

6. The circuit of claim 5, wherein the saturation circuit means comprises a cascade arrangement of an instrumentation amplifier and a high gain stage.

7. The circuit of claim 1, wherein the phase meter circuitry comprises a digital phase meter.

8. The circuit of claim 1, wherein the inertial measurement unit comprises a MEMS gyroscope.

9. A device, comprising:

the circuit of claim 1; and

a user circuit device coupled to the lock-in amplifier in the circuit,

wherein the user circuitry is configured to utilize the inertial measurement signal generated by the lock-in amplifier.

10. A method, comprising:

oscillating the inertial measurement unit via a drive signal provided by the drive circuit arrangement;

receiving, by a lock-in amplifier, a sensing signal from the inertial measurement unit and a reference demodulation signal from the driving signal;

providing, by the lock-in amplifier, an inertial measurement signal based on the sense signal and the reference demodulation signal, wherein the reference demodulation signal is subject to a variable phase error;

receiving, by phase meter circuitry, the drive signal and the sense signal, and providing a phase correction signal for the reference demodulated signal in dependence upon a phase difference between the drive signal and the sense signal; and

applying, by a correction node, the phase correction signal to the reference demodulation signal of the lock-in amplifier such that the phase error is maintained near a reference value in response to the phase correction signal being applied to the reference demodulation signal of the lock-in amplifier.

11. The method of claim 10, further comprising: the phase correction signal is trimmed to a reference phase correction value by a trimming node.

12. The method of claim 10, further comprising: the sensing signal is provided by sensing circuitry disposed between the inertial measurement unit and the lock-in amplifier.

13. The method of claim 12, wherein the sensing circuitry provides the sensing signal to both the lock-in amplifier and the phase meter circuitry.

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