CN115276256B - Bidirectional MC-WPT system and constant-current output phase-shifting control method thereof - Google Patents

Abstract

The present invention relates to the technical field of magnetic coupling wireless power transmission. The present invention relates to the technical field of MC-WPT, and specifically discloses a two-way MC-WPT system and its constant current output phase-shifting control method. The ground terminal communication module and the ground terminal current detector, and the vehicle terminal controller, the vehicle terminal communication module and the vehicle terminal current detector are set at the vehicle terminal, so that the energy is forwarded to the vehicle terminal as the energy transmitter at the ground terminal When transmitting in the opposite direction, the current I ₂ flowing through the vehicle-mounted battery is obtained by the vehicle-mounted current detector, and transmitted to the ground-side controller, so that the ground-side controller calculates the phase shift angle θ corresponding to ΔI according to the PI algorithm to act on The full-bridge conversion module at the ground end performs phase-shift control to realize constant current output to the vehicle end. Conversely, when the vehicle end transmits energy back to the ground end, it is similar to this process. Thus, the bidirectional constant current output control of the bidirectional MC‑WPT system is realized.

Description

Bidirectional MC-WPT system and its constant current output phase-shift control method

technical field

The invention relates to the technical field of magnetically coupled wireless power transmission (MC-WPT), in particular to a bidirectional MC-WPT system and a constant current output phase-shift control method thereof.

Background technique

Wireless Power Transfer (WPT) technology refers to a technology that uses a space medium carrier instead of the traditional wire contact method to realize the power transmission between the power supply and the load end. The space medium carrier includes magnetic field, electric field, laser, microwave, etc. , ultrasound, etc. Magnetic Coupling Wireless Power Transfer (MC-WPT) is currently a hot topic in the field of WPT technology. MC-WPT technology uses magnetic field as the transmission medium, and the wireless transmission of electric energy between the primary and secondary sides is through high-frequency The mutual conversion between alternating magnetic field and electric energy is realized. Since its birth, this technology has attracted the attention and research of scholars at home and abroad, and its application fields are becoming more and more extensive. At present, MC-WPT has been successfully applied to consumer electronics, biomedicine, electric vehicles, underwater and special places, etc. field.

The total amount of power supply at the grid side remains unchanged, while there are peak periods and valley periods at the power consumption side. The peak period of power consumption is generally during the day, and the power supply may not be enough, and at night it is a valley, and there is a certain amount of waste of power. Such a large number of electric vehicles enhances their energy interaction with the grid, that is, electric vehicle to grid technology (Vehicle to Grid, V2G), which is used to charge electric vehicles during the low power consumption period at night, and idle electric vehicles during peak periods during the day The energy is fed back to the power grid to cut peaks and fill valleys, improve the utilization rate of electric energy, and achieve a more balanced and effective use of energy. If the charging and discharging process between the electric vehicle and the power grid adopts MC-WPT technology, it will undoubtedly improve the interactive ability of the electric vehicle and the power grid, making the whole system more convenient, efficient, flexible and reliable, so as to make V2G more intelligent and fully utilized role. Therefore, the need for research on two-way magnetically coupled wireless power transfer (two-way MC-WPT) technology is becoming more and more urgent.

The research on unidirectional MC-WPT technology is very mature and has been widely used in various fields. The bidirectional MC-WPT technology is developed from the unidirectional MC-WPT technology, which can realize the interaction and networking of electric energy at the power supply and load terminals. , has received extensive attention and research in recent years.

For the two-way MC-WPT system of electric vehicles, when the energy is transmitted in the forward direction, when the energy is transmitted from the grid side to the load wirelessly, the load of different electric vehicles is different, and the load change or the fluctuation of the mutual inductance of the coil will cause the output characteristics of the system to be unstable. When the energy is reversely transmitted, although there is no load change problem on the grid side, mutual inductance fluctuations still exist, and the system output is also unstable.

For the bidirectional MC-WPT system, the structure design of the coupling mechanism plays a decisive role in the power transmission and anti-offset capability of the whole system. At present, the research results on the bidirectional MC-WPT system are fruitful, but the characteristics of the coupling mechanism and anti-offset characteristics are still limited in the forward and reverse transmission of the bidirectional MC-WPT system, and further improvement and optimization are still needed.

Contents of the invention

The present invention provides a bidirectional MC-WPT system and its constant current output phase-shift control method. The technical problem to be solved is: how to realize the constant current output of the bidirectional MC-WPT system, and how to optimize the magnetic coupling mechanism of the bidirectional MC-WPT system , so that it has a high level of anti-offset capability to ensure stable and efficient transmission performance.

In order to solve the above technical problems, the present invention firstly provides a MC-WPT system, including a ground terminal and a vehicle terminal;

The ground terminal includes a sequentially connected ground terminal DC power supply, a ground terminal full-bridge conversion module, a ground terminal resonant network, and a ground terminal coupling structure, and also includes a ground terminal controller connected to the ground terminal full bridge conversion module, and a ground terminal controller connected to the ground terminal. The ground-side communication module and the ground-side current detector of the ground-side controller;

The vehicle-mounted terminal includes sequentially connecting the vehicle-mounted terminal coupling structure, the vehicle-mounted terminal resonant network, the vehicle-mounted full-bridge conversion module, the filter circuit and the vehicle-mounted battery, and also includes a vehicle-mounted controller connected to the vehicle-mounted full-bridge conversion module, And a vehicle-mounted communication module and a vehicle-mounted current detector connected to the vehicle-mounted controller;

Both the ground-side full-bridge conversion module and the vehicle-mounted full-bridge conversion module include a full-bridge inverter and four power diodes connected in antiparallel with the four switching tubes of the full-bridge inverter;

When the ground terminal is used as the energy transmitting terminal to transmit energy forward to the vehicle terminal, the ground terminal full-bridge conversion module works in the inverter state to convert the ground terminal DC power supply into high-frequency AC power, and the The vehicle-mounted full-bridge conversion module is working in a rectification state, the four switch tubes of the vehicle-mounted full-bridge conversion module are turned off, and the vehicle-mounted full-bridge rectifier circuit is formed by corresponding four power diodes connected in reverse parallel. The high-frequency alternating current induced by the terminal coil is rectified into direct current, and finally supplied to the vehicle-mounted battery for charging; the vehicle-mounted current detector obtains the current I ₂ flowing through the vehicle-mounted battery, and is sent by the vehicle-mounted communication module To the ground end communication module, the ground end controller _compares I with the ground reference current value _Iset , calculates the phase shift angle θ, thereby generating a switching tube PWM drive signal with a phase shift angle of θ, acting on The ground end full-bridge conversion module;

Based on the same principle as the energy forward transmission process, the vehicle-mounted terminal acts as an energy transmitting terminal to perform energy reverse transmission to the ground terminal.

Preferably, the ground end controller compares I ₂ with the ground reference current value I _set to obtain a difference ΔI, and calculates the phase shift angle θ corresponding to ΔI according to the PI algorithm.

Preferably, both the resonant network at the ground end and the resonant network at the vehicle end use an LCC resonant network to form an LCC-LCC resonant topology.

Preferably, the LCC-LCC resonant topology satisfies the conditions:

Among them, w represents the operating angular frequency of the system, L _r1 , C _p , and C _r1 respectively represent the parameter values corresponding to the inductance, series capacitance, and parallel capacitance in the LCC resonant network at the ground end, and L _r2 , Cs, and C _r2 represent the Describe the parameter values corresponding to the inductance, series capacitance and parallel capacitance in the LCC resonant network on the vehicle side.

Preferably, the ground-end controller is provided with a drive circuit, the drive circuit is composed of two-stage drive, and the dual-channel drive chip is used as the first-stage drive for the 3.3V drive pulse signal output by the ground-end controller PWM1 and PWM2 are upgraded to 12V drive signals PWMH and PWML; the second-level drive is composed of an optocoupler isolation drive chip and its peripheral circuits, the output is connected to the totem pole drive structure, and the final output +18V/-3V is used to control the ground The gate drive voltages G_S and H_S for power MOSFET turn-on and turn-off in the end-to-full bridge conversion module;

The setting of the vehicle-mounted controller is the same as that of the ground controller.

Preferably, the ground-side coupling structure includes a ground-side coil, and the vehicle-side coupling structure includes a vehicle-side coil;

The ground-side coil is formed in series by an external close-wound coil wound by a close-wound method and an internal sparse-wound coil wound in the same direction as the external close-wound coil by a sparse-wound method; It is wound in a double-layer close-wound manner.

Preferably, the ground-side coil and the ground-side coil are wound with Litz wire of the same specification.

Preferably, the ground-side coupling structure includes a ground metal shielding plate, a ground magnetic core, and the ground-side coil arranged in layers from bottom to top;

The vehicle-mounted coupling structure includes the vehicle-mounted coil, the vehicle-mounted magnetic core, and the vehicle-mounted metal shielding plate arranged from bottom to top;

Both the ground magnetic core and the vehicle magnetic core are spliced by a plurality of square magnetic cores.

The present invention also provides a constant-current output phase-shift control method for a bidirectional MC-WPT system, which is characterized in that it includes a ground phase-shift control step for forward constant current output from the ground terminal to the vehicle terminal, and The on-vehicle phase-shifting control step of performing reverse constant current output to the ground end by the vehicle-mounted terminal, the ground phase-shifting control step comprising:

A1. Acquiring the current I ₂ flowing through the vehicle-mounted battery;

A2. Comparing the current I ₂ with the ground reference current value I _set to obtain the difference ΔI;

A3, judge whether ΔI is 0, if so, do nothing, if not, calculate the phase shift angle θ corresponding to ΔI according to the PI algorithm, and output a PWM driving signal with a phase shift angle θ to act on the full-bridge conversion module at the ground end;

The principle of the vehicle-mounted phase shift control step is the same as that of the ground phase shift control step.

The present invention provides a bidirectional MC-WPT system and its constant current output phase-shifting control method, by setting the ground terminal controller, the ground terminal communication module and the ground terminal current detector at the ground terminal, and setting the vehicle terminal controller at the vehicle terminal , the vehicle-mounted terminal communication module and the vehicle-mounted terminal current detector, so that when the ground terminal is used as the energy transmitting terminal to carry out energy forward transmission to the vehicle-mounted terminal, the current flowing through the vehicle-mounted terminal battery is obtained through the vehicle-mounted terminal current detector I ₂ , and send it to the ground-side controller, so that the ground-side controller calculates the phase-shift angle θ corresponding to ΔI according to the PI algorithm, and acts on the full-bridge conversion module of the ground-side to perform phase-shift control, so as to realize the constant current output to the vehicle-side. Conversely, when the vehicle end transmits energy back to the ground end, it is similar to this process. Thus, the bidirectional constant current output control of the bidirectional MC-WPT system is realized.

In the two-way MC-WPT system, the ground-side coil adopts dense outer coils and sparse inner coils (including outer densely wound coils and inner sparsely wound coils). For basic power transmission requirements, the sparse winding of the internal coil can make the magnetic field distribution more uniform, and comprehensively improve the anti-offset characteristics of the coil. It is worth mentioning that, compared with the traditional densely wound coil, the outer dense inner sparse coil can reduce the wire consumption and cost of the coil, reduce the coil self-inductance and internal resistance, thereby effectively reducing the voltage at both ends of the coil and coil loss. The vehicle-side coil is wound in a multi-layer dense winding method, which can improve the self-inductance and mutual inductance of the vehicle-side coil to ensure that the energy transmission can meet the corresponding performance requirements.

Description of drawings

Fig. 1 is a schematic diagram of the ground terminal coil in the magnetic coupling mechanism of the bidirectional MC-WPT system provided by the embodiment of the present invention;

Fig. 2 is a schematic diagram of the vehicle-mounted end in the magnetic coupling mechanism of the two-way MC-WPT system provided by the embodiment of the present invention;

3 is a trend diagram of mutual inductance and its rate of change when the x-axis of the four coils provided by the embodiment of the present invention is offset;

Fig. 4 is a trend diagram of the coupling coefficient and its rate of change when the x-axis of the four coils provided by the embodiment of the present invention is offset;

Fig. 5 is a front view of the magnetic coupling mechanism provided by the embodiment of the present invention;

Fig. 6 is a flow chart of the parameter design method of the magnetic coupling mechanism of the two-way MC-WPT system provided by the embodiment of the present invention;

Fig. 7 is a trend diagram of the mutual inductance and the rate of change of the coupling coefficient of the four coils provided by the embodiment of the present invention at the limit offset;

Fig. 8 is a structural diagram of a two-way MC-WPT system provided by an embodiment of the present invention;

Fig. 9 is a situation diagram of the forward energy transmission of the two-way MC-WPT system provided by the embodiment of the present invention;

Fig. 10 is a situation diagram of the reverse energy transmission of the two-way MC-WPT system provided by the embodiment of the present invention;

Fig. 11 is a schematic diagram of a phase-shift control circuit for a full-bridge inverter provided by an embodiment of the present invention;

Fig. 12 is a waveform diagram of switching tube drive and inverter output voltage and current provided by an embodiment of the present invention;

Fig. 13 is a simulation waveform diagram of load switching during energy forward transmission provided by an embodiment of the present invention;

Fig. 14 is a simulation waveform diagram of mutual inductance changes during energy forward transmission provided by an embodiment of the present invention;

Fig. 15 is a simulation waveform diagram of mutual inductance changes during energy reverse transmission provided by an embodiment of the present invention;

Fig. 16 is a comparison diagram between the simulated value of M and the measured value when the coupling mechanism provided by the embodiment of the present invention is shifted in the directions of x and y axes;

Fig. 17 is a working flowchart of the closed-loop constant current control provided by the embodiment of the present invention;

Fig. 18 is a driving circuit diagram provided by an embodiment of the present invention;

Fig. 19 is an experimental waveform diagram during energy forward transmission provided by an embodiment of the present invention;

Fig. 20 is a diagram of the output power and efficiency of the system when the energy is forwardly transmitted according to the embodiment of the present invention;

Fig. 21 is a waveform diagram of a load switching experiment during energy forward transmission provided by an embodiment of the present invention;

Fig. 22 is an experimental waveform diagram of the system during reverse energy transmission provided by the embodiment of the present invention;

Fig. 23 is a diagram of output power and efficiency of the system during reverse energy transmission provided by an embodiment of the present invention.

Detailed ways

The embodiment of the present invention will be explained in detail below in conjunction with the accompanying drawings. The examples given are only for the purpose of illustration, and cannot be interpreted as limiting the present invention. The accompanying drawings are only for reference and description, and do not constitute the scope of patent protection of the present invention. limitations, since many changes may be made in the invention without departing from the spirit and scope of the invention.

Embodiments of the present invention first provide a magnetic coupling mechanism for a bidirectional MC-WPT system, including a ground-side coupling structure and a vehicle-side coupling structure. The ground-side coupling structure includes a ground-side coil, and the vehicle-side coupling structure includes a vehicle-side coil.

As shown in Figure 1 and Figure 2, the ground-side coil consists of an outer densely wound coil (coil 1) wound by a densely wound method and an inner sparsely wound coil (coil 1) wound in the same direction as the outer densely wound coil by a sparsely wound method. The coils 2) are connected in series, and the vehicle-mounted coils are wound in a multi-layer dense winding manner. The number of turns of the outer densely wound coil is N1, and the outer dimension is x ₁ *y ₁ . The number of turns of the inner sparsely wound coil is N2, the outer dimension is x ₂ *y ₂ , and the turn spacing is d. The distance between the outer densely wound coil and the inner sparsely wound coil is Δd.

The outer dense inner sparse coil structure is based on the size requirements of the coil. The outer coil adopts a dense winding method to ensure that the coil self-inductance and mutual inductance meet the most basic power transmission requirements. The inner coil adopts a sparse winding method to make the magnetic field distribution more uniform. Integrated lift coil anti-offset characteristics. It is worth mentioning that, compared with the traditional densely wound coil, the outer dense inner sparse coil can reduce the wire consumption and cost of the coil, reduce the coil self-inductance and internal resistance, thereby effectively reducing the voltage at both ends of the coil and coil loss. In addition, similar coils with sparse outer and inner dense coils have the characteristics of low self-inductance and mutual inductance. When the power transmission of the same power level is realized, the coil current will be increased, resulting in more losses and reducing the transmission efficiency of the system. Therefore, this embodiment chooses Outer dense inner sparse winding method.

Grouped series-wound coils are similar to outer-dense inner-sparse coils. Both inner and outer coils are combined in series. The outer coils are all densely wound. The difference is that the former's inner coils adopt a densely wound winding method. Therefore, compared with the outer-dense inner-sparse coil, the grouped series-wound coil also has the problems of larger wire consumption, larger coil self-inductance and internal resistance, and higher cost.

In order to improve the lateral anti-deflection ability of the coupling mechanism, the coil winding method of grouping and series winding is adopted. The grouping series winding coil is similar to the outer dense inner sparse coil, and the method of combining the inner and outer coils in series is adopted. The outer coils are all tightly wound, the difference is that the inner coil of the former adopts a densely wound winding method. Therefore, compared with the outer-dense inner-sparse coil, the grouped series-wound coil also has the problems of larger wire consumption, larger coil self-inductance and internal resistance, and higher cost. The following content will compare and analyze the coupling characteristics and anti-offset characteristics of these different winding methods.

In order to analyze the coupling characteristics and anti-deviation characteristics in the horizontal direction of the four winding methods (close winding, sparse winding, grouped series winding, and outer dense inner sparse), 3D simulation models were established for simulation analysis by Mxawell simulation software. In order to ensure that the obtained data are comparable, the various parameters of the coil are set as follows:

(1) The Maxwell simulation conditions and solver settings are the same;

(2) The outer diameters of the coils on the ground side and the vehicle side are the same, the total number of turns of the ground side coils is set to 10 turns, the number of turns of the outer coil is 6 turns, the number of turns of the inner coil is 4 turns, the initial winding points of the inner and outer coils are the same, and the turn spacing It is set to 5mm, and the number of turns of the vehicle-mounted coil is set to 20 turns;

(3) The coupling mechanism does not add a magnetic core, and the transmission distance d1 is kept consistent, which is set to 20cm.

There is not much difference in the uniformity of the magnetic field distribution of the two winding methods of grouped series winding and outer dense inner sparse winding. There are still advantages to sparse coils.

The simulation results of the four winding methods (close winding, sparse winding, grouped series winding, and outer dense inner sparse) are sorted out, and the trend diagram of the coil mutual inductance M and mutual inductance change rate of the coupling mechanism with the x-axis offset distance is obtained. Figure 3 (a), (b) shows, and the coil coupling coefficient k and coupling coefficient change rate with the x-axis offset distance change trend diagram shown in Figure 4 (a), (b).

According to Figure 3, when the coil of the coupling mechanism is offset in the x-axis direction, the mutual inductance value of the outer dense inner sparse coil is slightly smaller than that of the densely wound and grouped series wound coils, but the mutual inductance change rate is faster than that of the other three coils at any time. As the offset distance increases, the gap will narrow. The main reason is that the self-inductance of sparsely wound coils is smaller than that of densely wound coils. After the offset distance increases, the main factors affecting the mutual inductance and its rate of change become The magnetic field generated by the external coil.

According to Figure 4, it can be seen that the coupling coefficient of the dense outer coil and sparse inner coil is greater than that of the densely wound and grouped series wound coils at any offset distance, and only smaller than that of the sparsely wound coil, indicating that the magnetic field distribution of the outer dense inner sparse coil is more uniform , which also verifies the previous conclusion that its mutual inductance change rate is the lowest among the four winding methods. In addition, it can be seen from Figure 4 that the rate of change of the coupling coefficient is also at a lower level among the four methods. From the perspective of mutual inductance, coupling coefficient and its rate of change, the outer-dense inner-sparse coil has better resistance to x-axis offset.

It can also be obtained that when the coil is shifted on the y-axis, the variation trend of each coefficient is roughly the same as the previous x-axis trend, and the similarities will not be repeated here. The difference is that when the offset distance reaches the limit of 300mm, the coupling coefficients of the four methods will tend to the same value. The reason is that the system offset is too large, resulting in a sharp decrease in the coil mutual inductance, and the magnetic field received by the secondary coil is already very large. Faint. When the y-axis offset distance is less than 200mm, the coupling coefficient can always remain above 0.1. At this time, the mutual inductance and coupling coefficient of the outer dense inner sparse coil can be maintained at a relatively large value, and the mutual inductance and coupling coefficient change rate are maintained at a certain value. relatively small value. Therefore, it can be considered that the outer-dense inner-sparse coil has relatively better anti-offset ability in the y-axis.

From the above analysis, it can be seen that compared with the other three winding methods, the outer-dense inner-sparse coil has better horizontal anti-offset ability of the coupling mechanism.

For the size of the coil on the vehicle side, its size is limited to 40*40cm according to the actual working conditions. During the energy transmission process of the system, the on-board coil acts as the energy receiving end when charging forward, and becomes the energy transmitting end when discharging in the reverse direction. According to the size requirements, the system becomes a structure of "small transmitting and large receiving" at this time.

The design of the coil on the vehicle side mainly considers the characteristics of the reverse transmission due to the small size of the transmitter, which reduces the transmission performance. If the same winding method as the large-size ground terminal is used, the self-inductance and mutual inductance of the coil will be reduced. Under the premise of ensuring the same power level of transmission, the coil excitation current _Ip and voltage _Uin of the vehicle terminal will increase, thereby increasing the system loss. Ultimately, the transmission efficiency is greatly reduced. Therefore, it is necessary to select an appropriate coil structure to improve the reverse transmission performance of the system.

According to the Neumann formula, the mutual inductance between the two coils is:

M is the mutual inductance, N _Tx , N _Rx , l _Tx , l _Rx are the number of turns of the primary and secondary coils and the length of each coil, Δx is the relative distance between the two coils when they are offset, and μ ₀ is the air permeability.

It can be seen that the mutual inductance is positively correlated with the number of turns and the length of the coil, and the number of turns and the length of the coil directly determine the self-inductance of the coil. Therefore, in order to ensure that the energy transmission can meet the corresponding performance requirements, it is necessary to ensure that the ground terminal and The self-inductance and mutual inductance of the coil on the vehicle side are large enough.

In order to improve the self-inductance and mutual inductance of the vehicle-mounted coil, common methods include enlarging the coil size and increasing the number of coil layers. Expanding the coil size in the horizontal direction is contradictory to the limited size of the vehicle-mounted coil in this embodiment, so enlarging the coil size is not suitable for this application. In the embodiment system, this embodiment chooses to increase the number of layers of coils on the vehicle side.

The size of the coil directly determines the self-inductance of the coil. The theoretical calculation formula of the inductance of the multi-layer planar spiral coil is as follows:

In the formula: r _s is the distance between the effective center of the coil and the geometric center, d ₂ is the effective width of the coil, h is the thickness of the multi-layer coil, d _o is the outer diameter of the coil, n is the number of coil layers; the unit of inductance is uH.

It can be seen from the above formula that the self-inductance of the coil is proportional to the square of the outer diameter of the coil and inversely proportional to the inner diameter of the coil. Obviously, compared with the inner diameter of the coil, the parameter of the outer diameter of the coil contributes more to the self-inductance value of the coil. Under the condition that the outer diameter of the coil is determined, even if the number of turns of the coil is set to be large (reducing the inner diameter), the self-inductance of the coil will increase. The effect is still limited. For the coil structure of "small transmitting and large receiving", the external size of the ground terminal is larger than that of the vehicle terminal coil, so in order to meet the requirements of coil self-inductance, the vehicle terminal coil of the bidirectional MC-WPT system in this embodiment adopts a double-layer densely wound coil. line mode.

As shown in FIG. 5 , the complete ground-side coupling structure includes a ground metal shielding plate, a ground magnetic core, and a ground-side coil arranged from bottom to top. The vehicle-side coupling structure includes a vehicle-side coil, a vehicle-mounted magnetic core, and a vehicle-mounted metal shielding plate arranged from the bottom up.

The primary and secondary coils of the bidirectional MC-WPT system have large leakage inductance, and in the case of limited coil size, it is difficult to meet the requirements of coil mutual inductance and coupling coefficient simply by increasing the number of coil turns. Adding a ferrite core can effectively narrow the magnetic field lines of the coil, optimize the magnetic field distribution, and greatly reduce the volume of the coupling mechanism and reduce the cost.

In the actual system, the use of the magnetic core needs to comprehensively consider the relevant parameters such as the magnetic permeability, magnetic saturation capacity and resistivity of the magnetic core, so as to reduce the volume and reduce the loss. In this embodiment, the magnetic permeability is 2400H/m PC40 core (manganese zinc ferrite). In order to increase the coupling coefficient and mutual inductance of the coupling mechanism as much as possible and improve the anti-offset capability of the system, the magnetic cores of the coupling mechanism are placed in a full-spread manner. Considering the large size of the ground and vehicle coils, it is unrealistic to use only one complete magnetic core. Therefore, several small square magnetic cores with a size of 10*10*0.5cm are spliced to form a magnetic core of the required size.

The energy of the MC-WPT system is transmitted through magnetic field coupling, and there is always magnetic flux leakage in the system. In order to reduce the interference caused by the magnetic flux leakage of the system to the instruments on the electric vehicle, it is necessary to shield the magnetic flux leakage of the system. Ferrite cores can also shield magnetic flux leakage to a certain extent while constricting the magnetic field, but the effect is limited. Generally, a ferrite core and a metal shielding layer are used to make the charging system achieve a nearly complete shielding effect. The magnetic core of the coupling mechanism is spliced by small pieces of magnetic cores, and there are gaps and there is still magnetic flux leakage. Therefore, it is used in A metal aluminum plate is added behind the magnetic core to shield the magnetic flux leakage of the system.

The study in this example is based on the static electric vehicle wireless charging system, so the longitudinal transmission distance of the system is considered to be fixed, and only the anti-deviation characteristics of the system when the coil is offset in the horizontal direction are considered. The size of the on-board side of the coupling mechanism is limited by the size of the chassis of the electric vehicle, so the coil on the on-board side of this embodiment is limited to be wound in a square area of 40*40cm. Considering that the actual distance between the car chassis and the ground is 15-20cm, the transmission distance of the system is set at d1=20cm. In order to improve the anti-offset capability of the electric vehicle wireless charging system in the horizontal direction, the size of the system ground end is usually designed to be larger than that of the vehicle end. The size of the system ground end coil in this embodiment is limited to 65*50cm.

Under the premise that the outer diameter of the coil is limited, the focus of the design and optimization of the outer-dense inner-sparse coil is to obtain the optimal parameters of the number of turns N1 and N2 of the outer and inner coils, the spacing d of the inner coils, and the spacing Δd between the inner and outer coils. good relationship. The key to the design and optimization of the double-layer close-wound coil lies in the optimization of the number of turns Ns of the coil. The following is a flow chart of a parameter design method for the magnetic coupling mechanism of the bidirectional MC-WPT system as shown in Figure 6. By designing and optimizing the coil parameters, the optimal coil structure is finally obtained. The specific steps are:

S1. Determine the minimum mutual inductance Mmin and minimum coupling coefficient Kmin of the magnetic coupling mechanism according to the performance requirements;

S2. Parametrically scan the number of turns of the vehicle-mounted coil to determine the optimal number of turns;

S3. Determine the external dimensions x ₁ *y ₁ of the external densely wound coil according to actual needs;

S4. Build a 3D simulation model, perform parametric scanning on the number of turns N1 of the external densely wound coil, obtain the mutual inductance of the magnetic coupling mechanism, and obtain the value range of the number of turns N1 when the mutual inductance of the magnetic coupling mechanism is greater than Mmin;

S5. Determine the external dimension x ₂ *y ₂ of the internal sparsely wound coil according to actual needs;

S6. Parametrically scan the number of turns N2 of the inner sparsely wound coil and the distance between the outer densely wound coil and the inner sparsely wound coil as Δd to obtain the coupling coefficient of the magnetic coupling mechanism respectively, and obtain the mutual inductance of the magnetic coupling mechanism greater than Mmin The value range of the number of turns N2 and Δd at the time;

S7. Parametrically scan the turn spacing d of the internal sparsely wound coil to obtain the mutual inductance and coupling coefficient of the magnetic coupling mechanism, and determine the optimal turn spacing d according to the mutual inductance and coupling coefficient;

S8. Determine a set of values within the respective value ranges of N1, N2, and Δd, and obtain the system anti-offset characteristics and transmission characteristics of the set of values and the optimal turn spacing;

S9. Judging whether the anti-offset characteristics and transmission characteristics of the system meet the design requirements. If yes, determine the values of N1, N2, Δd and d at this time as the final design parameter values. Otherwise, return to step S8.

According to the previous parameter design method, the optimal number of turns of the outer densely wound coil is 10 turns, the number of turns of the inner sparsely wound coil is 5 turns, and the optimal turn spacing is 0.8cm. The vehicle-side coil adopts a double-layer close-wound winding method. The number of turns of the coil near the ferrite core is 12 turns, and the other layer is 10 turns.

According to the requirements of practical applications, the maximum horizontal offset of the coupling mechanism is 8 cm on the x-axis and 12 cm on the y-axis. In order to measure the anti-offset ability of the designed coupling mechanism, under the extreme offset condition, the coil mutual inductance and coupling coefficient change rate of the other three coil structures and the coil structure in this paper were compared and analyzed. The comparison results are shown in Figure 7 below. It can be seen that the coil winding method adopted in this embodiment is at the minimum value in both the rate of change of the mutual inductance and the rate of change of the coupling coefficient, which is about 12% lower than that of the sparse winding method with the largest rate of change.

In order to improve the anti-offset characteristics of the two-way MC-WPT system of electric vehicles, this embodiment first gives the particularity of the coupling mechanism of the two-way MC-WPT system, and then analyzes different coil modes (unipolar , bipolar, close-wound and sparse-wound) anti-offset characteristics, and it is concluded that the proper selection of two coil structures, densely wound and sparsely wound, can improve the anti-offset ability of the coil. Then, considering the mutual inductance, coupling coefficient and its rate of change comprehensively, the coils on the ground side and the vehicle side are respectively densely wound on the outside and sparsely wound on the inside, and the corresponding parameter design and optimization schemes are given. Compared with other coil structures, the designed coil reduces the change rate by at most 12% while ensuring sufficient mutual inductance, so that the system has better anti-offset capability.

It should be pointed out that the two-way MC-WPT system applied to the magnetic coupling mechanism of this embodiment is shown in Figure 8, which includes a ground terminal and a vehicle terminal, and the ground terminal includes a sequentially connected ground terminal DC power supply and a ground terminal full-bridge conversion module. , ground-side resonant network, ground-side coupling structure, and the vehicle-mounted terminal includes sequentially connected vehicle-side coupling structure, vehicle-side resonant network, vehicle-side full-bridge conversion module, filter circuit and vehicle-side battery.

Both the ground-side full-bridge conversion module and the vehicle-side full-bridge conversion module include a full-bridge inverter and four power diodes connected in antiparallel with four switching tubes of the full-bridge inverter.

The overall structure of the MC-WPT system using the SS topology is relatively simple, but the system has limited anti-offset capability, and there is an overcurrent problem when the secondary side is running without load, which will reduce the safety and reliability of the system, so the application scenarios are subject to many restrictions. The LCL-LCL topology has advantages in anti-offset capability and no-load conditions, but compared with the SS topology, the transmission power is reduced, and it is not suitable for high-power wireless energy transfer systems. The LCC-LCC topology has the advantages of the LCL-LCL topology in special working conditions, and there is no problem of limited transmission power. Therefore, the resonant topology selected in this embodiment is the LCC-LCC topology, that is, the resonant network at the ground end and the resonant network at the vehicle end both use the LCC resonant network to form the LCC-LCC resonant topology.

In addition, the LCC-LCC resonant topology satisfies the conditions:

Among them, w represents the operating angular frequency of the system, L _r1 , C _p , and C _r1 represent the parameter values corresponding to the inductance, series capacitance, and parallel capacitance in the LCC resonant network at the ground end, respectively, and L _r2 , Cs, and C _r2 represent the LCC at the vehicle end, respectively. The parameter values corresponding to the inductance, series capacitance and shunt capacitance in the resonant network. The MC-WPT system using the LCC-LCC resonant topology can ensure that the system works in a completely resonant condition when the four conditions shown below are met. For the MC-WPT system using LCC-LCC resonant topology, if the position of the primary and secondary coils of the system remains unchanged and the parameters of the resonant network are determined, the amplitude of the output current _I2 is only determined by the input voltage U _in , so the LCC-LCC resonant The topology has the characteristic of maintaining constant current output when the system is input with constant voltage.

As shown in Figure 9, when the ground terminal performs energy reverse transmission to the vehicle terminal, the ground terminal full-bridge conversion module works in the inverter state, converting the ground terminal DC power into high-frequency AC power, while the vehicle terminal full-bridge conversion module works In the rectification state, the four switching tubes of the full-bridge conversion module on the vehicle end are turned off, and the four power diodes connected in reverse parallel form the vehicle-mounted full-bridge rectifier circuit, which rectifies the high-frequency alternating current induced by the coil on the vehicle end into direct current. Finally, it is supplied to the vehicle-mounted battery for charging;

As shown in Figure 10, when the vehicle end transmits energy forward to the ground end, the vehicle end full-bridge conversion module works in the inverter state, converting the DC power provided by the vehicle end battery into high-frequency AC power, while the ground end full bridge The conversion module works in the rectification state, that is, the four switching tubes of the ground-side full-bridge conversion module are turned off, and the ground-side full-bridge rectifier circuit is composed of four corresponding antiparallel power diodes, and the high-frequency AC induced by the ground-side coil It is rectified into DC, and finally supplied to the DC power supply of the ground terminal.

For the two-way MC-WPT system of electric vehicles, when the energy is transmitted in the forward direction, when the energy is transmitted from the grid side to the load wirelessly, the load of different electric vehicles is different, and the load change or the fluctuation of the mutual inductance of the coil will cause the output characteristics of the system to be unstable. When the energy is reversely transmitted, although there is no load change problem on the grid side, mutual inductance fluctuations still exist, and the system output is also unstable. In order to solve the above problems, this paper controls the current on the energy receiving side to achieve efficient and stable output of the MC-WPT system.

According to the above analysis results, the LCC-LCC resonant topology theory selected in the present invention should have the characteristics of constant current output, but the premise is to ensure that the mutual inductance value is stable and the resonance network is completely resonant. Considering the fact that the mutual inductance of the coupling mechanism coil offsets At the same time, there is a difference between the matching inductance and capacitance in the actual system and the theoretical calculation value (the resonant network is not in a completely resonant state), and it is usually impossible to maintain a constant current output. Therefore, in addition to improving the anti-offset characteristics of the coupling mechanism, it is still A corresponding control strategy is required to achieve constant current output.

The present invention selects phase shift control as the constant current control scheme of the bidirectional MC-WPT system. Although the adjustment range of the phase shift control is relatively narrow, the improvement of the anti-offset capability of the system and the realization of constant current output have been considered in the selection of the system resonance topology and the design and optimization of the coupling mechanism, so the adjustment of the phase shift angle can be reduced. This makes up for the narrow adjustment range of the phase shift control to a certain extent.

The schematic diagram of the phase-shifting circuit of the full-bridge inverter is shown in Figure 11. In the phase-shifting control mode, the switches S1 and S2 are still in complementary conduction, with S1 and S2 as the reference bridge arm, and S3 and S4 as the lagging arm. The phase difference is 180°-θ, and θ is called the phase shift angle.

The switching tube driving and inverter output voltage and current waveforms are shown in Figure 12. In one cycle, there are a total of 4 working modes. In working mode ①, the switching tubes S2 and S3 work, and the inverter output voltage is -Udc; In the working mode ②, the current is not commutated, the switch tube S2 and the power diode D4 work to ensure the continuous flow of the circuit, and the output voltage of the inverter is 0; in the working mode ③, the switch tubes S1 and S4 work, and the output voltage of the inverter Udc, in working mode ④, the current does not commutate, at this time the switch tube S1 and power diode D2 work to ensure the continuous flow of the circuit, and the output voltage of the inverter is 0. It can be seen that by adjusting the value of the phase shift angle θ (0<θ≤180°), the effective value of the output voltage of the full-bridge inverter circuit can be changed, thereby controlling the output power of the entire system.

The present invention selects phase-shift control as the constant current control scheme of the bidirectional MC-WPT system. Referring to FIG. 8 , it is a structural block diagram of the phase-shift control of the bidirectional MC-WPT system. In order to facilitate the analysis and experiment, when the energy is transmitted in the forward direction, the AC input of the power grid to the circuit before the inverter input is equivalent to the DC source of the ground terminal, and the load is set as a pure resistive load. The circuit before the inverter input is equivalent to a DC source on the vehicle side, and the load on the ground side is set to a purely resistive load at this time.

Therefore, in order to further realize bidirectional constant current output, as shown in Figure 8, the two-way MC-WPT system provided by this embodiment includes a ground terminal controller connected to the ground terminal full-bridge conversion module, and a ground terminal controller connected to the ground terminal. The ground terminal communication module and the ground terminal current detector, the vehicle terminal also includes the vehicle terminal controller connected to the vehicle terminal full bridge conversion module, and the vehicle terminal communication module and vehicle terminal current detector connected to the vehicle terminal controller.

When the ground terminal is used as the energy transmitter to transmit energy forward to the vehicle terminal, the vehicle terminal current detector obtains the current I ₂ flowing through the vehicle terminal battery, and is sent by the vehicle terminal communication module to the ground terminal communication module, and the ground terminal controller Comparing I ₂ with the ground reference current value _Iset , the phase shift angle θ is calculated, so as to generate the PWM drive signal of the switching tube with the phase shift angle θ, which acts on the full bridge conversion module at the ground end.

Based on the same principle as the energy forward transmission process, the vehicle-mounted terminal acts as an energy transmitter to perform reverse energy transmission to the ground terminal. That is to say, when the vehicle terminal is used as the energy transmitting terminal to transmit energy forward to the ground terminal, the ground terminal current detector obtains the current I ₁ flowing through the ground terminal DC power supply, and sends it to the vehicle terminal communication module by the ground terminal communication module , the vehicle-mounted controller compares I ₁ with the vehicle-mounted reference current value _I'set , and calculates the phase shift angle θ', thereby generating a switch tube PWM drive signal with a phase-shifted angle of θ', which acts on the vehicle-side full-bridge conversion module .

For the bidirectional MC-WPT system with constant current output, the controlled quantity changes with the transmission direction. When the energy is transmitted in the forward direction, the controlled quantity is the output current I ₂ of the vehicle terminal. At this time, the phase shift control is the ground terminal controller, and the controlled quantity becomes the ground terminal current I ₁ during the reverse transmission. The vehicle-side controller plays the role of phase-shift control. Therefore, if the system wants to realize bidirectional constant current transmission, a communication module needs to be added to send the collected currents I ₁ and I ₂ to corresponding controllers respectively.

Based on the introduction and analysis of the bidirectional MC-WPT system resonance parameter design and constant current control strategy, the circuit simulation model is built through the Matlab/Simulink simulation software platform. The Matlab software version used in this paper is Matlab 2019a. The circuit part of the simulation model is completely consistent with the block diagram of the constant current control strategy system shown in Figure 8.

The simulation parameter settings of the system are shown in Table 1 below.

Table 1 Bidirectional MC-WPT system simulation parameters

Figure 13 is the simulation waveform when the system performs load switching during forward energy transmission, where (a) and (b) in Figure 13 are enlarged diagrams of inverter output voltage waveforms before and after load switching, and (c) is the phase shift angle at The change trend graph before and after load switching, (d) is the DC output current waveform at both ends of the load. The current reference value of the system is 8.3A, and the load value is switched from 43Ω to 21Ω at 0.05s to simulate the process of switching the output power from full power operation to 50% power operation in the actual system. It can be seen that when the DC input voltage is constant, as the load resistance is switched from 43Ω to 21Ω, the effective value of the inverter voltage and the output voltage will decrease through phase shift control, while the output current can remain constant before and after the load switching.

The working mechanism is: when the control link is not added, switching the load (switching from 43Ω to 21Ω) will cause the equivalent impedance of the secondary side to drop suddenly, but the current on the primary side has not changed in the future, and the induced voltage on the secondary side remains unchanged, which will cause the load to flow through the load. sudden increase in current. After the primary side is added to the control, the load switching will cause the load current to rise suddenly, and there will be a difference with the reference current value. The phase-shift controller will reduce the phase-shift angle, and the effective value of the inverter output voltage will decrease. The induced voltage and current will also be reduced accordingly, and finally the load current will be stabilized at the set value, achieving the effect of constant output of the system. It is worth noting that there will be a sharp increase in the load current before and after the load switching, because the load switching process is too short for the LCC-LCC high-order system, the system has a certain reaction time, and the phase shift angle decreases. time delay, resulting in a surge in load current.

Figure 14 is the simulation waveform when the mutual inductance of the system changes during forward energy transmission, where (a) and (b) are the enlarged diagrams of the inverter output voltage waveform before and after the mutual inductance change, and (c) is the phase shift angle before and after load switching Variation trend graph, (d) is the DC output current waveform at both ends of the load. It can be seen from the waveform diagram that when the mutual inductance changes, the phase shift angle increases, the effective value of the primary coil current increases, and the output current remains unchanged.

The current reference value of the system is still set to 8.3A, and the mutual inductance of the coil changes at 0.04s (from M=38uH to M=30uH), which is used to simulate the actual working condition that the coil shifts and the mutual inductance changes. In the simulation, the mutual inductance The values before and after the change are the mutual inductance of 38uH when the coil is facing right and the mutual inductance of 30uH when the coil is under the extreme offset. The working mechanism is: keep the DC input voltage constant, reduce the mutual inductance of the system coil, reduce the induced voltage of the secondary side, and the load does not change, which directly leads to a decrease in the output current flowing through the load. At this time, there is a difference between the load current and the reference value. Under the action of the primary-side phase-shift controller, the phase-shift angle will be increased to increase the inverter output voltage, thereby increasing the primary-side coil current to offset the trend of decreasing output current. , and finally make the system output current constant.

Figure 15 shows the system simulation results under the working state of mutual inductance change during energy reverse transmission, including the enlarged diagram of the inverter output voltage waveform before and after the mutual inductance change, the diagram of the phase shift angle change, and the DC output current waveform at both ends of the load. It can be seen that when the energy is transmitted in the reverse direction, the simulation system can also keep the current at the output terminal constant by adjusting the phase shift angle. The basic principle of the control is exactly the same as that of the forward transmission, and will not be repeated here.

To sum up, whether it is energy forward transmission or reverse transmission, the simulation model can ensure that the output current remains constant during load switching and mutual inductance changes, and the simulation results correspond to the theoretical analysis one by one.

Experimental verification is carried out below.

First of all, the basic performance indicators of the two-way MC-WPT system designed in this paper are:

(1) Energy wireless transmission distance: 20cm;

(2) Coupling mechanism coil X/Y axis offset: 8cm/12cm;

(3) Operating frequency of switching devices: 85kHz;

(4) Full load input and output voltage: 360VDC;

(5) Forward and reverse full load power: 3000W;

(6) Transmission efficiency of the whole machine: over 85%.

In this paper, Litz wire is used to wind the primary and secondary coils. Litz wire is made of multiple thin wires twisted, which can effectively reduce the influence of skin effect on system parameters and efficiency. The diameter of Litz wire is selected according to the maximum current passing through the coil, and the maximum coil current I _coil can be obtained:

A 30% margin is considered during design, so the design reference value of the coil current is 22.1A. According to the Litz wire selection specification, the final selection specification is a Litz wire with a wire diameter of about 4.5mm, its wire core is 0.1mm*1000, and the withstand voltage level is 3kV.

The primary coil adopts the outer dense inner sparse winding method, the outer coil has 10 turns, the inner coil has 5 turns and the turn spacing is 0.8cm; the secondary coil is double-layer densely wound, and the number of turns is 12+10 turns. The actual measured values of the relevant parameters of the coupling mechanism are shown in Table 2 below, where the direct alignment of the coils means that the offsets of the primary and secondary coils on the x-axis and the y-axis are both 0, and the limit offset of the coils means that the offsets of the coils on the x-axis are 8cm, the y-axis offset is 12cm.

Table 2 The measured values of the parameters of the coupling mechanism

From the measured value of the coil, it can be found that the mutual inductance of the system changes from 38.5uH to 30uH, and the change rate is 22%, which is close to the simulation results in Chapter 3. In order to further verify the anti-offset characteristics of the system, the mutual inductance of the coils is measured when they are offset to different degrees in the x-axis and y-axis directions. The measured results are shown in Figure 16. It can be seen that there is a certain gap between the measured mutual inductance and the simulation results. The reason is that the actual winding density and turn spacing of the coils cannot be as accurate as the simulation, but the overall change trend and rate of change of the mutual inductance are the same as the simulation, indicating that the coils wound It has good anti-deviation ability in the horizontal direction, which can improve the redundancy of the closed-loop control of the system.

The main control module of the ground terminal and the vehicle terminal adopts the DSP control chip TMS320F28335 as the main controller, which is mainly responsible for control, analog-to-digital conversion, drive signal output, communication and other work. The workflow of the constant current output phase-shift control method of the bidirectional MC-WPT system is shown in Figure 17 below. When the system is started, the current detection part is used to detect the current to be controlled, the data is transmitted to the corresponding main control chip through communication and A/D conversion is completed, and then the actual sampling value is compared with the preset current value to obtain the difference, after The PI algorithm obtains the corresponding phase shift angle θ, and finally the main control chip outputs a PWM drive signal with a phase shift angle of θ. Specifically, corresponding to FIG. 8, the constant current output phase-shift control method includes the ground phase-shift control step of performing forward constant current output from the ground terminal to the vehicle terminal, and performing reverse constant current output from the vehicle terminal to the ground terminal. The vehicle-mounted phase shift control steps, the ground phase shift control steps include:

A1. Obtain the current I ₂ flowing through the vehicle-mounted battery;

A2. Comparing the current I ₂ with the ground reference current value I _set to obtain the difference ΔI;

A3. Determine whether ΔI is 0, and if so, do nothing; otherwise, calculate the phase shift angle θ corresponding to ΔI according to the PI algorithm and output a PWM driving signal with a phase shift angle θ to act on the full-bridge conversion module at the ground end.

The principle of the vehicle-mounted phase-shift control steps is the same as that of the ground-based phase-shift control steps. Specifically, the steps include:

B1. Obtain the current I ₁ flowing through the ground terminal battery;

B2. Comparing the current _I1 with the vehicle reference current value _I'set to obtain the difference ΔI';

B3. Judging whether ΔI' is 0, if so, do nothing, if not, calculate the phase shift angle θ' corresponding to ΔI' according to the PI algorithm, and output a PWM drive signal with a phase shift angle θ' to act on the full-bridge conversion module at the vehicle end .

The PI parameter setting adopts the trial and error method commonly used in engineering, that is, first set the initial parameter values Kp and Ti in the DSP program, and modify the PI one by one by observing the dynamic performance (power rise time, overshoot, etc.) in a low power state. parameter.

The driving circuit of the power device is shown in Figure 18. The driving circuit is composed of two-level driving. The dual-channel driving chip UCC27524 is used as the first-level driving. The main function is to upgrade the 3.3V driving pulse signals PWM1 and PWM2 output by the controller to 12V. Drive signals PWMH, PWML; the second level drive is composed of optocoupler isolation drive chip HCPL-3120 and its peripheral circuits, the output is connected to the totem pole drive structure, and the final output +18V/-3V is used to control the power MOSFET on and off Gate drive voltages G_S and H_S.

The power switch tube is very important for the full-bridge inverter circuit. According to the working conditions of the actual system, the switch tube should have sufficient voltage and current stress. MOSFET is a commonly used power switching device in MC-WPT system. According to its material, it can be divided into silicon (Si) MOSFET, silicon carbide (SiC) MOSFET, etc. According to some key characteristics of SiC MOSFET, considering sufficient voltage and current margin, Infineon FF23MR12W1M1 is selected as the inverter power device in this paper. Two power MOSFETs are directly connected in series to form a bridge arm. The basic operating parameters are shown in the table 3.

Table 3 Basic working parameters of FF23MR12W1M1

For the diode in reverse series with the power MOSFET in this system, its function is to realize rectification when the corresponding power MOSFET is in the off state. Considering the design margin of voltage and current in the system, the model of the reverse series diode used in this paper is G3S06010J. Its maximum reverse withstand voltage is 600V, and its maximum forward conduction current is 30A, which meets the design requirements of this system.

The current detection chip on the power circuit is the ACS712ELCTR-20A-T chip based on the Hall effect. Its maximum detection current is 20A, and it has a high-precision (detection sensitivity of 100mV/A, maximum sampling error of 1.5%) sampling output. Voltage, the measured value is the current value flowing through the main power circuit. The current is converted into a corresponding voltage signal I_ADC by the current detection chip, and the signal will be sent to the DSP for control and protection processing.

In order to verify that the system designed in this paper can achieve bidirectional energy transmission and has anti-offset capability, the experimental results are analyzed from two aspects of energy forward and reverse transmission.

When the system energy is transmitted in the forward direction and the transmission distance is 20cm, the waveform diagram and the indications of the DC source and the electronic load when the coil is in direct alignment and extreme offset are shown in Figure 19. The waveform diagram shows the experimental waveforms of the system inverter output voltage U _ab , inverter output current I _ab , DC output voltage V _out and DC output current I _out when the coils are differently offset. It can be seen from the figure that the system can maintain a constant current output of 8.32A under the conditions of direct alignment and extreme offset of the coil. The reason for the gap with the set current of 8.35A is that there is a certain error in the system current detection. Comparing the waveforms of the two The inverter output voltage U _ab can be found that the phase shift angle of the coil alignment system is significantly smaller than that of the coil in the limit offset state, indicating that when the mutual inductance decreases, the controller at the ground end of the system realizes the control of the current by increasing the phase shift angle.

Fig. 20 shows the variation curves of the transmission power and efficiency of the system coils under the conditions of different offsets in the x-axis and y-axis directions during the forward constant current transmission process. It can be seen from the figure that the system power can be close to 3000W, and the fluctuation is very small under different offsets of the x and y axes; however, as the offset of the x and y axes increases, the system transmission efficiency will decrease, and the highest transmission efficiency can reach 89.7%, the designed system meets the corresponding performance index requirements. For the control effect of the control method used on the output power (output power fluctuation rate), this paper uses the difference rate ΔP between the actual output power P _o and the expected power P _e (P _e =3000W) as a measure of power fluctuation, ΔP=( P _o -P _e )/P _e , according to the measured output power, the maximum output power fluctuation rate when the energy is transmitted forward is

ΔP _max =(P _o-min -P _e )/P _e =1.16%

Fig. 21 is an experimental waveform diagram of the system output voltage V _out and the output current I _out when the system energy is transmitted in the forward direction and the load is switched. It can be seen that after the load changes, the system can maintain a constant output current through phase-shift control. Different load resistances (43Ω, 32Ω, and 21Ω) correspond to the system’s full load, 75% of rated power, and 50% of rated power. power.

Similar to the forward transmission experiment, as shown in Figure 22 below, when the energy is transmitted in the reverse direction and the transmission distance is 20cm, the waveform diagram and the DC source and electronic load indications are given when the coil is in direct alignment and extreme offset. The waveform diagrams show the experimental waveforms of the system inverter output voltage U _ab , inverter output current I _ab , DC output voltage Vout and DC output current I _out under different offset conditions. It can be seen from the figure that the system can maintain a constant current output of 8.32A under the conditions of direct alignment and extreme offset of the coil. Comparing the root mean square value of the inverter output voltage U _ab of the two, it can be concluded that it is similar to the forward transmission experiment. The conclusion is that when the mutual inductance of the system changes, the output current of the ground terminal can be kept constant by adjusting the phase shift angle of the vehicle-side controller. The only difference is that the controller and the controlled quantity are interchanged.

Fig. 23 shows the variation curves of the transmission power and efficiency of the system coils under the conditions of different offsets in the x-axis and y-axis directions during the reverse constant current transmission process. It can be seen from the figure that compared with the forward transmission, there is not much difference in the size and fluctuation of the transmission power; in terms of efficiency, the overall trend of the efficiency of the reverse transmission is the same as that of the forward transmission (decreases with the increase of the offset distance) , but the overall efficiency of reverse transmission is higher, the highest efficiency can reach more than 90.3%, even the lowest efficiency is 0.3 percentage points higher than the highest efficiency of forward transmission. The coil at the ground end is large, and the coil current required to transmit the same power is smaller, resulting in a reduction in the internal resistance loss of the coil and an increase in transmission efficiency. Similarly, according to the measured output power, the maximum output power fluctuation during energy reverse transmission can be obtained as

ΔP' _max = (P' _o-min - P _e )/P _e = 0.91%

Based on the experimental results and analysis of the forward and reverse energy transmission of the previous system, it can be concluded that the transmission power and efficiency of the system can meet the design requirements under the two energy transmission modes, and the maximum output power fluctuation rate is about 1.0%. , in the working condition that the coil offset causes the mutual inductance to change and the working condition of the forward transmission load switching, the constant output of the current can be realized through the phase shift control.

In summary, the present invention builds a two-way MC-WPT system with an input voltage of 360V, a constant output current of 8.3A, and a full load power of 3000W. The actual measurement results show that when the coil of the coupling mechanism is offset, the measured values of the coil parameters are basically consistent with the simulation results, indicating that the wound coil has a good ability to resist offset in the horizontal direction, which can improve the redundancy of the closed-loop control of the system ; The experimental results verify that the transmission power and efficiency of the system can meet the design requirements under the different working conditions of the two energy transmission modes, and can achieve constant current output through phase shift control.

The above-mentioned embodiment is a preferred embodiment of the present invention, but the embodiment of the present invention is not limited by the above-mentioned embodiment, and any other changes, modifications, substitutions, combinations, Simplifications should be equivalent replacement methods, and all are included in the protection scope of the present invention.

Claims (8)

1. The two-way MC-WPT system is characterized in that: it includes a ground terminal and a vehicle terminal; The ground terminal includes a sequentially connected ground terminal DC power supply, a ground terminal full-bridge conversion module, a ground terminal resonant network, and a ground terminal coupling structure, and also includes a ground terminal controller connected to the ground terminal full bridge conversion module, and a ground terminal controller connected to the ground terminal. The ground-side communication module and the ground-side current detector of the ground-side controller; The vehicle-mounted terminal includes sequentially connecting the vehicle-mounted terminal coupling structure, the vehicle-mounted terminal resonant network, the vehicle-mounted full-bridge conversion module, the filter circuit and the vehicle-mounted battery, and also includes a vehicle-mounted controller connected to the vehicle-mounted full-bridge conversion module, And a vehicle-mounted communication module and a vehicle-mounted current detector connected to the vehicle-mounted controller; Both the ground-side full-bridge conversion module and the vehicle-mounted full-bridge conversion module include a full-bridge inverter and four power diodes connected in antiparallel with the four switching tubes of the full-bridge inverter; When the ground terminal is used as the energy transmitting terminal to transmit energy forward to the vehicle terminal, the ground terminal full-bridge conversion module works in the inverter state to convert the ground terminal DC power supply into high-frequency AC power, and the The vehicle-mounted full-bridge conversion module is working in a rectification state, the four switch tubes of the vehicle-mounted full-bridge conversion module are turned off, and the vehicle-mounted full-bridge rectifier circuit is formed by corresponding four power diodes connected in reverse parallel. The high-frequency alternating current induced by the terminal coil is rectified into direct current, and finally supplied to the vehicle-mounted battery for charging; the vehicle-mounted current detector obtains the current I ₂ flowing through the vehicle-mounted battery, and is sent by the vehicle-mounted communication module To the ground end communication module, the ground end controller _compares I with the ground reference current value _Iset , calculates the phase shift angle θ, thereby generating a switching tube PWM drive signal with a phase shift angle of θ, acting on The ground end full-bridge conversion module; Based on the same principle as the energy forward transmission process, the vehicle-mounted terminal acts as an energy transmitting terminal to perform energy reverse transmission to the ground terminal; The ground-side coupling structure includes a ground-side coil, and the vehicle-side coupling structure includes a vehicle-side coil; The ground-side coil is formed in series by an external close-wound coil wound by a close-wound method and an internal sparse-wound coil wound in the same direction as the external close-wound coil by a sparse-wound method; Made of double-layer dense winding; The number of turns N1 and N2 of the outer and inner coils of the ground-side coil, the spacing d between the inner coils and the spacing Δd between the inner and outer coils are determined by the following steps: S1. Determine the minimum mutual inductance Mmin and minimum coupling coefficient Kmin of the magnetic coupling mechanism according to the performance requirements; S2. Parametrically scan the number of turns of the vehicle-mounted coil to determine the optimal number of turns; S3. Determine the external dimensions x ₁ *y ₁ of the external densely wound coil according to actual needs; S4. Build a 3D simulation model, perform parametric scanning on the number of turns N1 of the external densely wound coil, obtain the mutual inductance of the magnetic coupling mechanism, and obtain the value range of the number of turns N1 when the mutual inductance of the magnetic coupling mechanism is greater than Mmin; S5. Determine the external dimension x ₂ *y ₂ of the internal sparsely wound coil according to actual needs; S6, the number of turns N2 of the inner sparsely wound coil and the interval distance between the outer densely wound coil and the inner sparsely wound coil are parametrically scanned to obtain the coupling coefficient of the magnetic coupling mechanism respectively, and the coupling coefficient of the magnetic coupling mechanism is greater than The value range of the number of turns N2 and Δd at Kmin; S7. Parametrically scan the turn spacing d of the internal sparsely wound coil to obtain the mutual inductance and coupling coefficient of the magnetic coupling mechanism, and determine the optimal turn spacing d according to the mutual inductance and coupling coefficient; S8. Determine a set of values within the respective value ranges of N1, N2, and Δd, and obtain the system anti-offset characteristics and transmission characteristics of the set of values and the optimal turn spacing; S9. Judging whether the anti-offset characteristics and transmission characteristics of the system meet the design requirements. If yes, determine the values of N1, N2, Δd and d at this time as the final design parameter values. Otherwise, return to step S8. 2. _The two-way MC-WPT system according to claim 1, characterized in that: the ground-side controller compares I with the ground reference current value I _set to obtain a difference ΔI, and calculates the corresponding value of ΔI according to the PI algorithm The phase shift angle θ. 3. The two-way MC-WPT system according to claim 1, characterized in that: both the ground-side resonant network and the vehicle-side resonant network use an LCC resonant network to form an LCC-LCC resonant topology. 4. The two-way MC-WPT system according to claim 3, wherein the LCC-LCC resonant topology satisfies the conditions: Among them, w represents the operating angular frequency of the system, L _r1 , C _p , and C _r1 respectively represent the parameter values corresponding to the inductance, series capacitance, and parallel capacitance in the LCC resonant network at the ground end, and L _r2 , Cs, and C _r2 represent the Describe the parameter values corresponding to the inductance, series capacitance and parallel capacitance in the LCC resonant network on the vehicle side. 5. The two-way MC-WPT system according to claim 1, characterized in that: the ground-side controller is provided with a drive circuit, the drive circuit is composed of two-stage drive, and the dual-channel drive chip is used as the first-stage drive, It is used to upgrade the 3.3V driving pulse signals PWM1 and PWM2 output by the ground-end controller to 12V driving signals PWMH and PWML; the second-level drive is composed of an optocoupler isolation drive chip and its peripheral circuits, and the output is connected to the totem pole drive structure, and finally output +18V/-3V gate drive voltages G_S and H_S for controlling the turn-on and turn-off of the power MOSFET in the ground-side full-bridge conversion module; The setting of the vehicle-mounted controller is the same as that of the ground controller. 6. The two-way MC-WPT system according to claim 5, characterized in that: The ground-side coil and the ground-side coil are wound with Litz wire of the same specification. 7. The two-way MC-WPT system according to claim 5, characterized in that: The ground-side coupling structure includes a ground metal shielding plate, a ground magnetic core, and the ground-side coil arranged from bottom to top; The vehicle-mounted coupling structure includes the vehicle-mounted coil, the vehicle-mounted magnetic core, and the vehicle-mounted metal shielding plate arranged from bottom to top; Both the ground magnetic core and the vehicle magnetic core are spliced by a plurality of square magnetic cores. 8. The constant current output phase-shifting control method of the two-way MC-WPT system according to any one of claims 1 to 7, characterized in that it includes a ground terminal for positive constant current output from the ground terminal to the vehicle terminal Phase-shifting control step, and the vehicle-mounted phase-shifting control step of performing reverse constant current output from the vehicle-mounted terminal to the ground terminal, the ground phase-shifting control step includes: A1. Acquiring the current I ₂ flowing through the vehicle-mounted battery; A2. Comparing the current I ₂ with the ground reference current value I _set to obtain the difference ΔI; A3, judge whether ΔI is 0, if so, do nothing, if not, calculate the phase shift angle θ corresponding to ΔI according to the PI algorithm, and output a PWM driving signal with a phase shift angle θ to act on the full-bridge conversion module at the ground end; The principle of the vehicle-mounted phase shift control step is the same as that of the ground phase shift control step.