CN115276256B - Bidirectional MC-WPT system and constant-current output phase-shifting control method thereof - Google Patents

Bidirectional MC-WPT system and constant-current output phase-shifting control method thereof Download PDF

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Publication number
CN115276256B
CN115276256B CN202210841334.6A CN202210841334A CN115276256B CN 115276256 B CN115276256 B CN 115276256B CN 202210841334 A CN202210841334 A CN 202210841334A CN 115276256 B CN115276256 B CN 115276256B
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vehicle
ground
coil
current
terminal
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CN115276256A (en
Inventor
龚文兰
王智慧
唐春森
肖静
吴晓锐
韩帅
吴宁
莫宇鸿
陈绍南
陈卫东
郭敏
郭小璇
左志平
李小飞
周浩
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Chongqing University
Electric Power Research Institute of Guangxi Power Grid Co Ltd
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Chongqing University
Electric Power Research Institute of Guangxi Power Grid Co Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/10—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling
    • H02J50/12—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling of the resonant type
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L53/00—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles
    • B60L53/10—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles characterised by the energy transfer between the charging station and the vehicle
    • B60L53/12—Inductive energy transfer
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L55/00—Arrangements for supplying energy stored within a vehicle to a power network, i.e. vehicle-to-grid [V2G] arrangements
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
    • H01F5/00—Coils
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/28—Arrangements for balancing of the load in a network by storage of energy
    • H02J3/32—Arrangements for balancing of the load in a network by storage of energy using batteries with converting means
    • H02J3/322—Arrangements for balancing of the load in a network by storage of energy using batteries with converting means the battery being on-board an electric or hybrid vehicle, e.g. vehicle to grid arrangements [V2G], power aggregation, use of the battery for network load balancing, coordinated or cooperative battery charging
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/005—Mechanical details of housing or structure aiming to accommodate the power transfer means, e.g. mechanical integration of coils, antennas or transducers into emitting or receiving devices
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/70—Energy storage systems for electromobility, e.g. batteries

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Transportation (AREA)
  • Mechanical Engineering (AREA)
  • Electric Propulsion And Braking For Vehicles (AREA)

Abstract

The invention relates to the technical field of magnetic coupling wireless power transmission, in particular to a bidirectional MC-WPT system and a constant current output phase-shifting control method thereof, wherein a ground end controller, a ground end communication module and a ground end current detector are arranged at a ground end, and a vehicle-mounted end controller, a vehicle-mounted end communication module and a vehicle-mounted end current detector are arranged at a vehicle-mounted end, so that when the ground end is used as an energy transmitting end to forward transmit energy to the vehicle-mounted end, the vehicle-mounted end current detector is used for acquiring current I flowing through a vehicle-mounted end battery 2 And the phase shift angle theta corresponding to the delta I is calculated by the ground end controller according to a PI algorithm, acts on the ground end full-bridge conversion module to carry out phase shift control, and constant current output of the vehicle-mounted end is realized. Otherwise, when the vehicle-mounted terminal carries out energy reverse transmission on the ground terminal, the process is similar to the process. Thereby realizing the bidirectional constant current output control of the bidirectional MC-WPT system.

Description

Bidirectional MC-WPT system and constant-current output phase-shifting control method thereof
Technical Field
The invention relates to the technical field of magnetic coupling wireless power transmission (MC-WPT), in particular to a bidirectional MC-WPT system and a constant current output phase shift control method thereof.
Background
The wireless power transmission (Wireless Power Transfer, WPT) technology refers to a technology for realizing power transmission between a power source and a load end by means of a space medium carrier instead of a traditional wire contact mode, wherein the space medium carrier comprises magnetic fields, electric fields, lasers, microwaves, ultrasonic waves and the like. The magnetic coupling wireless power transmission (Magnetic Coupling Wireless Power Transfer, MC-WPT) is a great research in the current WPT technology field, the MC-WPT technology takes a magnetic field as a transmission medium, and the wireless transmission of the electric energy between the primary side and the secondary side is realized through the mutual conversion of the high-frequency alternating magnetic field and the electric energy. The technology has been brought to the attention and research of students at home and abroad since birth, the application field is more and more extensive, and the MC-WPT is successfully applied to various fields such as consumer electronics, biomedical, electric automobiles, underwater, special places and the like.
The total power supply amount at the power grid end is unchanged, while the power utilization end has a peak period and a valley period, the power utilization peak period is generally in the daytime, the power supply is possibly insufficient, the power supply is in the valley at night, and the power is wasted to a certain extent. The number of the electric vehicles is huge, the energy interaction between the electric vehicles and a power Grid is enhanced, namely the technology from the electric vehicles to the power Grid (V2G), the electric vehicles are used for charging at night in the electricity consumption low valley period, and the idle electric vehicles in the daytime peak period feed energy back to the power Grid end, so that peak clipping and valley filling are realized, the electric energy utilization rate is improved, and more balanced and effective utilization of energy is realized. If the MC-WPT technology is adopted in the charging and discharging process between the electric automobile and the power grid, the interaction capability of the electric automobile and the power grid can be improved undoubtedly, so that the whole system is more convenient, efficient, flexible and reliable, and the function of V2G can be exerted more intelligently and fully. Thus, there is an increasing need for research into two-way magnetically coupled wireless power transfer (two-way MC-WPT) technology.
The related research of the unidirectional MC-WPT technology is mature, and the bidirectional MC-WPT technology is widely applied to various fields, is developed from the unidirectional MC-WPT technology, can realize the interaction and networking of electric energy at a power supply and a load end, and has been widely focused and researched in recent years.
For the bidirectional MC-WPT system of the electric automobile, when energy is transmitted in the forward direction, and the energy is transmitted to a load in a wireless mode from a power grid side, the loads of different electric automobiles are different, and the output characteristic of the system is unstable due to load change or coil mutual inductance fluctuation. When energy is reversely transmitted, although the problem of load change does not exist on the power grid side, the mutual inductance fluctuation phenomenon still exists, and the system output is unstable.
For a bidirectional MC-WPT system, the design of the structure of the coupling mechanism plays a significant role in the power transmission and anti-offset capability of the whole system. At present, research results of a bidirectional MC-WPT system are rich, but the characteristics of a coupling mechanism and anti-offset characteristics and the like are still limited during forward and reverse transmission of the bidirectional MC-WPT system, and further improvement and optimization are still needed.
Disclosure of Invention
The invention provides a bidirectional MC-WPT system and a constant current output phase shift control method thereof, which solve the technical problems that: how to realize constant current output of the bidirectional MC-WPT system and how to optimize the magnetic coupling mechanism of the bidirectional MC-WPT system, so that the bidirectional MC-WPT system has higher level anti-offset capability to ensure stable and efficient transmission performance.
In order to solve the technical problems, the invention firstly provides an MC-WPT system, which comprises a ground end and a vehicle-mounted end;
the ground terminal comprises a ground terminal direct current power supply, a ground terminal full-bridge conversion module, a ground terminal resonant network, a ground terminal coupling structure, a ground terminal controller, a ground terminal communication module and a ground terminal current detector, wherein the ground terminal direct current power supply, the ground terminal full-bridge conversion module, the ground terminal resonant network and the ground terminal coupling structure are sequentially connected;
the vehicle-mounted terminal comprises a vehicle-mounted terminal coupling structure, a vehicle-mounted terminal resonant network, a vehicle-mounted terminal full-bridge conversion module, a filter circuit, a vehicle-mounted terminal battery, a vehicle-mounted terminal controller, a vehicle-mounted terminal communication module and a vehicle-mounted terminal current detector, wherein the vehicle-mounted terminal controller is connected with the vehicle-mounted terminal full-bridge conversion module;
the ground-end full-bridge conversion module and the vehicle-mounted-end full-bridge conversion module comprise a full-bridge inverter and four power diodes which are connected with four switching tubes of the full-bridge inverter in one-to-one reverse parallel connection;
when the ground end is used as an energy transmitting end to transmit energy forward to the vehicle-mounted endThe ground-end full-bridge conversion module works in an inversion state, converts the ground-end direct-current power supply into high-frequency alternating current, and the vehicle-mounted-end full-bridge conversion module works in a rectification state, four switching tubes of the vehicle-mounted-end full-bridge conversion module are turned off, a vehicle-mounted full-bridge rectification circuit is formed by four corresponding power diodes in inverse parallel connection, the high-frequency alternating current sensed by the vehicle-mounted-end coil is rectified into direct current, and finally the direct current is supplied to the vehicle-mounted-end battery for charging; the vehicle-mounted end current detector obtains the current I flowing through the vehicle-mounted end battery 2 The vehicle-mounted end communication module sends the I to the ground end communication module, and the ground end controller sends the I to the ground end communication module 2 With ground reference current value I set Comparing, and calculating a phase shift angle theta, so as to generate a switching tube PWM driving signal with the phase shift angle theta, and acting on the ground end full-bridge conversion module;
based on the same principle as the forward energy transmission process, the vehicle-mounted end serves as an energy transmitting end to transmit energy reversely to the ground end.
Preferably, the ground end controller will I 2 With ground reference current value I set And comparing to obtain a difference value delta I, and calculating a phase shift angle theta corresponding to the delta I according to a PI algorithm.
Preferably, the ground end resonant network and the vehicle end resonant network both adopt LCC resonant networks to form LCC-LCC resonant topology.
Preferably, the LCC-LCC resonant topology satisfies the condition:
wherein w represents the system working angular frequency, L r1 、C p 、C r1 Respectively representing parameter values corresponding to inductance, series capacitance and parallel capacitance in the LCC resonant network of the ground end, L r2 、Cs、C r2 And respectively representing parameter values corresponding to the inductance, the series capacitance and the parallel capacitance in the LCC resonant network of the vehicle-mounted terminal.
Preferably, the ground end controller is provided with a driving circuit, the driving circuit is formed by two-stage driving, a two-channel driving chip is used as a first-stage driving, and the driving circuit is used for lifting 3.3V driving pulse signals PWM1 and PWM2 output by the ground end controller to driving signals PWMH and PWML of 12V; the second-stage drive consists of an optocoupler isolation drive chip and a peripheral circuit thereof, outputs a totem pole drive structure, and finally outputs +18V/-3V grid drive voltages G_S and H_S for controlling the power MOSFET in the ground-end full-bridge conversion module to be turned on and off;
The vehicle-mounted end controller is arranged the same as the ground end controller.
Preferably, the ground end coupling structure comprises a ground end coil, and the vehicle-mounted end coupling structure comprises a vehicle-mounted end coil;
the ground end coil is formed by connecting an outer close-wound coil wound by a close-wound method and an inner loose-wound coil wound in the same direction with the outer close-wound coil by a loose-wound method in series; the vehicle-mounted end coil is wound in a double-layer close-winding mode.
Preferably, the ground end coil and the ground end coil are wound by litz wires with the same specification.
Preferably, the ground end coupling structure comprises a ground metal shielding plate, a ground magnetic core and the ground end coil which are arranged in a bottom-up level manner;
the vehicle-mounted end coupling structure comprises a vehicle-mounted end coil, a vehicle-mounted magnetic core and a vehicle-mounted metal shielding plate which are arranged in a bottom-to-top level manner;
the ground magnetic core and the vehicle-mounted magnetic core are formed by splicing a plurality of square magnetic cores.
The invention also provides a constant current output phase-shift control method of the bidirectional MC-WPT system, which is characterized by comprising a ground phase-shift control step of carrying out forward constant current output from the ground end to the vehicle-mounted end and a vehicle-mounted phase-shift control step of carrying out reverse constant current output from the vehicle-mounted end to the ground end, wherein the ground phase-shift control step comprises the following steps:
A1, acquiring current I flowing through the vehicle-mounted end battery 2 ；
A2, current I 2 With ground reference current value I set Comparing to obtain a difference delta I;
a3, judging whether the delta I is 0, if so, not taking the delta I as a phase shift angle theta corresponding to the delta I, and if not, calculating the phase shift angle theta according to a PI algorithm, and outputting a PWM driving signal with the phase shift angle theta to act on the ground-end full-bridge conversion module;
the principle of the vehicle-mounted phase shift control step is the same as that of the ground phase shift control step.
The invention provides a bidirectional MC-WPT system and a constant current output phase shift control method thereof, wherein a ground end controller, a ground end communication module and a ground end current detector are arranged at a ground end, and a vehicle-mounted end controller, a vehicle-mounted end communication module and a vehicle-mounted end current detector are arranged at a vehicle-mounted end, so that when the ground end is used as an energy transmitting end to transmit energy to the vehicle-mounted end in a forward direction, the vehicle-mounted end current detector is used for acquiring current I flowing through a vehicle-mounted end battery 2 And the phase shift angle theta corresponding to the delta I is calculated by the ground end controller according to a PI algorithm, acts on the ground end full-bridge conversion module to carry out phase shift control, and constant current output of the vehicle-mounted end is realized. Otherwise, when the vehicle-mounted terminal carries out energy reverse transmission on the ground terminal, the process is similar to the process. Thereby realizing the bidirectional constant current output control of the bidirectional MC-WPT system.
In the bidirectional MC-WPT system, the ground end coil adopts an outer dense inner sparse coil (comprising an outer dense coil and an inner sparse coil), the outer dense coil adopts a dense winding mode to ensure that the coil self-inductance and mutual inductance meet the most basic power transmission requirement, the inner sparse coil adopts a sparse winding mode to ensure that the magnetic field distribution is more uniform, and the anti-offset characteristic of the coil is comprehensively improved. It is worth mentioning that, compare in the close coil of tradition, the close internal sparse coil of outer density can reduce coil use line volume and cost, reduce coil self-inductance and internal resistance to effectively reduce the voltage and the coil loss at coil both ends. The vehicle-mounted end coil is wound in a multi-layer close winding mode, so that self inductance and mutual inductance of the vehicle-mounted end coil can be improved, and energy transmission can meet corresponding performance requirements.
Drawings
Fig. 1 is a schematic diagram of a ground end coil in a magnetic coupling mechanism of a bidirectional MC-WPT system provided by an embodiment of the present invention;
fig. 2 is a schematic diagram of a vehicle-mounted end in a magnetic coupling mechanism of a bidirectional MC-WPT system according to an embodiment of the present invention;
FIG. 3 is a trend graph of mutual inductance and rate of change thereof when the x-axis of four coils provided by an embodiment of the present invention are offset;
FIG. 4 is a graph showing the trend of the coupling coefficient and the change rate thereof when the x-axis of the four coils provided by the embodiment of the invention are shifted;
FIG. 5 is a front view of a magnetic coupling mechanism provided by an embodiment of the present invention;
FIG. 6 is a flow chart of a method for parameter design of a magnetic coupling mechanism of a bi-directional MC-WPT system provided by an embodiment of the invention;
FIG. 7 is a trend chart of the mutual inductance and the coupling coefficient change rate of four coils provided by the embodiment of the invention at the limit offset;
figure 8 is a block diagram of a bi-directional MC-WPT system provided by an embodiment of the present invention;
figure 9 is a situation diagram of a bidirectional MC-WPT system provided by an embodiment of the present invention when forward energy is transferred;
figure 10 is a situation diagram of a bidirectional MC-WPT system provided by an embodiment of the present invention when reverse energy is transmitted;
fig. 11 is a schematic diagram of a phase-shift control circuit of a full-bridge inverter according to an embodiment of the present invention;
FIG. 12 is a waveform diagram of the switching tube driving and inverting output voltage and current provided by an embodiment of the present invention;
FIG. 13 is a simulated waveform diagram of load switching during forward energy transfer provided by an embodiment of the present invention;
FIG. 14 is a simulated waveform diagram of the change in mutual inductance during forward transmission of energy provided by an embodiment of the present invention;
FIG. 15 is a simulated waveform diagram of the mutual inductance change during energy reverse transmission provided by an embodiment of the present invention;
FIG. 16 is a graph showing the comparison of M simulation values and actual measurement values of a coupling mechanism provided by an embodiment of the present invention when the x-axis and y-axis directions are offset;
FIG. 17 is a flow chart of operation of closed loop constant current control provided by an embodiment of the present invention;
FIG. 18 is a driving circuit diagram provided by an embodiment of the present invention;
FIG. 19 is a waveform diagram of an experiment for forward transmission of energy provided by an embodiment of the present invention;
FIG. 20 is a graph of output power and efficiency of the system for forward energy transfer according to an embodiment of the present invention;
fig. 21 is a waveform diagram of a load switching experiment during forward transmission of energy according to an embodiment of the present invention;
FIG. 22 is an experimental waveform diagram of a system for energy reverse transmission according to an embodiment of the present invention;
fig. 23 is a diagram of output power and efficiency of the system during energy reverse transmission according to an embodiment of the present invention.
Detailed Description
The following examples are given for the purpose of illustration only and are not to be construed as limiting the invention, including the drawings for reference and description only, and are not to be construed as limiting the scope of the invention as many variations thereof are possible without departing from the spirit and scope of the invention.
The embodiment of the invention firstly provides a magnetic coupling mechanism of a bidirectional MC-WPT system, which comprises a ground end coupling structure and a vehicle-mounted end coupling structure, wherein the ground end coupling structure comprises a ground end coil, and the vehicle-mounted end coupling structure comprises a vehicle-mounted end coil.
As shown in fig. 1 and 2, the ground end coil is formed by connecting an outer close-wound coil (coil 1) wound by a close-wound method and an inner loose-wound coil (coil 2) wound in the same direction with the outer close-wound coil by a loose-wound method in series, and the vehicle-mounted end coil is wound by a multi-layer close-wound method. The number of turns of the outer close-wound coil is N1, and the outer dimension is x 1 *y 1 . The number of turns of the inner sparse winding coil is N2, and the outer dimension is x 2 *y 2 The turn pitch is d. The spacing distance between the outer close winding coil and the inner loose winding coil is deltad。
The outer dense inner sparse type coil structure is characterized in that according to the size requirement of the coil, the outer coil can ensure that the self inductance and the mutual inductance of the coil meet the most basic power transmission requirement in a dense winding mode, and the inner coil can enable the magnetic field to be distributed more uniformly in a sparse winding mode, so that the anti-offset characteristic of the coil is comprehensively improved. It is worth mentioning that, compare in the close coil of tradition, the close internal sparse coil of outer density can reduce coil use line volume and cost, reduce coil self-inductance and internal resistance to effectively reduce the voltage and the coil loss at coil both ends. In addition, similar outer sparse inner dense coils have the characteristic of low self inductance and mutual inductance, and when the electric energy transmission with the same power level is realized, coil current can be increased, so that more loss is caused, and the transmission efficiency of a system is reduced.
The grouping series winding type coil is similar to the outer dense inner sparse type coil, the inner coil and the outer coil are combined in series, and the outer coil is tightly wound, wherein the difference is that the inner coil adopts a tightly wound winding mode. Therefore, compared with the outer-dense inner-sparse coil, the grouping series winding coil has the problems of larger wire consumption, larger coil self-inductance and internal resistance and higher cost.
In order to improve the transverse anti-offset capability of the coupling mechanism, a coil winding mode of grouping series winding is adopted, the grouping series winding type coil is similar to an outer dense inner sparse type coil, a mode of combining an inner coil and an outer coil in series is adopted, and the outer coils are densely wound, wherein the difference is that an inner coil of the former adopts a densely winding type coil. Therefore, compared with the outer-dense inner-sparse coil, the grouping series winding coil has the problems of larger wire consumption, larger coil self-inductance and internal resistance and higher cost. The following will comparatively analyze the coupling and anti-migration properties of these several different winding modes.
In order to analyze the coupling characteristics and the horizontal anti-offset characteristics of four winding modes (dense winding, sparse winding, grouping series winding, dense outside and sparse inside), a 3D simulation model is respectively built for simulation analysis by Mxawell simulation software. To ensure that the resulting data are comparative, the various parameter settings of the coil are required as follows:
(1) The Maxwell simulation conditions and the solver are set the same;
(2) The outer diameters of the ground end coil and the vehicle-mounted end coil are the same, the total turns of the ground end coil are 10 turns, the turns of the external coil are 6 turns, the turns of the internal coil are 4 turns, the initial winding points of the internal coil and the external coil are the same, the turn-to-turn distance is 5mm, and the turns of the vehicle-mounted end coil are 20 turns;
(3) The coupling mechanism was not added with a magnetic core, and the transmission distance d1 was kept uniform and set to 20cm.
The magnetic field distribution uniformity degree of the two winding modes of grouping series winding and outer dense inner sparse winding is not quite different to the naked eye, but the sparse winding type coil has the advantages in that the number of turns of the inner coil is the same, and the sparse winding type coil has relatively less wire consumption.
Simulation results of 4 winding modes (dense winding, sparse winding, grouping series winding and outer dense inner sparse) are arranged to obtain trend graphs of coil mutual inductance M and mutual inductance change rate of the coupling mechanism along with the change of the x-axis offset distance, wherein the trend graphs of coil coupling coefficient k and coupling coefficient change rate along with the change of the x-axis offset distance are shown in (a) and (b) of fig. 3.
As can be seen from fig. 3, when the coupling mechanism coil is offset in the x-axis direction, the mutual inductance value of the outer dense inner sparse coil is slightly smaller than that of the dense winding and the grouping series winding coil, but the mutual inductance change rate is lower than that of the other three winding modes at any time, and the gap is reduced along with the increase of the offset distance, mainly because the self inductance of the sparse winding coil is smaller than that of the dense winding coil, and after the offset distance is increased, the main factor affecting the mutual inductance and the change rate thereof becomes the magnetic field generated by the outer coil.
As can be seen from fig. 4, the coupling coefficient of the outer dense inner sparse coil is larger than that of the dense winding type and the grouping series winding type at any offset distance, and is only smaller than that of the sparse winding type coil, which indicates that the magnetic field of the outer dense inner sparse coil is more uniformly distributed, and the conclusion that the mutual inductance change rate of the outer dense inner sparse coil is the lowest in the four winding manners is also verified. It can also be seen from fig. 4 that the rate of change of the coupling coefficient is also at a lower level of the four ways. The X-axis offset resistance of the outer-dense inner-sparse coil is relatively better from the viewpoints of comprehensive mutual inductance, coupling coefficient and change rate thereof.
It is also available that when the coil is shifted in the y-axis, the trend of each coefficient is approximately the same as the trend of the previous x-axis, and the same points are not described in detail. The difference is that when the offset distance reaches the limit of 300mm, the coupling coefficients of the four modes tend to be the same value, because the system offset is too large, resulting in the coil mutual inductance being drastically reduced, and the magnetic field received by the secondary coil is already weak. When the offset distance of the y axis is smaller than 200mm, the coupling coefficient can be always kept above 0.1, and the mutual inductance and the coupling coefficient of the outer dense inner sparse coil can be kept at a relatively large value, and the change rate of the mutual inductance and the coupling coefficient is kept at a relatively small value. It is therefore considered that the outer dense inner sparse coil has relatively better resistance to deflection in the y-axis.
From the analysis, compared with other three winding modes, the outer-dense inner-sparse coil has better horizontal anti-deflection capability of the coupling mechanism.
And limiting the size of the vehicle-mounted end coil to 40 x 40cm according to the actual working condition. In the energy transmission process of the system, the vehicle-mounted end coil is used as an energy receiving end when in forward charging, and becomes an energy transmitting end when in reverse discharging, and the system is changed into a small-transmitting and large-receiving structure according to the size requirement.
The design of the vehicle-mounted end coil mainly considers the characteristic of transmission performance reduction caused by smaller size of the transmitting end in reverse transmission. If the coil self inductance and the mutual inductance are reduced by adopting the same winding mode with the large-size ground end, the excitation current I of the coil at the vehicle-mounted end is ensured to be transmitted at the same power level p Sum voltage U in And increases, thereby increasing system loss, ultimately resulting in a significant reduction in transmission efficiency. Therefore, the reverse transmission performance of the system needs to be improved by selecting a proper coil structure.
The mutual inductance between two coils is known from the Neiman formula:
m is mutual inductance and N Tx ，N Rx ，l Tx ，l Rx The number of turns of the primary and secondary side coils and the length of each turn of the coil are respectively, deltax is the relative distance when the two coils deviate, mu 0 Is air permeability.
It can be seen that the mutual inductance is in positive correlation with the number of turns and the length of the coil, and the number of turns and the length of the coil directly determine the self-inductance of the coil, so that in order to ensure that the energy transmission can reach the corresponding performance requirements, the self-inductance and the mutual inductance of the coils at the ground end and the vehicle-mounted end need to be ensured to be large enough.
In order to improve the self inductance and mutual inductance of the vehicle-mounted end coil, the common method is to enlarge the coil size and increase the number of coil layers, and the enlargement of the coil size in the horizontal direction contradicts the limitation of the vehicle-mounted end coil size in the embodiment, so that the enlargement of the coil size is not suitable for the system in the embodiment, and the embodiment selects to increase the number of coil layers in the vehicle-mounted end.
The coil size directly determines the self-inductance of the coil, and the inductance theoretical calculation formula of the multilayer planar spiral coil is as follows:
wherein: r is (r) s D is the distance between the effective center and the geometric center of the coil 2 For the effective width of the coil, h is the thickness of the multi-layer coil, d o The outer diameter of the coil is represented, and n is the number of layers of the coil; the inductance unit is uH.
From the above formula, it can be seen that when the coil self-inductance is proportional to the square of the coil outer diameter, it is inversely proportional to the coil inner diameter. It is apparent that the coil outer diameter parameter contributes more to the coil self-inductance value than the coil inner diameter, and that the coil self-inductance improving effect is still limited even if the number of coil turns is set to be large (the inner diameter is reduced) under the condition that the coil outer diameter is determined. For the coil structure of 'small emission and large receiving', the ground end is larger than the external size of the coil at the vehicle-mounted end, so that the coil at the vehicle-mounted end of the bidirectional MC-WPT system adopts a double-layer close-wound winding mode in order to meet the requirement of coil self-inductance.
As shown in fig. 5, the complete ground end coupling structure comprises a ground metal shielding plate, a ground magnetic core and a ground end coil which are arranged in a bottom-up level. The vehicle-mounted end coupling structure comprises a vehicle-mounted end coil, a vehicle-mounted magnetic core and a vehicle-mounted metal shielding plate which are arranged in a bottom-to-top level mode.
The primary and secondary coils of the bidirectional MC-WPT system have larger leakage inductance, and under the condition of limited coil size, the requirement of coil mutual inductance and coupling coefficient is hardly met by increasing the number of turns of the coils. The ferrite core is added to effectively converge magnetic force lines of the coil, optimize magnetic field distribution, greatly reduce the volume of the coupling mechanism and reduce the cost.
In practical systems, the use of the magnetic core needs to comprehensively consider relevant parameters such as magnetic permeability, magnetic saturation capacity and resistivity of the magnetic core, so as to reduce volume and loss, and in this embodiment, a PC40 magnetic core (manganese-zinc ferrite) with magnetic permeability of 2400H/m is selected. In order to increase the coupling coefficient and mutual inductance of the coupling mechanism as much as possible and improve the anti-offset capability of the system, the magnetic core of the coupling mechanism selects a full-spread placement mode. Considering that the coil sizes of the ground end and the vehicle-mounted end are larger, it is not practical to use only one complete magnetic core, so that a plurality of small square magnetic cores with the sizes of 10 x 0.5cm are adopted to splice the magnetic cores with the required sizes.
The energy of the MC-WPT system is transmitted in a magnetic field coupling mode, and the system always has magnetic leakage, so that the magnetic leakage of the system is required to be shielded in order to reduce the interference of the magnetic leakage of the system on instruments on the electric automobile. Ferrite cores can shield leakage magnetic flux to a certain extent while converging magnetic fields, but have limited effects. The charging system is generally enabled to achieve an approximately complete shielding effect by adopting a mode of adding a metal shielding layer to a ferrite magnetic core, the magnetic core of the coupling mechanism is formed by splicing small magnetic cores, and magnetic leakage exists in gaps, so that the shielding of the magnetic leakage of the system is achieved by adding a metal aluminum plate behind the magnetic core.
The embodiment researches a wireless charging system based on a static electric automobile, so that the longitudinal transmission distance of the system is considered to be constant, and only the anti-offset characteristic of the system when the coil is offset in the horizontal direction is considered. The size of the vehicle-mounted end of the coupling mechanism is limited by the size of the chassis of the electric vehicle, so that the vehicle-mounted end coil of the embodiment is limited to be wound in a square area of 40 x 40 cm. Considering that the distance from the chassis of the vehicle to the ground is 15-20cm in practice, the transmission distance of the system is set to d1=20 cm. In order to improve the anti-offset capability of the wireless charging system of the electric automobile in the horizontal direction, the size of the ground end of the system is usually designed to be larger than that of the vehicle-mounted end, and the size of the coil of the ground end of the system in the embodiment is limited to 65 x 50cm.
On the premise of limiting the outer diameter of the coil, the design and optimization of the outer-dense inner-sparse coil are focused on obtaining the optimal relation among the parameters of the outer coil turns N1 and the inner coil turns N2, the inner coil turn spacing d and the inner coil spacing delta d. The key to the design and optimization of the double-layer close-wound coil is the optimization of the number of turns Ns of the coil. The following provides a parameter design method flow of the magnetic coupling mechanism for the bidirectional MC-WPT system shown in fig. 6, and the optimal coil structure is finally obtained by designing and optimizing the coil parameters. The method comprises the following specific steps:
s1, determining the minimum mutual inductance Mmin and the minimum coupling coefficient Kmin of a magnetic coupling mechanism according to performance requirements;
s2, carrying out parameterized scanning on the turns of the coil at the vehicle-mounted end to determine the optimal turns;
s3, determining the external dimension x of the external closely-wound coil according to actual requirements 1 *y 1 ；
S4, a 3d simulation model is built, parameterization scanning is conducted on the number of turns N1 of the external close-wound coil, mutual inductance of the magnetic coupling mechanism is obtained, and the value range of the number of turns N1 when the mutual inductance of the magnetic coupling mechanism is larger than Mmin is obtained;
s5, determining the external dimension x of the internal sparse winding coil according to actual requirements 2 *y 2 ；
S6, carrying out parameterization scanning on the number of turns N2 of the inner loose coil and the interval distance delta d between the outer close coil and the inner loose coil to respectively obtain the coupling coefficient of the magnetic coupling mechanism, and obtaining the value range of the number of turns N2 and delta d when the mutual inductance of the magnetic coupling mechanism is larger than Mmin;
S7, performing parameterization scanning on the turn distance d of the internal sparse coil to obtain mutual inductance and coupling coefficient of the magnetic coupling mechanism, and determining the optimal turn distance d according to the mutual inductance and the coupling coefficient;
s8, determining a group of values in the respective value ranges of N1, N2 and Deltad, and acquiring the system anti-offset characteristic and the transmission characteristic of the group of values and the optimal turn spacing;
s9, judging whether the anti-offset characteristic and the transmission characteristic of the system meet the design requirements, if so, determining that the values of N1, N2, delta d and d at the moment are the final design parameter values, and if not, returning to the step S8.
According to the design method of the previous parameters, the optimal number of turns of the outer close winding coil is 10 turns, the number of turns of the inner close winding coil is 5 turns, and the optimal turn spacing is 0.8cm. The vehicle-mounted end coil adopts a double-layer close-wound winding mode, one layer of coil turns close to the ferrite core are 12 turns, and the other layer of coil turns is 10 turns.
According to the practical application requirement, the horizontal maximum offset of the coupling mechanism is 8cm for x-axis offset and 12cm for y-axis offset. To measure the anti-deflection capability of the designed coupling mechanism, the coil mutual inductance and the coupling coefficient change rate of the other three coil structures and the coil structures are compared and analyzed under the limit deflection condition, and the comparison result is shown in the following figure 7. It can be seen that the coil winding method adopted in this embodiment is at the minimum value in both the mutual inductance change rate and the coupling coefficient change rate, and is reduced by about 12% compared with the solvophobic winding method with the maximum change rate.
In order to improve the anti-offset characteristic of the bidirectional MC-WPT system of the electric automobile, the embodiment firstly gives out the specificity of the coupling mechanism of the bidirectional MC-WPT system, then analyzes the anti-offset characteristic of different coil modes (unipolar, bipolar, densely-wound and loosely-wound) through a finite element simulation software Maxwell, and obtains a conclusion that the anti-offset capability of the coil can be improved by properly selecting the two coil structures of densely-wound and loosely-wound. And then comprehensively considering mutual inductance, coupling coefficient and change rate thereof, and respectively adopting an external-dense internal-sparse mode and a double-layer dense winding mode for the ground end coil and the vehicle-mounted end coil to provide corresponding parameter design and optimization schemes. Compared with other coil structures, the designed coil reduces the change rate by 12% at most under the condition of ensuring sufficient mutual inductance, so that the system has better anti-offset capability.
It should be noted that, as shown in fig. 8, the bidirectional MC-WPT system applied by the magnetic coupling mechanism of this embodiment includes a ground terminal and a vehicle-mounted terminal, where the ground terminal includes a ground terminal dc power supply, a ground terminal full-bridge conversion module, a ground terminal resonant network, and a ground terminal coupling structure that are sequentially connected, and the vehicle-mounted terminal includes a vehicle-mounted terminal coupling structure, a vehicle-mounted terminal resonant network, a vehicle-mounted terminal full-bridge conversion module, a filter circuit, and a vehicle-mounted terminal battery that are sequentially connected.
The ground-end full-bridge conversion module and the vehicle-end full-bridge conversion module comprise a full-bridge inverter and four power diodes which are connected with four switching tubes of the full-bridge inverter in one-to-one reverse parallel.
The MC-WPT system adopting the SS topological structure has a simpler overall structure, but the anti-offset capability of the system is limited, and the problem of overcurrent exists when the secondary side runs in a no-load manner, so that the safety and the reliability of the system can be reduced, and the application scene is limited in many ways. LCL-LCL topology has advantages in working conditions such as anti-offset capability and no-load, but transmission power is reduced compared with SS topology, and is not suitable for a wireless energy transmission system with high power level. The LCC-LCC topology has the advantages of the LCL-LCL topology under special working conditions, and meanwhile, the problem of limited transmission power is avoided. Therefore, the resonance topology selected in this embodiment is an LCC-LCC topology, that is, the ground-side resonance network and the vehicle-side resonance network both adopt LCC resonance networks to form an LCC-LCC resonance topology.
In addition, the LCC-LCC resonant topology satisfies the condition:
wherein w represents the system working angular frequency, L r1 、C p 、C r1 Respectively representing corresponding parameter values of inductance, series capacitance and parallel capacitance in LCC resonant network of ground terminal, L r2 、Cs、C r2 Inductance, series capacitance and parallel electricity in LCC resonance network respectively representing vehicle-mounted end And the corresponding parameter value is contained. The MC-WPT system adopting the LCC-LCC resonance topology can ensure that the system works in a complete resonance condition when 4 conditions shown below are met. For MC-WPT system adopting LCC-LCC resonance topology, if the primary and secondary coil positions of the system remain unchanged and the resonance network parameters are determined, the current I is output 2 Is limited only by the input voltage U in It was determined that the LCC-LCC resonant topology therefore has the characteristic of maintaining a constant current output at constant voltage input to the system.
As shown in fig. 9, when the ground terminal performs energy reverse transmission to the vehicle-mounted terminal, the ground terminal full-bridge conversion module works in an inversion state to convert the ground terminal direct current power supply into high frequency alternating current, the vehicle-mounted terminal full-bridge conversion module works in a rectification state, four switching tubes of the vehicle-mounted terminal full-bridge conversion module are turned off, a vehicle-mounted full-bridge rectification circuit is formed by four corresponding power diodes in reverse parallel connection, the high frequency alternating current sensed by the vehicle-mounted terminal coil is rectified into direct current, and finally the direct current is supplied to the vehicle-mounted terminal battery for charging;
as shown in fig. 10, when the vehicle-mounted terminal performs forward energy transmission to the ground terminal, the vehicle-mounted terminal full-bridge conversion module works in an inversion state to convert the dc power provided by the vehicle-mounted terminal battery into high-frequency ac power, and the ground terminal full-bridge conversion module works in a rectification state, i.e., four switching tubes of the ground terminal full-bridge conversion module are turned off, and the corresponding four power diodes connected in reverse parallel form a ground full-bridge rectification circuit to rectify the high-frequency ac power sensed by the ground terminal coil into dc power and finally supply the ground terminal dc power.
For the bidirectional MC-WPT system of the electric automobile, when energy is transmitted in the forward direction, and the energy is transmitted to a load in a wireless mode from a power grid side, the loads of different electric automobiles are different, and the output characteristic of the system is unstable due to load change or coil mutual inductance fluctuation. When energy is reversely transmitted, although the problem of load change does not exist on the power grid side, the mutual inductance fluctuation phenomenon still exists, and the system output is unstable. To solve the above problem, the current on the energy receiving side is controlled to realize the efficient and stable output of the MC-WPT system.
According to the analysis result, the LCC-LCC resonance topology theory selected by the invention has the characteristic of constant current output, but the premise is to ensure stable mutual inductance value and complete resonance of the resonance network, and consider that the mutual inductance can change when the coil of the coupling mechanism is offset in practice, meanwhile, the resonance inductance capacitance value in the actual system is different from the theoretical calculation value (the resonance network is not in a complete resonance state), so that constant current output is kept, and besides the anti-offset characteristic of the coupling mechanism is improved, the constant current output still needs to be realized by a corresponding control strategy.
The invention selects phase-shift control as a constant-current control scheme of a bidirectional MC-WPT system. Although the adjustment range of the phase shift control is narrower, the anti-offset capacity of the system is improved and the constant current output is realized in the process of selecting the system resonance topology and designing and optimizing the coupling mechanism, so that the requirement on the adjustment range of the phase shift angle can be reduced, and the defect of the narrower adjustment range of the phase shift control is overcome to a certain extent.
The schematic diagram of the phase shift circuit of the full-bridge inverter is shown in fig. 11, under the phase shift control mode, the switching tubes S1 and S2 are still in complementary conduction, S1 and S2 are used as reference bridge arms, S3 and S4 are used as hysteresis arms, the phase difference between the two is 180 ° - θ, and θ is called phase shift angle.
As shown in fig. 12, the waveforms of the switching tube driving and inversion output voltage and current have 4 working modes in total in one period, and when the switching tube is in the working mode (1), the switching tubes S2 and S3 work, and the inverter output voltage is-Udc; when the power supply is in the working mode (2), the current is not commutated, the working of the switching tube S2 and the power diode D4 ensures the follow current of the circuit, and the output voltage of the inverter is 0; when in the working mode (3), the switching tubes S1 and S4 work, the output voltage of the inverter is Udc, when in the working mode (4), the current is not commutated, and at the moment, the switching tube S1 and the power diode D2 work to ensure that the circuit is free-wheeling, and the output voltage of the inverter is 0. It can be seen that by adjusting the value of the phase shift angle θ (0 < θ. Ltoreq.180°), the effective value of the output voltage of the full-bridge inverter circuit can be changed, thereby controlling the output power of the whole system.
The invention selects phase-shifting control as a constant-current control scheme of a bidirectional MC-WPT system, and refers to FIG. 8, which is a phase-shifting control structure block diagram of the bidirectional MC-WPT system. In order to facilitate analysis and experiment, when energy is transmitted in the forward direction, a circuit before the power grid alternating current is input to the inversion input is equivalent to a ground-end direct current source, a load is set to be a pure resistance load, and when energy is transmitted in the reverse direction, a circuit before the vehicle-mounted-end battery is input to the inversion input is equivalent to a vehicle-mounted-end direct current source, and at the moment, the load of the ground end is set to be the pure resistance load.
Therefore, in order to further realize bidirectional constant current output, as shown in fig. 8, the ground end of the bidirectional MC-WPT system provided in this embodiment includes a ground end controller connected to the ground end full-bridge conversion module, a ground end communication module connected to the ground end controller, and a ground end current detector, and the vehicle-mounted end further includes a vehicle-mounted end controller connected to the vehicle-mounted end full-bridge conversion module, and a vehicle-mounted end communication module connected to the vehicle-mounted end controller, and a vehicle-mounted end current detector.
When the ground end is used as an energy transmitting end to transmit energy forward to the vehicle-mounted end, the vehicle-mounted end current detector acquires current I flowing through the vehicle-mounted end battery 2 The vehicle-mounted terminal communication module sends the I to the ground terminal communication module, and the ground terminal controller sends the I to the ground terminal communication module 2 With ground reference current value I set And comparing, and calculating a phase shift angle theta, so as to generate a switching tube PWM driving signal with the phase shift angle theta, and acting on the ground end full-bridge conversion module.
Based on the same principle as the forward energy transmission process, the vehicle-mounted end serves as an energy transmitting end to transmit energy reversely to the ground end. That is, when the vehicle-mounted terminal is used as the energy transmitting terminal to transmit energy forward to the ground terminal, the ground terminal current detector acquires the current I flowing through the ground terminal DC power supply 1 The ground end communication module sends the I to the vehicle-mounted end communication module, and the vehicle-mounted end controller sends the I to the vehicle-mounted end communication module 1 With the on-vehicle reference current value I' set And comparing, and calculating a phase shift angle theta ', so as to generate a switching tube PWM driving signal with the phase shift angle theta' and act on the vehicle-mounted end full-bridge conversion module.
For a constant current output bi-directional MC-WPT system, the controlled amount varies with the direction of transmission. When the energy is transmitted forward, the controlled quantity is the output current I of the vehicle-mounted end 2 At this timeThe phase-shift control works as a ground-side controller, and the controlled quantity becomes the ground-side current I during the reverse transmission 1 The vehicle-mounted end controller plays a role in phase shift control. Therefore, in order to realize bidirectional constant current transmission, a communication module is needed to be added to collect the current I 1 、I 2 Respectively to the corresponding controllers.
Based on the design of the resonance parameters of the bidirectional MC-WPT system and the introduction and analysis of the constant current control strategy, a circuit simulation model is built through a Matlab/Simulink simulation software platform. The version of Matlab software used herein is Matlab 2019a. The circuit part of the simulation model is completely consistent with the constant current control strategy system block diagram shown in fig. 8.
The simulation parameters of the system are set up as shown in table 1 below.
Table 1 simulation parameters of bidirectional MC-WPT System
Fig. 13 is a simulation waveform when the system performs load switching during forward energy transfer, where (a) and (b) in fig. 13 are respectively amplified views of the inverted output voltage waveforms before and after the load switching, (c) is a trend of the phase shift angle before and after the load switching, and (d) is a dc output current waveform at both ends of the load. The current reference value of the system is 8.3A, and the load value is switched from 43Ω to 21Ω at the moment of 0.05s, so as to simulate the process of switching the output power from full power operation to 50% power operation in the actual system. It can be seen that when the dc input voltage is constant, as the load resistance is switched from 43Ω to 21Ω, the effective value of the inverter voltage and the output voltage are reduced by the phase shift control, and the output current can be kept constant before and after the load switching.
The working mechanism is as follows: when a control link is not added, the equivalent impedance of the secondary side is suddenly reduced by switching the load (from 43Ω to 21Ω), the primary side current is not changed, and the induced voltage of the secondary side is unchanged, so that the current flowing through the load is suddenly increased. After the primary side is controlled, load switching can lead to sudden rise of load current, a difference value appears between the load current and a reference current value, a phase shifting controller reduces a phase shifting angle, an effective value of an inversion output voltage can be reduced, an induction voltage and a current of a secondary side coil can be correspondingly reduced, and finally, the load current is stabilized at a set value, so that the effect of outputting constant current by a system is achieved. It is worth to say that, load current will have a surge process before and after load switching, because the load switching process is too short for the LCC-LCC higher-order system, the system has a certain reaction time, the phase shift angle is reduced with a time delay, and the surge of load current will occur.
Fig. 14 is a simulation waveform of the system in the case of the mutual inductance change during the forward transmission of energy, wherein (a) and (b) are respectively amplified diagrams of the inverted output voltage waveforms before and after the mutual inductance change, (c) is a trend diagram of the phase shift angle before and after the load switching, and (d) is a waveform of the direct output current at both ends of the load. As can be seen from the waveform diagram, when the mutual inductance changes, the phase shift angle increases, the effective value of the primary coil current increases, and the output current can be kept unchanged.
The current reference value of the system is still set to be 8.3A, and the mutual inductance of the coil changes at the moment of 0.04s (the current is reduced from M=38uH to M=30uH) and is used for simulating the working condition that the coil is offset in practice to cause the mutual inductance change, and the values before and after the mutual inductance change in simulation are respectively the mutual inductance 38uH of the coil in the positive direction and the mutual inductance 30uH of the coil under the limit offset. The working mechanism is as follows: the direct current input voltage is kept constant, the mutual inductance of the system coil is reduced, the secondary side induced voltage is reduced, the load is unchanged, and the output current flowing through the load is directly reduced. At this time, the load current and the reference value have a difference value, and under the action of the primary phase-shifting controller, the phase-shifting angle is increased to increase the inversion output voltage, so as to increase the primary coil current, counteract the trend of decreasing the output current, and finally make the output current of the system constant.
Fig. 15 shows the simulation results of the system in the working state of the mutual inductance change during the energy reverse transmission, wherein the simulation results comprise an amplified graph of the waveform of the inversion output voltage before and after the mutual inductance change, a phase-shift angle change graph and the waveform of the direct current output current at the two ends of the load. It can be seen that when energy is reversely transmitted, the simulation system can also make the current of the output end constant by adjusting the phase shift angle, and the basic principle of control is exactly the same as that of forward transmission, and is not repeated here.
In summary, no matter the energy is transmitted forward or backward, the simulation model can ensure that the output current is kept constant when the load is switched and the mutual inductance is changed, and the simulation result corresponds to the theoretical analysis one by one.
The following experiments were performed for verification.
First, the basic performance indexes of the bidirectional MC-WPT system designed in the text are as follows:
(1) Wireless energy transmission distance: 20cm;
(2) Coupling mechanism coil X/Y axis offset: 8cm/12cm;
(3) Switching device operating frequency: 85kHz;
(4) Full load input and output voltage: 360VDC;
(5) Forward and reverse full power: 3000W;
(6) The transmission efficiency of the whole machine is as follows: 85% or more.
Litz wire is used herein to wind the primary and secondary coils. The litz wire is formed by twisting a plurality of fine wires, the influence of skin effect on system parameters and efficiency can be effectively reduced, the diameter of the litz wire is selected according to the maximum current passing through the coil, and the maximum coil current I can be obtained coil ：
The design reference value of the coil current is 22.1A, considering a margin of 30% at the time of design. According to litz wire selection specifications, litz wire with a wire diameter of about 4.5mm is finally selected, the wire core of the litz wire is 0.1mm multiplied by 1000, and the withstand voltage grade is 3kV.
The primary coil adopts an outer dense inner sparse winding method, the outer coil is 10 turns, the inner coil is 5 turns, and the inter-turn distance is 0.8cm; the secondary side coil is tightly wound in a double-layer manner, and the number of turns is 12+10. The measured values of the relevant parameters of the coupling mechanism are shown in the following table 2, wherein the coil opposite direction means that the offset of the primary side coil and the secondary side coil in the x axis and the y axis is 0, and the coil limit offset means that the offset of the coil in the x axis is 8cm and the offset in the y axis is 12cm.
Table 2 actual measurement values of parameters of the coupling mechanism
From the actual measurement value of the coil, the coil is in a state of being opposite to the limit offset, the system mutual inductance is changed from 38.5uH to 30uH, the change rate is 22%, and the simulation result is close to that of the third chapter. To further verify the anti-offset characteristics of the system, the mutual inductance of each coil is measured when the coils are offset to different degrees in the x-axis direction and the y-axis direction, and the actual measurement result is shown in fig. 16. It can be seen that a certain gap exists between the actually measured mutual inductance and the simulation result, because the actual wound coil density and the turn-to-turn distance cannot be as accurate as the simulation, but the overall mutual inductance change trend and the overall mutual inductance change rate are the same as the simulation, which shows that the wound coil has good horizontal anti-offset capability and can improve the redundancy of closed-loop control of the system.
The main control modules of the ground end and the vehicle-mounted end adopt a DSP control chip TMS320F28335 as a main controller, and are mainly responsible for the work of control, analog-to-digital conversion, driving signal output, communication and the like. The working flow of the constant current output phase shift control method of the bidirectional MC-WPT system is shown in the following figure 17. When the system is started, firstly, detection of current to be controlled is completed through a current detection part, data are transmitted to a corresponding main control chip through communication and A/D conversion is completed, then an actual sampling value is compared with a preset current value to obtain a difference value, a corresponding phase shift angle theta is obtained through a PI algorithm, and finally, the main control chip outputs a PWM driving signal with the phase shift angle theta. Specifically, corresponding to fig. 8, the constant current output phase shift control method includes a ground phase shift control step of performing forward constant current output from a ground end to a vehicle-mounted end, and a vehicle-mounted phase shift control step of performing reverse constant current output from the vehicle-mounted end to the ground end, where the ground phase shift control step includes:
a1, acquiring current I flowing through a vehicle-mounted end battery 2 ；
A2, current I 2 With ground reference current value I set Comparing to obtain a difference delta I;
a3, judging whether the delta I is 0, if so, not taking the delta I as a phase shift angle theta corresponding to the delta I, and if not, calculating the phase shift angle theta according to a PI algorithm, and outputting a PWM driving signal with the phase shift angle theta to act on the ground-end full-bridge conversion module.
The principle of the vehicle-mounted phase shift control step is the same as that of the ground phase shift control step, and specifically, the method comprises the following steps:
b1, obtaining the current I flowing through the battery at the ground end 1 ；
B2, current I 1 With the on-vehicle reference current value I' set Comparing to obtain a difference delta I';
and B3, judging whether the delta I 'is 0, if so, not, if not, calculating a phase shift angle theta' corresponding to the delta I 'according to a PI algorithm, and outputting a PWM driving signal with the phase shift angle theta' to act on the vehicle-mounted full-bridge conversion module.
The PI parameter setting adopts a trial-and-error method commonly used in engineering, that is, initial parameter values Kp and Ti are set in a DSP program, and PI parameters are modified one by observing dynamic performance (power rise time, overshoot, etc.) in a smaller power state.
As shown in fig. 18, the driving circuit of the power device is formed by two-stage driving, the two-channel driving chip UCC27524 is used as the first-stage driving, and the main functions of the driving circuit are to boost the 3.3V driving pulse signals PWM1 and PWM2 output by the controller into the driving signals PWMH and PWML of 12V; the second stage driving is composed of an optocoupler isolation driving chip HCPL-3120 and peripheral circuits thereof, the output is connected with a totem pole driving structure, and finally, grid driving voltages G_S and H_S which are used for controlling the power MOSFET to be turned on and off are output by +18V/-3V.
The power switching tube is important for the full-bridge inverter circuit, and the switching tube has enough voltage and current stress according to the working condition of an actual system. MOSFETs are power switching devices commonly used in MC-WPT systems, and can be classified into silicon (Si) MOSFETs, silicon carbide (SiC) MOSFETs, and the like, depending on the materials thereof. Based on some key characteristics of SiC MOSFETs, in view of sufficient voltage and current margin, the intel slush FF23MR12W1M1 is selected to be used as an inverter power device, and 2 power MOSFETs are directly connected in series to form a bridge arm, and basic operation parameters are shown in table 3.
TABLE 3 FF23MR12W1M1 basic operating parameters
For the diode reversely connected with the power MOSFET in series in the system, the function is to realize rectification when the corresponding power MOSFET is in an off state, and the design margin of voltage and current in the system is considered, the model of the reverse-series diode used in the system is G3S06010J, the maximum reverse withstand voltage is 600V, and the maximum forward conduction current is 30A, so that the design requirement of the system is met.
The current detection chip on the power circuit is an ACS712ELCTR-20A-T chip based on the Hall effect, the maximum detection current is 20A, the maximum detection sensitivity is 100mV/A, the maximum sampling error is 1.5 percent, the detected value is a current value flowing through the main power circuit, the current is converted into a corresponding voltage signal I_ADC through the current detection chip, and the signal is sent to the DSP for control and protection.
The experimental results were then analyzed from both the forward and reverse energy transmission aspects in order to verify that the system designed herein was capable of bi-directional energy transmission and has anti-migration capabilities.
When the system energy is transmitted forward, the transmission distance is 20cm, the waveform diagram and the direct current source and electronic load indication of the coil under the conditions of opposite direction and limit deviation are shown in figure 19. The waveform diagrams respectively show the inversion output voltage U of the system under the condition of different coil offsets ab Inversion output current I ab Dc output voltage V out And a direct current output current I out Is a test waveform of (a). As can be seen from the graph, the system can keep 8.32A constant current output under the working conditions of opposite direction and limit deviation of the coil, and the difference between the coil and the set current of 8.35A is that certain error exists in the detection of the system current, and the inversion output voltage U in the waveforms of the coil and the set current is compared ab The phase shift angle of the system when the coil is aligned is obviously smaller than that of the coil in a limit offset state, which indicates that the control of the ground end of the system is realized when the mutual inductance is reducedThe controller controls the current by increasing the phase shift angle.
Fig. 20 shows a variation curve of transmission power and efficiency of the system coil under the condition of different offsets in the x-axis direction and the y-axis direction respectively in the forward constant current transmission process. As can be seen from the graph, the system power can be close to 3000W, and the fluctuation is small under the condition of different offsets of the x-axis and the y-axis; and as the offset in the x-axis direction and the y-axis direction increases, the transmission efficiency of the system can be reduced, the highest transmission efficiency can reach 89.7%, and the designed system meets the corresponding performance index requirements. The actual output power P will be referred to herein for the effect of the control method used on the output power (output power ripple rate) o And the expected power P e (P e Difference rate Δp=3000W as a measure of power fluctuation, Δp= (P) o -P e )/P e The maximum output power fluctuation rate in the forward transmission of energy can be obtained according to the actually measured output power
ΔP max ＝(P o-min -P e )/P e ＝1.16％
FIG. 21 is a system output voltage V for forward transmission of system energy and switching of load out And output current I out Is a waveform of the experiment. It can be seen that the system can maintain constant output current through phase shift control after load changes, and different load resistance sizes (43Ω, 32Ω and 21Ω) respectively correspond to full load, 75% rated power and 50% rated power.
Similar to the forward transmission experiment, the waveform diagram and DC source and electronic load readings are given when the coil is in the opposite direction and the limit offset when the energy is transmitted in the reverse direction and the transmission distance is 20cm as shown in the following figure 22. The waveform diagram shows the inversion output voltage U of the system under different offset conditions ab Inversion output current I ab A DC output voltage Vout and a DC output current I out Is a test waveform of (a). The graph shows that the system can keep 8.32A constant current output under the working conditions of opposite direction and limit deviation of the coil, and the two inverted output voltages U are compared ab The root mean square value of (2) can be used for obtaining a conclusion similar to a forward transmission experiment, namely, when the mutual inductance of the system changes, the system is switched on The phase-shifting angle of the over-regulated vehicle-mounted end controller realizes the constancy of the ground end output current, and the difference is only that the controller and the controlled quantity are mutually exchanged.
Fig. 23 shows a change curve of transmission power and efficiency of the system coil under the condition of different offsets in the x-axis direction and the y-axis direction respectively in the reverse constant current transmission process. As can be seen from the figure, the magnitude of the transmission power and the fluctuation are not much different from those of the forward transmission; in terms of efficiency, the overall trend of reverse transmission is the same as that of forward transmission (the overall trend is reduced along with the increase of offset distance), but the overall efficiency of reverse transmission is higher, the highest efficiency can reach more than 90.3%, and even the lowest efficiency is 0.3 percent higher than the highest efficiency of forward transmission, because the on-board end coil is used as an excitation end to be self-induced to be larger than the ground end coil during reverse transmission, the required coil current is smaller during the transmission of the same power, the loss of the internal resistance of the coil is reduced, and the transmission efficiency is also increased. Similarly, the maximum output power fluctuation rate in energy reverse transmission can be obtained according to the actual output power
ΔP′ max ＝(P′ o-min -P e )/P e ＝0.91％
Combining the experimental results and analysis of the forward and reverse transmission of the energy of the previous system, a conclusion can be drawn: under two energy transmission modes, the system can achieve the design requirements on transmission power and efficiency, the maximum output power fluctuation rate is about 1.0%, and constant output of current can be achieved through phase shift control under the working condition that mutual inductance changes caused by coil deviation and the working condition that forward transmission loads are switched.
In conclusion, the bidirectional MC-WPT system with 360V input voltage, 8.3A output current and 3000W full load power is built. The actual measurement result shows that when the coil of the coupling mechanism is deviated, the actual measurement value of the coil parameter is basically consistent with the simulation result, which shows that the wound coil has better horizontal deviation resistance and can improve the redundancy of closed-loop control of the system; experimental results prove that the system can achieve the design requirements on transmission power and efficiency under different working conditions of two energy transmission modes, and constant current output can be achieved through phase shift control.
The above examples are preferred embodiments of the present invention, but the embodiments of the present invention are not limited to the above examples, and any other changes, modifications, substitutions, combinations, and simplifications that do not depart from the spirit and principle of the present invention should be made in the equivalent manner, and the embodiments are included in the protection scope of the present invention.

Claims (8)

1. The bidirectional MC-WPT system is characterized in that: the system comprises a ground end and a vehicle-mounted end;
the ground terminal comprises a ground terminal direct current power supply, a ground terminal full-bridge conversion module, a ground terminal resonant network, a ground terminal coupling structure, a ground terminal controller, a ground terminal communication module and a ground terminal current detector, wherein the ground terminal direct current power supply, the ground terminal full-bridge conversion module, the ground terminal resonant network and the ground terminal coupling structure are sequentially connected;
The vehicle-mounted terminal comprises a vehicle-mounted terminal coupling structure, a vehicle-mounted terminal resonant network, a vehicle-mounted terminal full-bridge conversion module, a filter circuit, a vehicle-mounted terminal battery, a vehicle-mounted terminal controller, a vehicle-mounted terminal communication module and a vehicle-mounted terminal current detector, wherein the vehicle-mounted terminal controller is connected with the vehicle-mounted terminal full-bridge conversion module;
the ground-end full-bridge conversion module and the vehicle-mounted-end full-bridge conversion module comprise a full-bridge inverter and four power diodes which are connected with four switching tubes of the full-bridge inverter in one-to-one reverse parallel connection;
when the ground end is used as an energy transmitting end to forward transmit energy to the vehicle-mounted end, the ground end full-bridge conversion module works in an inversion state to convert the ground end direct current power supply into high-frequency alternating current, the vehicle-mounted end full-bridge conversion module works in a rectification state, four switching tubes of the vehicle-mounted end full-bridge conversion module are turned off, a vehicle-mounted full-bridge rectification circuit is formed by four corresponding power diodes in reverse parallel connection, the high-frequency alternating current sensed by the vehicle-mounted end coil is rectified into direct current, and finally the direct current is supplied to the vehicle-mounted end battery for charging; the vehicle-mounted end current detector obtains the current flowing through the vehicle-mounted end battery Is the current I of (2) 2 The vehicle-mounted end communication module sends the I to the ground end communication module, and the ground end controller sends the I to the ground end communication module 2 With ground reference current value I set Comparing, and calculating a phase shift angle theta, so as to generate a switching tube PWM driving signal with the phase shift angle theta, and acting on the ground end full-bridge conversion module;
based on the same principle as the forward energy transmission process, the vehicle-mounted end is used as an energy transmitting end to transmit energy reversely to the ground end;
the ground end coupling structure comprises a ground end coil, and the vehicle-mounted end coupling structure comprises a vehicle-mounted end coil;
the ground end coil is formed by connecting an outer close-wound coil wound by a close-wound method and an inner loose-wound coil wound in the same direction with the outer close-wound coil by a loose-wound method in series; the vehicle-mounted end coil is wound in a double-layer close-winding mode;
the number of turns N1, N2 of the outer and inner coils of the ground end coil, the inner coil turn spacing d, and the inner and outer coil spacing Δd are determined by:
s1, determining the minimum mutual inductance Mmin and the minimum coupling coefficient Kmin of a magnetic coupling mechanism according to performance requirements;
s2, carrying out parameterized scanning on the turns of the coil at the vehicle-mounted end to determine the optimal turns;
S3, determining the external dimension x of the external closely-wound coil according to actual requirements 1 *y 1 ；
S4, a 3d simulation model is built, parameterization scanning is conducted on the number of turns N1 of the external close-wound coil, mutual inductance of the magnetic coupling mechanism is obtained, and the value range of the number of turns N1 when the mutual inductance of the magnetic coupling mechanism is larger than Mmin is obtained;
s5, determining the external dimension x of the internal sparse winding coil according to actual requirements 2 *y 2 ；
S6, carrying out parameterization scanning on the number of turns N2 of the inner loose coil and the interval distance delta d between the outer close coil and the inner loose coil to respectively obtain the coupling coefficient of the magnetic coupling mechanism, and obtaining the value range of the number of turns N2 and delta d when the coupling coefficient of the magnetic coupling mechanism is larger than Kmin;
s7, performing parameterization scanning on the turn distance d of the internal sparse coil to obtain mutual inductance and coupling coefficient of the magnetic coupling mechanism, and determining the optimal turn distance d according to the mutual inductance and the coupling coefficient;
s8, determining a group of values in the respective value ranges of N1, N2 and Deltad, and acquiring the system anti-offset characteristic and the transmission characteristic of the group of values and the optimal turn spacing;
s9, judging whether the anti-offset characteristic and the transmission characteristic of the system meet the design requirements, if so, determining that the values of N1, N2, delta d and d at the moment are the final design parameter values, and if not, returning to the step S8.
2. The bi-directional MC-WPT system of claim 1 wherein: the ground end controller will I 2 With ground reference current value I set And comparing to obtain a difference value delta I, and calculating a phase shift angle theta corresponding to the delta I according to a PI algorithm.
3. The bi-directional MC-WPT system of claim 1 wherein: the ground end resonant network and the vehicle-mounted end resonant network both adopt LCC resonant networks to form LCC-LCC resonant topology.
4. A bi-directional MC-WPT system as claimed in claim 3, characterized in that the LCC-LCC resonant topology satisfies the condition:
wherein w represents the system working angular frequency, L r1 、C p 、C r1 Respectively representing parameter values corresponding to inductance, series capacitance and parallel capacitance in the LCC resonant network of the ground end, L r2 、Cs、C r2 And respectively representing parameter values corresponding to the inductance, the series capacitance and the parallel capacitance in the LCC resonant network of the vehicle-mounted terminal.
5. The bi-directional MC-WPT system of claim 1 wherein: the ground end controller is provided with a driving circuit, the driving circuit is formed by two-stage driving, a two-channel driving chip is used as a first-stage driving, and the driving circuit is used for lifting 3.3V driving pulse signals PWM1 and PWM2 output by the ground end controller to driving signals PWMH and PWML of 12V; the second-stage drive consists of an optocoupler isolation drive chip and a peripheral circuit thereof, outputs a totem pole drive structure, and finally outputs +18V/-3V grid drive voltages G_S and H_S for controlling the power MOSFET in the ground-end full-bridge conversion module to be turned on and off;
The vehicle-mounted end controller is arranged the same as the ground end controller.
6. The bi-directional MC-WPT system of claim 5 wherein:
the ground end coil and the ground end coil are wound by litz wires with the same specification.
7. The bi-directional MC-WPT system of claim 5 wherein:
the ground end coupling structure comprises a ground metal shielding plate, a ground magnetic core and the ground end coil which are arranged in a bottom-to-top level manner;
the vehicle-mounted end coupling structure comprises a vehicle-mounted end coil, a vehicle-mounted magnetic core and a vehicle-mounted metal shielding plate which are arranged in a bottom-to-top level manner;
the ground magnetic core and the vehicle-mounted magnetic core are formed by splicing a plurality of square magnetic cores.
8. The constant current output phase shift control method of a bidirectional MC-WPT system according to any one of claims 1 to 7, characterized by comprising a ground phase shift control step of performing forward constant current output from the ground terminal to the vehicle-mounted terminal, and a vehicle-mounted phase shift control step of performing reverse constant current output from the vehicle-mounted terminal to the ground terminal, the ground phase shift control step comprising:
a1, acquiring current I flowing through the vehicle-mounted end battery 2 ；
A2, current I 2 With ground reference current value I set Comparing to obtain a difference delta I;
a3, judging whether the delta I is 0, if so, not taking the delta I as a phase shift angle theta corresponding to the delta I, and if not, calculating the phase shift angle theta according to a PI algorithm, and outputting a PWM driving signal with the phase shift angle theta to act on the ground-end full-bridge conversion module;
the principle of the vehicle-mounted phase shift control step is the same as that of the ground phase shift control step.
CN202210841334.6A 2022-07-18 2022-07-18 Bidirectional MC-WPT system and constant-current output phase-shifting control method thereof Active CN115276256B (en)

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