CN115214394B

CN115214394B - Dynamic wireless charging system of electric automobile and lateral offset power fluctuation suppression method

Info

Publication number: CN115214394B
Application number: CN202210840236.0A
Authority: CN
Inventors: 肖静; 唐春森; 王智慧; 龚文兰; 吴晓锐; 韩帅; 陈绍南; 吴宁; 莫宇鸿; 陈卫东; 郭敏; 郭小璇; 左志平; 李小飞; 史可
Original assignee: Chongqing University; Electric Power Research Institute of Guangxi Power Grid Co Ltd
Current assignee: Chongqing University; Electric Power Research Institute of Guangxi Power Grid Co Ltd
Priority date: 2022-07-18
Filing date: 2022-07-18
Publication date: 2023-04-21
Anticipated expiration: 2042-07-18
Also published as: CN115214394A

Abstract

The invention provides an electric automobile dynamic wireless charging system and a transverse offset power fluctuation suppression method, wherein the system comprises an electric energy transmitting end and an electric energy receiving end, the electric energy transmitting end is provided with a plurality of sections of energy transmitting guide rails which are arranged side by side one by one, each section of energy transmitting guide rail comprises an energy transmitting coil wound along a first direction and a primary side compensation coil which is connected with the energy transmitting coil in series and wound along the opposite direction of the first direction, the electric energy receiving end comprises an energy receiving coil and a variable integrated compensation inductance, the variable integrated compensation inductance comprises an alternating current inductance coil serving as a secondary side compensation network and two direct current control windings for changing the equivalent self inductance of the alternating current inductance coil, and the two direct current control windings are respectively arranged at the inner side and the outer side of the alternating current inductance coil according to opposite winding directions. The effect is that: the transverse offset resistance is further improved on the basis of power fluctuation suppression when the guide rail is switched, and the compensation of the system output power drop when the transverse offset is realized by using smaller direct current.

Description

Dynamic wireless charging system of electric automobile and lateral offset power fluctuation suppression method

Technical Field

The invention relates to a wireless power transmission technology, in particular to a dynamic wireless charging system of an electric automobile and a lateral offset power fluctuation suppression method.

Background

Wireless power transmission (wireless power transfer, WPT) achieves non-electrical contact transmission of electrical energy through magnetic fields, electric fields, lasers, microwaves and other mediums. The technology can effectively solve the problems of limited flexibility and potential safety hazard of equipment caused by the traditional wired power-taking mode. At present, in the application fields of electric automobiles, consumer electronics products, household appliances and the like, expert scholars at home and abroad have developed researches on the technology and obtain more theoretical results.

For an electric automobile, in order to realize continuous wireless power supply in the driving process, a dynamic wireless charging system of the electric automobile is proposed at present, and wireless energy dynamic transmission is realized through a wireless energy generating coil in a multi-section short guide rail form. Meanwhile, aiming at the problem of power fluctuation in the dynamic charging process, the research team has proposed an electric automobile dynamic wireless charging system for restraining power fluctuation through magnetic integration, and the research team specifically refers to Chinese patent invention: 202111524549.7 it is found that, during the use, the electric automobile inevitably generates a lateral offset and always continuously moves along the road under a certain lateral offset state, and tests show that, although the power drop caused by the switching of the guide rail in the prior art realizes power compensation, the output power still drops under the lateral offset, as shown in fig. 1, under the condition that y=5 cm and y=10 cm are lateral offsets, although the equivalent mutual inductance basically maintains stable during the position movement of the receiving coil, under different offset conditions, the equivalent mutual inductance still changes, and the farther the offset is, the lower the equivalent mutual inductance is, so that a system and a method capable of realizing the suppression of the lateral offset power fluctuation still need to be continuously studied.

Disclosure of Invention

In order to solve the problem of power drop caused by guide rail switching in a transverse offset state, the primary aim of the invention is to provide a dynamic wireless charging system of an electric automobile, which can realize power fluctuation suppression caused by transverse offset of the electric automobile on the basis of the prior art.

In order to achieve the above purpose, the specific technical scheme adopted by the invention is as follows:

the utility model provides an electric automobile developments wireless charging system, includes electric energy transmitting terminal and electric energy receiving terminal, its key lies in, electric energy transmitting terminal adopts multistage energy transmission guide rail to set up side by side one, and every section energy transmission guide rail includes along the energy transmission coil of first direction coiling and with energy transmission coil establish ties and along the primary side compensation coil of first direction opposite direction coiling, electric energy receiving terminal includes energy receiving coil and variable integrated compensation inductance, variable integrated compensation inductance includes as the alternating current inductance coil of secondary side compensation network and is used for changing two DC control windings of alternating current inductance coil equivalent self-inductance, two DC control windings set up respectively according to opposite coiling the inside and outside both sides of alternating current inductance coil.

Alternatively, a dc power supply is applied to both dc control windings, wherein the direction of the magnetic field generated by the current flowing through the inner dc control winding is opposite to the direction of the magnetic field generated by the current flowing through the outer dc control winding.

Optionally, the number of turns of the inner dc control winding is the same as the number of turns of the outer dc control winding.

Optionally, a horizontal distance is arranged between two adjacent sections of energy emission guide rails according to a first preset value.

Optionally, the electric energy transmitting end is provided with a primary LCC resonance compensation network, and the energy transmitting coil is also provided with a primary series resonance capacitor.

Optionally, the energy transmitting coil is wound horizontally according to a rectangular coil, the primary side compensation coil is wound in the energy transmitting coil according to the same plane, the energy receiving coil is wound horizontally according to the rectangular coil, the width of the outer contour of the energy receiving coil is the same as the width of the energy transmitting coil, and the alternating current induction coil and the two direct current control windings in the variable integrated compensation inductor are wound in the energy receiving coil according to the same plane.

Optionally, a primary magnetic core plate is arranged on the bottom surfaces of the energy transmitting coil and the primary compensating coil, and a secondary magnetic core plate and a shielding plate are arranged on the top surfaces of the energy receiving coil and the variable integrated compensating inductor.

Based on the system, the invention also provides a lateral offset power fluctuation suppression method of the dynamic wireless charging system of the electric automobile, which is characterized by comprising the following steps:

s1: detecting the transverse offset of the electric automobile;

s2: determining the falling condition of a first mutual inductance coefficient according to the transverse offset, wherein the first mutual inductance coefficient is the mutual inductance coefficient between two adjacent energy transmitting coils and an energy receiving coil;

s3: and the second mutual inductance is changed by adjusting the driving currents of the two direct current control windings, so that the output power of the system is maintained within a preset range, and the second mutual inductance is the mutual inductance between the energy receiving coil and the variable integrated compensation inductor.

Optionally, according to system parameter configuration, a mapping relation between the transverse offset and the first mutual inductance is pre-established through simulation, and meanwhile, a mapping relation between the driving currents of the two direct current control windings and the second mutual inductance is pre-established through simulation.

Optionally, in step S3, as follows

R _e To determine the variation relation between the first mutual inductance and the second mutual inductance, wherein P _out For output power, M _ABs As the first mutual inductance coefficient, M _sf3 For the second mutual inductance coefficient, omega is the system working angular frequency, L _f For compensating inductance value, L in primary side LCC resonance compensation network _se R is equivalent inductance value of secondary side resonance network _e For equivalent load +.>

Phasor expression of inverter alternating current output voltage, when electric automobile transversely deviates to cause first mutual inductance coefficient M _ABs When the current is changed, the second mutual inductance M is changed by adjusting the driving current of the two direct current control windings _sf3 So that the output power P _out Maintaining a predetermined range.

The invention has the following effects:

according to the dynamic wireless charging system and the lateral offset power fluctuation suppression method for the electric automobile, provided by the invention, based on the application of the variable integrated inductance coil, the lateral offset resistance is further improved on the basis of power fluctuation suppression during rail switching, and the compensation of the system output power drop during lateral offset is realized by using smaller direct current.

Drawings

In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the drawings used in the description of the embodiments or the prior art will be briefly described below.

FIG. 1 is a graph of the variation of equivalent mutual inductance at different lateral offsets in the prior art;

fig. 2 is a schematic structural diagram of an electric vehicle dynamic wireless charging system according to an embodiment;

fig. 3 is a schematic circuit diagram of an electric vehicle dynamic wireless charging system according to an embodiment;

FIG. 4 is an equivalent decoupling circuit of a secondary side compensation network;

FIG. 5 is an equivalent circuit considering same-side inductive coupling;

FIG. 6 is a schematic diagram of a variable integrated inductor;

FIG. 7 is a schematic diagram of the windings and current direction of a variable integrated inductor;

FIG. 8 is a typical magnetization curve of a magnetic material;

FIG. 9 is a hysteresis loop with polarized DC bias field magnetization;

FIG. 10 is a flow chart of a design of a variable inductance magnetic integrated coupling mechanism in an embodiment;

FIG. 11 is a front view of the structure and dimensions of the magnetically integrated coupling mechanism;

FIG. 12 is a structural top view and dimensions of a magnetically integrated coupling mechanism;

FIG. 13 is a graph of coupling coefficient k versus gap distance g;

FIG. 14 shows the number of turns n _dc1 And number of turns n _dc2 The mutual inductance difference between the secondary side coil and the control coil is influenced;

FIG. 15 is a composite mutual inductance design area in an integrated control coil;

FIG. 16 is a diagram showing the self inductance of an integrated compensating inductor and its mutual inductance with a receiver coil in different turns combinations;

fig. 17 integrates the self inductance of the compensating inductor and its mutual inductance with the receiving coil with the same number of turns inside and outside;

FIG. 18 is a graph showing the effect of different DC currents in an integrated control coil on the flux density mode;

FIG. 19 is an inverter output waveform for the system at different receive coil positions;

fig. 20 is a graph of simulated output power and efficiency as it moves along the transmit coil with a lateral offset of 10 cm.

Reference numerals: 1-energy transmitting coil, 11-primary side compensation coil, 2-energy receiving coil, 21-AC inductance coil, 22-DC control winding.

Detailed Description

Embodiments of the technical scheme of the present invention will be described in detail below with reference to the accompanying drawings. The following examples are only for more clearly illustrating the technical aspects of the present invention, and thus are merely examples, and are not intended to limit the scope of the present invention.

It is noted that unless otherwise indicated, technical or scientific terms used herein should be given the ordinary meaning as understood by one of ordinary skill in the art to which this invention pertains.

As shown in fig. 2, this embodiment provides a dynamic wireless charging system for an electric automobile, including an electric energy transmitting end and an electric energy receiving end, the electric energy transmitting end adopts multiple sections of energy transmitting guide rails to set up side by side one by one, each section of energy transmitting guide rail includes an energy transmitting coil 1 wound along a first direction and a primary side compensation coil 11 connected in series with the energy transmitting coil 1 and wound along a direction opposite to the first direction, the electric energy receiving end includes an energy receiving coil 2 and a variable integrated compensation inductance, the variable integrated compensation inductance includes an ac inductance coil 21 serving as a secondary side compensation network and two dc control windings 22 for changing the equivalent self inductance of the ac inductance coil 21, and the two dc control windings 22 are respectively set on the inner side and the outer side of the ac inductance coil 21 according to opposite winding directions.

In specific implementation, a horizontal distance is set between two adjacent sections of energy emission guide rails according to a first preset value, the energy emission coil 1 is wound horizontally according to a rectangular coil, the primary compensation coil 11 is wound horizontally according to the same plane inside the energy emission coil 1, the energy receiving coil 2 is wound horizontally according to the rectangular coil, the width of the outer contour of the energy receiving coil 2 is the same as the width of the energy emission coil 1, and the alternating current induction coil 21 and the two direct current control windings 22 in the variable integrated compensation inductor are wound horizontally according to the same plane inside the energy receiving coil 2. A primary magnetic core plate is typically provided on the bottom surface of the energy transmitting coil 1 and the primary compensation coil 11, and a secondary magnetic core plate and a shield plate are provided on the top surface of the energy receiving coil 2 and the variable integrated compensation inductance.

In this embodiment, the power transmitting end is provided with a primary LCC resonance compensation network, and the power transmitting coil is further provided with a primary series resonance capacitor.

In combination with the above description, the system circuit structure is shown in FIG. 3, in which U _in Is the AC output voltage of the inverter, R _L Is a load resistor, coil L _A1 And L _A2 The energy transmitting coil 1 and the primary compensating coil 11 in the corresponding first section of energy transmitting guide rail are connected in reverse series as a first transmitter; coil L _B1 And L _B2 The energy transmitting coil 1 and the primary compensating coil 11 in the second section of energy transmitting guide rail are correspondingly connected in reverse series as a second transmitter; coil L _s And L _f3 The integrated energy receiving coil 2 and the variable integrated compensation inductor are correspondingly integrated as a receiver, and the integrated coils and the main coupling coil share the same ferrite and the shielding plate. L (L) ₁ And L ₂ Is compensation inductance, C ₁ 、C ₂ 、C _f1 、C _f2 、C _p1 、C _p2 、C _s 、C _f3 Is compensation capacitance, C _o Is a filter capacitor. As can be seen from FIG. 3, there are 15 important coupling inductances in the system, M _A1B2 、M _A1B1 、M _A2B2 、M _A2B1 、M _A1A2 、M _B1B2 、M _A1f3 、M _A2f3 、M _B1f3 、M _B2f3 、M _A1s 、M _A2s 、M _B1s 、M _B2s And M _sf3 . Wherein M is _A1B2 、M _A1B1 、M _A2B2 、M _A2B1 、M _A1A2 、M _B1B2 Is the mutual inductance between the two sets of primary side coils. M is M _A1f3 、M _A2f3 、M _B1f3 、M _B2f3 、M _A1s 、M _A2s 、M _B1s 、M _B2s Is the mutual inductance between the secondary side coil and the primary side coil. M is M _sf3 The mutual inductance between the energy receiving coil 2 and the alternating current inductance coil 21 in the variable integrated compensation inductance is compensated for. I _A And I _B Is the current of the transmitting coil, I _s Is the current of the receiving coil. I ₁ And I ₂ Inverter output currents of two compensation networks respectively, I ₃ Is the load output current and ω is the system operating frequency.

For ease of understanding and description, the mutual inductance involved in the system is defined as:

M _A1A2 and M _B1B2 Is the internal mutual inductance directly in series in the circuit, and is usually fixed. They can therefore be directly equivalent to the elements in a circuit as follows:

at the power receiving end, the internal mutual inductance between the power receiving coil and the integrated inductance coil is decoupled through an equivalent T-shaped network, as shown in fig. 4, M _sf3 The resulting T-network affects the circuit impedance on the secondary side, which also provides a way to increase the output power.

L _f1 And L _f2 Equivalent inductances representing the series branches are respectively expressed as:

the resonant network is configured to:

furthermore, a simplified circuit considering the coupling between the same-side coils is shown in fig. 5.

The KVL equation is obtained:

by solving the equation, the values of the transmitting coil current and the output current are calculated as follows:

according to equation (6), neither the energy-transmitting coil current nor the output current contains a load R _e Thus, constant current output characteristics of the primary side and secondary side circuits are maintained.

In view of symmetry of system parameters, the corresponding parameters are set as:

L _f1 ＝L _f2 ＝L _f (7)

for the sake of simplicity of expression, the cross-inductance is expressed as:

the output power is expressed as:

the Chinese invention patent is submitted in the early stage according to the team: 202111524549.7, the system design and optimization scheme can realize stable M in the running process of the electric automobile _ABs The fluctuation of output power during the system rail switching without lateral offset is effectively suppressed.

As can be seen from the analysis result of FIG. 1, M is calculated during the movement of the electric vehicle in the lateral offset state _ABs Remain relatively stable at lower values. According to formula (9), M can be adjusted _sf3 To increase power transfer capability and thereby suppress overall output power sag under lateral misalignment.

However, based on analysis of LCC-LCC wireless energy transfer system with magnetically integrated coupling mechanism, M _sf3 The adjustment of (c) necessarily affects the resonance state. The total output current of the inverter calculated according to equation (5) is expressed as:

wherein L is _ses ＝L _se +M _sf3 ,L _sed ＝L _se -L _f3 The total input impedance of the system is expressed as:

/>

the phase angle between the inverter output voltage and current, denoted as γ, may be used to reflect the resonant state of the system. Positive and negative values of γ represent a perceptual offset and a capacitive offset, respectively. Further, the absolute value thereof indicates the degree to which the system deviates from the resonance point. The tangent value of γ is expressed as:

M _sf3 and L _f3 Is a parameter introduced by the integrated inductor and the effect of these two parameters on the output power and ZVS conditions requires further in-depth analysis. M is M _sf3 The variation of (a) is defined as ΔM _sf3 ，L _f3 The variation of (1) is defined as DeltaL _f3 Formula (9) is rewritten as:

according to equation (13), the output power is subjected to ΔM _sf3 The effect, it can be used as a control variable. At the same time, the output power is not affected by DeltaL _f3 The effect, this is consistent with the constant output current characteristics of the secondary side.

Formula (12) is rewritten as:

wherein, the liquid crystal display device comprises a liquid crystal display device,

in formula (14), T ₁ Is the original ZVS configuration part, T ₂ Is formed by DeltaM _sf3 And DeltaL _f3 The resulting additional parts, which together constitute the tangent of the input impedance angle of the system, determine the resonance state and ZVS conditions. In order to obtain a relatively stable and optimized ZVS operating state, T, during driving of an electric vehicle on a road ₂ Desired T relative to configuration ₁ Is small enough to be negligible. Thus, by careful proper design, the output power at lateral offset can be increased and the system can be operated at reasonable ZVS conditions.

For the variable integrated compensation inductor, in implementation, as shown in fig. 6 and 7, the integrated inductor winding N _ac Wound in the middle, two DC control windings N _dc1 And N _dc2 And the inner side and the outer side are respectively wound, and direct current power supplies are loaded on the two direct current control windings, wherein the directions of magnetic fields generated by currents flowing through the inner direct current control winding and the outer direct current control winding are opposite.

Since the magnetic flux generated by the ac winding flows through the dc windings, an undesirable voltage is generated in both dc windings. Due to the non-linear B-H behavior, ideal decoupling between the inductor and the control winding is difficult. But the ac voltage induced in the dc winding can be relieved by the following measures.

(1) The two direct current windings are respectively adjacent to the inner edge and the outer edge of the alternating current winding, so that the two direct current windings and the alternating current winding have tight coupling characteristics. Thus, the control of the direct current will be more efficient.

(2) The two direct current windings are in reverse series. In the intermediate winding, the induced voltages are in opposite directions, taking into account the direction of the current, which causes the voltages on the direct current coils to cancel each other out.

The magnetic core can increase the inductance of the coil, and the magnetic permeability of the magnetic core influences the inductance value. For an integrated inductor, the magnetic field density is labeled B and the magnetic field strength is labeled H. When alternating current passes through the coil, the magnetization process of the core can be expressed as a relationship of H and B, as shown in fig. 8. The process includes an unsaturated zone, a transition zone, and a saturated zone. At the initial time, the values of H and B are both zero. In the unsaturated region, H and B have strong positive correlation and nonlinear relationship. In the transition region, the increase in B is relatively slow. At the saturation region, B tends to stabilize. The slope of the relationship between H and B reflects the speed of magnetizing the core, corresponding to the equivalent relative permeability μ of the core material.

The general definition of inductance is the ratio of the total flux linkage to the current through the winding, expressed as:

in the expression, ψ (t) is the total flux linkage. Φ is the magnetic flux, which is positively correlated with the relative permeability of the core material. N is the number of turns of the wire and i is the current in the winding. In conjunction with fig. 9 and equation (15), the self-inductance of the variable integrated inductor is positively correlated with the equivalent relative permeability of the core material.

The relative permeability of the core is typically constant during operation of the wireless power transfer system. In particular, the polarization bias generated by the direct current affects the equivalent relative permeability of the core. The hysteresis loop of the polarized dc bias field magnetization is shown in fig. 9. When the core is magnetized by a polarized dc bias field, the ac magnetization produces a small lancet-shaped hysteresis loop that decreases in alignment with the decreasing ac H field. The area of the loop is typically small and its average slope may reflect the incremental relative permeability Δμ:

in combination with the definition of inductance and the influence of direct current on the magnetization process of the core, it is therefore possible to adjust the inductance by means of the permeability of the inductive core.

In combination with the above description, the present embodiment also provides a design flow of the magnetic integrated coupling mechanism, and in the implementation process, in order to reduce the influence of the addition of the outer control coil on the self inductance of the receiving coil and amplify the influence of the dc magnetic field, the distance between the outer control coil and the energy receiving coil should be small enough. In addition, in order to make the total induced voltage on the control coil as small as possible, the size of the inner control coil should be as close as possible to the outer control coil, and thus the distance between the inner control coil and the receiving coil should be small enough. In view of the above, the proposed design flow of the magnetic coupling mechanism is shown in fig. 10.

Firstly, an energy receiving coil is designed according to application scene requirements and related standards, and an energy transmitting coil is designed by comprehensively considering a coupling coefficient k and practical application conditions.

Secondly, in the case of determining the primary coupling coil, the primary-side integrated primary-side compensation coil is designed to obtain a stable M of different receiver positions _ABs 。

Third, on the basis of the energy transmitting coil and the primary side compensation coil, a secondary side compensation inductance coil is designed to ensure M of different receiver positions _ABf ≤εM _ABs 。

Fourth, the relation of the number of turns inside and outside the control coil is reasonably set, so that the total induction voltage is small enough, and a series of turns selection areas of the inside and outside control coil are obtained through a traversal method.

Fifth, the mutual inductance M is determined according to the mutual inductance drop and the expression of the output power during the transverse shift _sf3 And the number of turns of the control coil is designed accordingly.

Sixth, according to L _f3 And M _sf3 The relationship between the changes, ZVS conditions are evaluated according to equation (14).

Finally, on the basis of simulation design of the main coupling coil and the integrated coil, a magnetic coupling mechanism model is built in the Maxwell to obtain system parameters.

The overall system architecture using Maxwell modeling is shown in fig. 11 and 12, and parameters describing the dimensions of the coupling mechanism are defined in table 1. A typical magnetic coupling mechanism of 4.5kW power rating was thus established to verify the design methodology.

Table 1 parameter definition of variable inductance magnetic integration coupling mechanism

For the design of the initial magnetic coupling mechanism, the receiver structure is designed according to the SAE J2954 WPT3Z3 VA standard, the energy receiving coil is designed to be 400mm by 5mm, and the number of turns n _w3 Designed as 10. Ferrite plates and aluminum shielding plates are used for magnetic field enhancement and electromagnetic shielding, respectively. The secondary side ferrite plate has dimensions of "400mm x 10mm", the aluminum shield has dimensions of "420mm x 5mm", and the transmission distance is d=150 mm.

Typically, coupling coefficient and cost are two main indicators of the design of the primary magnetic coupling mechanism, in the present system, coupling coefficient k is closely related to power transfer performance. For a length of road, a shorter length transmission coil requires more resonant compensation networks, magnetic cores, and more costly Litz wire. Taking a transmitting coil as an example, the length, the cost, the magnetic leakage and the coupling coefficient of the electric automobile are fully considered, and an initial magnetic coupling mechanism is designed. In this case, the transmitting coil is dimensioned as "900mm 450mm 5mm", with a number of turns n _w1 Designed as 6.

In the present system, cross-coupling between two adjacent transmit coils will cause the active coils to be affected by nearby inactive coils. In addition, excessive cross coupling may make the series resonant compensation capacitor difficult to configure in practical applications. The larger the gap distance, the larger the mutual inductance drop, and the larger the output power fluctuation. To avoid excessive cross coupling, the gap distance g between two adjacent transmit coils should be optimized. The relationship between the coupling coefficient k and the gap distance g is shown in fig. 13. The results showed that after the gap distance reached 30mm, the k value was negligible and slowly decreased, so the gap distance g was set to 30mm.

The integrated reverse coil is sized as "650mm by 190mm by 5mm", with n turns _w2 Designed as 3. The results show that the output power fluctuation is kept within + -4% over the entire range of movement. Therefore, without lateral offset, power fluctuations during transmit coil transitions can be effectively suppressed. In addition, a secondary side integrated compensation inductance coil is further designed to optimize ZVS conditions, the integrated compensation inductance coil is designed to be 170mm by 5mm, and the number of turns n is n _w4 Designed as 6.

For integrated control coils, the addition of integrated control coils inevitably brings about a coupling with other monopole coils in the coupling mechanism, which coupling should be impaired according to theoretical analysis. Since the mutual inductance between it and the primary side coil is sufficiently small, only the mutual inductance between it and the secondary side coil is considered in the design.

According to the newman equation, the mutual inductance between the transmit coil and the receive coil can be expressed as:

wherein N is _a 、N _b 、

The number of turns of the coils is respectively the infinitesimal of the transmitting coil and the receiving coil. r is (r) _ab Is->

And->

Between (a) and (b)Distance. Mu (mu) ₀ Is the permeability of the vacuum.

For fixed position integrated control coils, the number of turns is a determining factor affecting the mutual inductance, according to equation (17). Therefore, by traversing the number of turns of the inner and outer control coils, the mutual inductance between the integrated control coil and the secondary side coil in different turn combination configurations is obtained. The mutual inductance between the integrated compensation inductance coil and the integrated control coil is M respectively _d1f3 And M _d2f3 . The mutual inductance between the receiving coil and the integrated control coil is M respectively _d1 And M _d2 . The difference in equivalent mutual inductance of the integrated compensation inductance coil and the integrated control coil is expressed as DeltaM _df3 . The equivalent mutual inductance difference between the receiving coil and the integrated control coil is expressed as delta M _ds 。

As can be seen in conjunction with FIG. 14, the number of turns n is plotted in FIGS. 14 (a) and 14 (b) _dc1 And number of turns n _dc2 The mutual inductance difference between the affected secondary side coil and the control coil. Due to I _s And I ₃ There is a 90 degree phase difference between the two, and the total equivalent induced voltage generated by the receiving coil and the integrated inductor coil can be regarded as the maximum value of the two. In a prototype design at a power level of 4.5kW, to ensure a small induced voltage, the composite mutual inductance difference maximum (Δm _df3 ，ΔM _ds ) Can be set below 6 muH. To determine the selection range of the number of turns of the inner and outer integrated control coils, the composite mutual inductance in the integrated control coils is obtained by superposition and plotted in fig. 15. In fig. 15, the region smaller than 6 μh is marked as a design range of the number of turns.

In theory, the variable integrated inductor can be tuned to any number of turns within the design range, except where the desired dc current range is different. To achieve control of the output power at a small control current, the turns configuration is further optimized by simulation. According to FIG. 15, in the design range, when the number of external turns n _d1 When=3, the optional internal turns n _d2 Is relatively large in range. Thus, n is first analyzed _d1 The system characteristics of the inner and outer turns at=3, the self inductance of the integrated compensation inductance coil and the mutual inductance between the integrated compensation inductance coil and the receiving coil at different turns combinations are plotted in fig. 16. In the figure, the upper curve represents L _f3 The lower curve represents M _sf3 The results show two features. First, in a design-wide turn configuration, L _f3 And M _sf3 Effectively varying with increasing direct current. Second, relative to n _d1 ，n _d2 The smaller the corresponding L _f3 And M _sf3 The more significantly it is affected by the direct current. With the increase of the direct current, L is expected in consideration of design purposes _f3 Less variation, M _sf3 The variation is large. Therefore, in order to realize the compromise between self inductance and mutual inductance, n is set in the design scope _d1 And n _d2 Are set equal.

The self inductance of the integrated compensating inductor and its mutual inductance with the receiving coil with the same number of turns on the inside and outside is shown in fig. 17. The results indicate that different turns configurations correspond to different M _sf3 Adjusting the range. In the prototype design of this example 4.5kW power level, n is chosen based on the mutual inductance dip at a lateral offset y=10 cm in fig. 16 _d1 =5 and n _d2 ＝5。

In order to further verify the influence of the polarization bias generated by the direct current on the equivalent relative magnetic permeability of the magnetic core, finite element simulation analysis is carried out on the magnetic flux density mode distribution of the integrated control coils with the direct currents with different magnitudes. The simulated flux density pattern is shown in fig. 18 based on the system parameters of the previous design.

According to the simulation result in fig. 18, the direct current in the integrated control coil can effectively change the magnetic flux density mode of the integrated inductance coil, and change the relative permeability of the magnetic core material, thereby changing the self-inductance value of the integrated compensation inductance coil. Therefore, output power fluctuation caused by guide rail switching in a transverse offset state in the dynamic wireless charging system of the electric automobile can be effectively restrained.

To further verify the correctness of the theoretical analysis, a 4.5kW power level EVDWC system prototype was constructed below. The feasibility of the method is verified through simulation and experiments.

TABLE 2 simulation parameters and calculated resonance parameters for coupling mechanism

A system circuit simulation model was built according to the simulation parameters in table 2, with the inverter output voltage and the load output voltage set to 400V. The lateral offset is set to y=10cm, and 10A of direct current is applied to the integrated control coil. The inverter output waveforms at the different receiver coil positions are shown in fig. 19, where v _inv Is the inverter output voltage, i _invA And i _invB Is the inverter output current corresponding to two transmitters, i _inv Is i _invA And i _invB Is added to the total current. The output voltage phase of the inverter slightly advances the total current, and ZVS conditions under normal parameter configuration are ensured while limiting reactive power circulating in the resonant circuit to reduce power loss.

The results of the simulation of the output power and efficiency for the different receive coil positions at a lateral offset of 10cm are shown in fig. 20. The results show that the efficiency remains high at different receiving positions. And in the moving process of the receiving coil, the whole output power is improved, and the power fluctuation under the transverse offset is restrained. The output power fluctuation is suppressed to within + -5% with respect to the average power.

Based on the theoretical analysis and simulation verification, the embodiment also provides a method for suppressing the lateral offset power fluctuation of the dynamic wireless charging system of the electric automobile, which comprises the following steps:

s1: detecting the transverse offset of the electric automobile;

s2: determining the falling condition of a first mutual inductance coefficient according to the transverse offset, wherein the first mutual inductance coefficient is the mutual inductance coefficient between two adjacent energy transmitting coils and an energy receiving coil; in specific implementation, according to system parameter configuration, a mapping relation between the transverse offset and the first mutual inductance is established in advance through simulation

S3: and the second mutual inductance is changed by adjusting the driving currents of the two direct current control windings, so that the output power of the system is maintained within a preset range, and the second mutual inductance is the mutual inductance between the energy receiving coil and the variable integrated compensation inductor. And in the specific implementation, the mapping relation between the driving currents of the two direct-current control windings and the second mutual inductance coefficient is established in advance through simulation.

In step S3, can be according to

Determining the change relation between the first mutual inductance coefficient and the second mutual inductance coefficient, and when the electric automobile transversely deviates to cause the first mutual inductance coefficient M _ABs When the current is changed, the second mutual inductance M is changed by adjusting the driving current of the two direct current control windings _sf3 So that the output power P _out Maintaining a predetermined range.

By adopting the method, the compensation of the system output power drop can be realized by using smaller direct current.

Finally, it should be noted that the above embodiments are only for illustrating the technical solution of the present invention, and are not limiting; although the invention has been described in detail with reference to the foregoing embodiments, it will be understood by those of ordinary skill in the art that: the technical scheme described in the foregoing embodiments can be modified or some or all of the technical features thereof can be replaced by equivalents; such modifications and substitutions do not depart from the spirit of the invention and are intended to be within the scope of the appended claims and description.

Claims

1. The system comprises an electric energy transmitting end and an electric energy receiving end, wherein the electric energy transmitting end adopts a plurality of sections of energy transmitting guide rails which are arranged side by side one by one, each section of energy transmitting guide rail comprises an energy transmitting coil wound along a first direction and a primary side compensation coil which is connected in series with the energy transmitting coil and wound along the opposite direction of the first direction, the electric energy receiving end comprises an energy receiving coil and a variable integrated compensation inductance, the variable integrated compensation inductance comprises an alternating current induction coil serving as a secondary side compensation network and two direct current control windings for changing the equivalent self inductance of the alternating current induction coil, and the two direct current control windings are respectively arranged on the inner side and the outer side of the alternating current induction coil according to opposite winding directions; the method for suppressing the transverse offset power fluctuation of the system is characterized by comprising the following steps of:

s1: detecting the transverse offset of the electric automobile;

s3: changing a second mutual inductance coefficient by adjusting driving currents of the two direct-current control windings, so that the output power of the system is maintained within a preset range, wherein the second mutual inductance coefficient is a mutual inductance coefficient between the energy receiving coil and the variable integrated compensation inductor;

in step S3 according to

To determine the variation relation between the first mutual inductance and the second mutual inductance, wherein P _out For output power, M _ABs As the first mutual inductance coefficient, M _sf3 For the second mutual inductance coefficient, omega is the system working angular frequency, L _f For compensating inductance value, L in primary side LCC resonance compensation network _se R is equivalent inductance value of secondary side resonance network _e For equivalent load +.>

Phasor expression of inverter alternating current output voltage, when electric automobile transversely deviates to cause first mutual inductance coefficient M _ABs When the current is changed, the second mutual inductance M is changed by adjusting the driving current of the two direct current control windings _sf3 So that the saidOutput power P _out Maintaining a predetermined range.

2. The method for suppressing lateral offset power fluctuation of an electric vehicle dynamic wireless charging system according to claim 1, wherein the method comprises the steps of: and according to system parameter configuration, a mapping relation between the transverse offset and the first mutual inductance is established in advance through simulation, and meanwhile, a mapping relation between the driving currents of the two direct-current control windings and the second mutual inductance is established in advance through simulation.