CN115208317A - Ultra-low phase noise microstrip oscillator - Google Patents

Ultra-low phase noise microstrip oscillator Download PDF

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CN115208317A
CN115208317A CN202210837286.3A CN202210837286A CN115208317A CN 115208317 A CN115208317 A CN 115208317A CN 202210837286 A CN202210837286 A CN 202210837286A CN 115208317 A CN115208317 A CN 115208317A
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terminal short
short
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肖飞
邵兰淳
杨汇峰
肖礼康
何俊岭
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University of Electronic Science and Technology of China
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B1/00—Details
    • H03B1/04—Reducing undesired oscillations, e.g. harmonics
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
    • H03B5/18—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising distributed inductance and capacitance

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Abstract

The invention provides a microstrip oscillator with ultra-low phase noise, which introduces a quasi-band-pass frequency selection network. The quasi-band-pass frequency-selecting network is directly coupled by two quarter-wave resonators through source/load, and can realize quasi-second-order generalized Chebyshev band-pass frequency response. Each transmission zero is arranged on each side of the passband, so that the frequency selectivity can be effectively improved, and higher group delay is formed. In addition, due to the special structure of the feeder line, three transmission zeros are additionally introduced, and the out-of-band rejection performance is greatly improved. The test results of the examples show that: the oscillation frequency is 1.976GHz, the output power is 9.61dBm, the phase noise is-129.85 dBc/Hz @100kHz, the out-of-band second harmonic suppression reaches 48.21dBc, and the quality factor of the oscillator at the position deviated from the carrier wave by 100kHz is-200.66 dBc/Hz.

Description

Microstrip oscillator with ultra-low phase noise
Technical Field
The invention belongs to the technical field of communication, and particularly relates to a microstrip oscillator with ultralow phase noise.
Background
In recent years, with the rapid development of personal mobile communication, the microwave and wireless markets have attracted attention. The microwave oscillator is an indispensable component of a frequency generation source, serves as a key module of circuits such as a phase-locked loop, frequency synthesis and clock recovery, and is widely applied to electronic systems such as mobile phones, satellite communication terminals, mechanisms, radars, missile guidance systems, military communication systems, digital wireless communication, optical multiplexers and optical transmitters. The phase noise of the microwave oscillator, which is used as a reference source for various frequency sources and a key device for generating a time frequency reference, is becoming a key factor limiting the performance of various circuits and systems, and has a decisive influence on the performance, size, weight and cost of electronic systems, which is a difficulty in the design and integration of microwave circuits. Therefore, it is of great importance to study a microwave oscillator having low phase noise.
Disclosure of Invention
The invention aims to overcome the defect of poor phase noise of the conventional microwave oscillator and provides a micro-strip oscillator with ultra-low phase noise. The microwave oscillator is based on a microstrip structure, and has the advantages of low phase noise, high output power, good harmonic suppression, easiness in processing, low cost and the like.
The microstrip oscillator of the invention is shown in figure 1, and is characterized in that: the base terminal of the transistor (BJT) is connected with a first wire (T1), and the collector terminal is connected with a second wire (T2); a direct current power supply is loaded on a bias line section (PT), the bias line section (PT) is connected with a grounding capacitor (SC) and is connected to a first line section (T1) through a resistor (R), and the bias line section (PT) is simultaneously connected to a second line section (T2) to realize the power supply of a transistor (BJT); the left end of the first node (T1) is connected with the right end of the first capacitor (C1); the left end of the first capacitor (C1) is connected with the right end of the first terminal short-circuit line section (ST 1), and the first terminal short-circuit line section (ST 1) is grounded through the first metalized through hole (S1); a first open-circuit branch section (OT 1) is loaded at one point in the middle of a first terminal short-circuit line section (ST 1); one end of the first terminal short-circuit section (ST 1) is in gap coupling with the first terminal short-circuit resonator (SR 1), and the first terminal short-circuit resonator (SR 1) is grounded through the second metalized through hole (S2); the right end of the second wire joint (T2) is connected with the left end of a second capacitor (C2); the right end of the second capacitor (C2) is connected with the left end of a second terminal short-circuit line section (ST 2), and the second terminal short-circuit line section (ST 2) is grounded through a third metalized through hole (S3); a second open-circuit branch section (OT 2) is loaded at one point in the middle of a second terminal short-circuit line section (ST 2); one end of a second terminal short-circuit line section (ST 2) is in gap coupling with a second terminal short-circuit resonator (SR 2), and the second terminal short-circuit resonator (SR 2) is grounded through a fourth metalized through hole (S4); the first terminal short-circuit link (ST 1) is coupled to the second terminal short-circuit link (ST 2) by means of a crossing finger structure (CF); the first terminal short-circuit resonator (SR 1) and the second terminal short-circuit resonator (SR 2) are in gap coupling; energy is output through an output branch (OUT) loaded at a middle point of the second terminal short-circuit line section (ST 2).
The partial structure of the microstrip oscillator of the invention is shown in fig. 2 and is composed of the following components: the first terminal short circuit section (ST 1) is grounded through a first metalized through hole (S1); one end of the first terminal short-circuit section (ST 1) is in gap coupling with the first terminal short-circuit resonator (SR 1), and the first terminal short-circuit resonator (SR 1) is grounded through the second metalized through hole (S2); the second terminal short circuit section (ST 2) is grounded through a third metalized through hole (S3); one end of a second terminal short-circuit section (ST 2) is in gap coupling with a second terminal short-circuit resonator (SR 2), and the second terminal short-circuit resonator (SR 2) is grounded through a fourth metallized through hole (S4); the first terminal short-circuit link (ST 1) is coupled to the second terminal short-circuit link (ST 2) by means of a crossing finger structure (CF); the first terminal short-circuit resonator (SR 1) and the second terminal short-circuit resonator (SR 2) are in gap coupling; for the sake of simplicity, this partial structure is referred to as a quasi-bandpass frequency-selective network. The quasi-bandpass frequency-selecting network directly determines the performance of the microstrip oscillator.
The structural parameters of the quasi-band-pass frequency-selecting network are labeled as shown in fig. 3. Except for the cross finger structure (CF), the rest of the quasi band-pass frequency-selecting network is left-right symmetrical about a vertical plane. Wherein l 1 The line length of the first terminal short-circuit resonator (SR 1) and the second terminal short-circuit resonator (SR 2) in gap coupling is shown; l. the 2 Represents the line length of the first short-circuited termination resonator (SR 1) and also represents the line length of the second short-circuited termination resonator (SR 2); l 3 A line length representing a crossed finger structure (CF); l 4 Represents the line length of the second terminal short-circuit section (ST 2) from the end to the coupling position with the second terminal short-circuit resonator (SR 2); l 5 Showing the second terminal short-circuited stub (ST 2) and the second terminal short-circuited resonator(SR 2) coupled horizontal section line length; l 6 Represents the wire length of the first terminal short circuit section (ST 1) from the cross finger structure (CF) access position to the first metallized through hole (S1); w is a 1 Represents a line width of the first short-circuited resonator (SR 1); w is a 2 Represents the line width of the second terminal short-circuited section (ST 2); w is a 3 Represents a line width of a cross finger structure (CF); g 1 Represents the gap width between the first terminal short-circuit resonator (SR 1) and the second terminal short-circuit resonator (SR 2); g 2 Represents the width of a gap between one end of a first terminal short-circuit section (ST 1) and a first terminal short-circuit resonator (SR 1); d represents the diameter of the second metallized via (S2).
The quasi-bandpass frequency-selective network can theoretically realize a second-order bandpass frequency response, as shown in fig. 4, with one transmission zero on each side of its passband. The normalized coupling matrix [ M ] corresponding thereto is as follows:
Figure BDA0003749086660000021
wherein, the first terminal short-circuited section (ST 1) is denoted by S, the second terminal short-circuited section (ST 2) is denoted by L, the first terminal short-circuited resonator (SR 1) is denoted by 1, and the second terminal short-circuited section (ST 2) is denoted by 2. m is ij (i and j take S, L, 1 and 2, respectively) represent normalized coupling coefficients between the first terminal short-circuited section (ST 1), the second terminal short-circuited section (ST 2), the first terminal short-circuited resonator (SR 1) and the second terminal short-circuited resonator (SR 2).
By the following formula, the coupling coefficient m can be normalized ij Determining a coupling coefficient k ij 。
Figure BDA0003749086660000022
External quality factor Q e Can be determined from the normalized coupling coefficient by the following formula.
Figure BDA0003749086660000023
Wherein Q eS Represents the external quality factor between the first short-circuited termination (ST 1) and the first short-circuited termination (SR 1). Q eL Represents the external quality factor between the second terminal short-circuit section (ST 2) and the second terminal short-circuit resonator (SR 2). FBW is the relative bandwidth.
And determining the initial structure parameter value of the quasi-band-pass frequency selection network by using a traditional filter synthesis method according to the oscillation frequency requirement of the microstrip oscillator. The initial values of the electrical lengths of the first terminal short-circuit resonator (SR 1) and the second terminal short-circuit section (ST 2) are set to be quarter wavelengths corresponding to the oscillation frequency; other parameter g 1 And g 2 Is determined by comparing the ideal and actual coupling coefficients and the ideal and actual external figures of merit. Based on the initial values, structural parameter values are further adjusted through simulation optimization, the insertion loss of the quasi-band-pass frequency-selecting network is minimized and the group delay is maximized at the oscillation frequency, and final structural parameter values can be determined.
The simulation result of the quasi-band-pass frequency-selective network is shown in fig. 5. Having five transmission zeros at finite frequencies, each at f TZ1 、f TZ2 、f TZ3 、f TZ4 And f TZ5 And (6) labeling. Wherein the two resonators of the first terminal short-circuit resonator (SR 1) and the second terminal short-circuit link (ST 2) are directly coupled by source/load to form f TZ2 And f TZ3 . The two transmission zeros are respectively positioned on each side of the passband, so that the frequency selectivity is effectively improved, and the formation of higher group delay is facilitated. The other three transmission zeros are positioned in the stop band, so that the out-of-band rejection capability is greatly improved. These three transmission zeros are associated with the specific configuration of the first terminal short-circuited section (ST 1) and the second terminal short-circuited section (ST 2), and the equivalent circuit diagram thereof is shown in fig. 6. Input admittance Y in Is shown as
Y in ＝-jY 1 cotθ 1 (4)
Wherein, Y 1 And theta 1 Respectively, characteristic admittance and electrical length. When Y is in Produced when = ∞ timeTransmission zero.
The oscillator has the advantages that: low phase noise, high power output, good out-of-band harmonic suppression, easy processing, low cost, and the like.
Drawings
FIG. 1: a microstrip oscillator schematic;
FIG. 2: a quasi-band-pass frequency-selecting network schematic diagram;
FIG. 3: a quasi-bandpass frequency-selecting network structure parameter schematic diagram;
FIG. 4 is a schematic view of: an ideal second-order generalized Chebyshev band-pass frequency response diagram;
FIG. 5 is a schematic view of: a simulation result diagram of the quasi-band-pass frequency-selecting network;
FIG. 6: an equivalent circuit diagram;
FIG. 7 is a schematic view of: actual coupling coefficient k 12 Following gap g 1 A graph of the variation;
FIG. 8: actual external quality factor Q e Following the gap g 2 A graph of the variation;
FIG. 9: an S parameter simulation result diagram of the first embodiment;
FIG. 10: a simulation result diagram of group delay of the first embodiment;
FIG. 11: structural parameter l of embodiment one 4 A graph of the effect on its performance;
FIG. 12: the influence of the number of nodes N of the cross finger structure (CF) of the first embodiment on the performance of the cross finger structure;
FIG. 13 is a schematic view of: a phase noise test result chart of the second embodiment;
FIG. 14 is a schematic view of: and the output spectrum test result chart of the second embodiment.
Detailed Description
In order to embody the inventive and novel aspects of the present invention, the following description will be made in conjunction with the accompanying drawings and specific examples, but the embodiments of the present invention are not limited thereto.
In the embodiment, a common microstrip substrate with a relative dielectric constant of 3.66 and a thickness of 0.508mm is selected.
The first embodiment is a quasi-bandpass frequency-selecting network. If the oscillation frequency of the microstrip oscillator is set to 2.0GHz, the first embodiment uses a filterThe method is designed to determine the initial values of the structural parameters. Center frequency f of embodiment one 0 =2.0GHz, relative bandwidth FBW =2%. The normalized coupling coefficient matrix of the quasi-band-pass frequency-selecting network is
Figure BDA0003749086660000031
The coupling coefficient matrix can be calculated by the formula (2) as
Figure BDA0003749086660000032
From this, an ideal coupling coefficient k of the first short-circuited resonator (SR 1) and the second short-circuited resonator (SR 2) can be obtained 12 =0.03094. Calculating to obtain the ideal external quality factor Q by the formula (3) e ＝Q eS ＝Q eL ＝20.89。
The actual coupling coefficient k is given in fig. 7 12 Following gap g 1 The curve of the change. Fig. 8 shows the actual external quality factor Q e Following gap g 2 The curve of the change. Using the previously calculated values of the ideal coupling coefficient and the ideal external quality factor, the corresponding g can be determined from these curves 1 And g 2 Value, i.e. g 1 =1.3mm and g 2 =0.1mm as their initial value.
At the center frequency f based on the initial values of the structural parameters obtained previously 0 Adjusting the structural parameter values to minimize the insertion loss and maximize the group delay of the quasi-bandpass frequency-selecting network, and finally determining a group of structural parameter values as (unit: mm): l 1 ＝5.80、l 2 ＝23.58、l 3 ＝2.00、l 4 ＝11.20、l 5 ＝4.80、l 6 ＝31.56、w 1 ＝0.80、w 2 ＝0.60、w 3 ＝0.22、g 1 ＝1.65、g 2 =0.28 and d =0.30. The simulation results of example one are shown in fig. 9 and 10. The minimum insertion loss in the band is 4.51dB, and the group delay peak value is 11.25ns obtained at 2.004 GHz. Five transmission zeros are respectively positioned at 0.54GHz, 1.87GHz, 2.09GHz, 2.63GHz and 3.61GHz at 2f 0 The inhibition at (d) was-28.79 dB.
To illustrate the performance flexibility of embodiment one, a typical structural parameter l is given in FIG. 11 4 Impact on performance of example one. When l is 4 When varied, it affects primarily three transmission zeros, i.e. f TZ1 、f TZ4 And f TZ5 For transmission zero point f TZ2 And f TZ3 There is no effect. This fully illustrates that three transmission zeros f TZ1 、f TZ4 And f TZ5 In connection with the special construction of the first terminal short-circuit link (ST 1) and the second terminal short-circuit link (ST 2). When l is 4 At increasing time f TZ1 、f TZ4 And f TZ5 Are moving in the low frequency direction. Based on the equivalent circuit of FIG. 6, when Y is 1 =0.014S and θ 1 In the case of =0.762rad, the transmission zeros calculated using equation (4) are 2.706GHz and 5.413GHz, respectively, and the fourth transmission zero f obtained by simulation TZ4 =2.63GHz, fifth transmission zero f TZ5 The frequency of the equivalent circuit is effectively verified by the frequency of 3.59GHz, and a generation mechanism of transmission zero points is disclosed.
Fig. 12 shows the effect of the number of nodes N of the cross-finger structure (CF) on the performance of the quasi-bandpass frequency-selective network. When N =0, only f is present TZ1 、f TZ2 And f TZ3 These three transmission zeros. When N =1, four transmission zeros, i.e., f, occur TZ1 、f TZ2 、f TZ3 And f TZ4 . As N continues to increase, the strength of the coupling between the first terminal short-circuited section (ST 1) and the second terminal short-circuited section (ST 2) also increases, at which time the fifth transmission zero point f occurs TZ5 . Therefore, the performance of the quasi-band-pass frequency selection network can be effectively controlled by adjusting N.
Example two is a microstrip oscillator based on example one, and the test results are shown in fig. 13 and 14. When the DC bias voltage is 2.7V and the power supply current is 12mA, the measured oscillation frequency is 1.976GHz, the output power is 9.61dBm, and the phase noise is-129.85 dBc/Hz @100kHz. Meanwhile, the quasi-bandpass frequency-selecting network has good out-of-band performance, and the second harmonic suppression of the second embodiment reaches 48.21dBc. Example two the oscillator quality factor at 100kHz off-carrier is-200.66 dBc/Hz. In the second embodiment, the phase noise, the output power, the second harmonic suppression and the quality factor of the oscillator exceed the similar indexes of other S-band microstrip oscillators published and reported at home and abroad at present, and the method has remarkable technical progress.
The embodiments listed above fully demonstrate that the microstrip oscillator of the present invention has the advantages of low phase noise, high output power, good out-of-band harmonic suppression, easy debugging, etc., and has significant technical progress. It will be appreciated by those of ordinary skill in the art that the embodiments described herein are intended to assist the reader in understanding the principles of the invention and are to be construed as being without limitation to such specifically recited embodiments and examples. Those skilled in the art can make various other specific changes and combinations based on the teachings of the present invention without departing from the spirit of the invention, and these changes and combinations are within the scope of the invention.

Claims (7)

1. A microstrip oscillator, characterized by: the base terminal of the transistor (BJT) is connected with a first wire (T1), and the collector terminal is connected with a second wire (T2); the direct current power supply is loaded on a bias line node (PT), the bias line node (PT) is connected with a grounding capacitor (SC) and is connected to a first node (T1) through a resistor (R), and the bias line node (PT) is simultaneously connected to a second node (T2) to realize the power supply of a transistor (BJT); the left end of the first node (T1) is connected with the right end of the first capacitor (C1); the left end of the first capacitor (C1) is connected with the right end of the first terminal short-circuit line section (ST 1), and the first terminal short-circuit line section (ST 1) is grounded through the first metalized through hole (S1); a first open-circuit branch (OT 1) is loaded at one point in the middle of a first terminal short-circuit link (ST 1); one end of the first terminal short-circuit section (ST 1) is in gap coupling with the first terminal short-circuit resonator (SR 1), and the first terminal short-circuit resonator (SR 1) is grounded through the second metalized through hole (S2); the right end of the second wire joint (T2) is connected with the left end of a second capacitor (C2); the right end of the second capacitor (C2) is connected with the left end of the second terminal short-circuit line section (ST 2), and the second terminal short-circuit line section (ST 2) is grounded through a third metalized through hole (S3); a second open-circuit branch section (OT 2) is loaded at one point in the middle of a second terminal short-circuit line section (ST 2); one end of a second terminal short-circuit section (ST 2) is in gap coupling with a second terminal short-circuit resonator (SR 2), and the second terminal short-circuit resonator (SR 2) is grounded through a fourth metallized through hole (S4); the first terminal short-circuit link (ST 1) is coupled to the second terminal short-circuit link (ST 2) by means of a crossing finger structure (CF); the first terminal short-circuit resonator (SR 1) and the second terminal short-circuit resonator (SR 2) are in gap coupling; energy is output through an output branch (OUT) loaded at a middle point of the second terminal short-circuit line section (ST 2).
2. The microstrip oscillator according to claim 1, comprising a quasi-bandpass frequency-selective network configured to: the first terminal short circuit section (ST 1) is grounded through a first metalized through hole (S1); one end of the first terminal short-circuit section (ST 1) is in gap coupling with the first terminal short-circuit resonator (SR 1), and the first terminal short-circuit resonator (SR 1) is grounded through the second metalized through hole (S2); the second terminal short circuit section (ST 2) is grounded through a third metalized through hole (S3); one end of a second terminal short-circuit section (ST 2) is in gap coupling with a second terminal short-circuit resonator (SR 2), and the second terminal short-circuit resonator (SR 2) is grounded through a fourth metallized through hole (S4); the first terminal short-circuit link (ST 1) is coupled to the second terminal short-circuit link (ST 2) by means of a crossing finger structure (CF); the first short-circuited resonator (SR 1) is slot-coupled to the second short-circuited resonator (SR 2).
3. The microstrip oscillator according to claim 1, wherein said quasi-bandpass frequency-selective network theoretically implements a second-order bandpass frequency response, with a transmission zero on each side of its passband; the normalized coupling matrix [ M ] corresponding thereto is as follows:
Figure FDA0003749086650000011
wherein S denotes a first terminal short-circuited section (ST 1), L denotes a second terminal short-circuited section (ST 2), 1 denotes a first terminal short-circuited resonator (SR 1), and 2 denotes a second terminal short-circuited resonator (SR 1)And a wire knot (ST 2). m is ij (i and j take S, L, 1 and 2 respectively) to represent normalized coupling coefficients among the first terminal short-circuit section (ST 1), the second terminal short-circuit section (ST 2), the first terminal short-circuit resonator (SR 1) and the second terminal short-circuit resonator (SR 2);
by the following formula, the coupling coefficient m can be normalized ij Determining a coupling coefficient k ij ；
Figure FDA0003749086650000021
External quality factor Q e Can be determined from the normalized coupling coefficient by the following equation:
Figure FDA0003749086650000022
wherein Q is eS Represents the external quality factor, Q, between the first short-circuited termination (ST 1) and the first short-circuited resonator (SR 1) eL Represents the external quality factor between the second terminal short-circuited stub (ST 2) and the second terminal short-circuited resonator (SR 2), FBW being the relative bandwidth.
4. The microstrip oscillator according to claim 1, comprising a quasi band-pass frequency-selective network having five transmission zeros at finite frequencies, respectively denoted by f TZ1 、f TZ2 、f TZ3 、f TZ4 And f TZ5 Labeling is carried out; wherein the two resonators of the first terminal short-circuit resonator (SR 1) and the second terminal short-circuit node (ST 2) are directly coupled by source/load to form f TZ2 And f TZ3 (ii) a The two transmission zeros are respectively positioned at each side of the passband, so that the frequency selectivity is effectively improved, and higher group delay is formed; the other three transmission zero points are positioned in the stop band, so that the out-of-band inhibition capability is greatly improved; these three transmission zeros are associated with the particular configuration of the first (ST 1) and second (ST 2) terminal short circuit nodes; input admittance Y of its equivalent circuit in Is shown as
Y in ＝-jY 1 cotθ 1
Wherein, Y 1 And theta 1 Respectively representing characteristic admittance and electrical length; when Y is in Transmission zero occurs when = ∞.
5. The microstrip oscillator according to claim 1, wherein the design method of the quasi-bandpass frequency-selecting network comprises: determining an initial structure parameter value of a quasi-band-pass frequency selection network by using a traditional filter synthesis method according to the oscillation frequency requirement of a microstrip oscillator; the initial values of the electrical lengths of the first terminal short-circuit resonator (SR 1) and the second terminal short-circuit section (ST 2) are set to be quarter wavelengths corresponding to the oscillation frequency; other parameters g 1 And g 2 Is determined by comparing the ideal and actual coupling coefficients and the ideal and actual external figures of merit; based on the initial values, structural parameter values are further adjusted through simulation optimization, the insertion loss of the quasi-band-pass frequency-selecting network is minimized and the group delay is maximized at the oscillation frequency, and final structural parameter values can be determined.
6. The microstrip oscillator according to claim 1, wherein the quasi-bandpass frequency-selecting network has the structural parameters: l 1 A line length l representing the gap coupling of the first short-circuited resonator (SR 1) and the second short-circuited resonator (SR 2) 2 Denotes the line length of the first short-circuited resonator (SR 1) and also denotes the line length of the second short-circuited resonator (SR 2) | 3 Line length, l, representing cross finger structure (CF) 4 Denotes a line length l from the end of the second short-circuited stub (ST 2) to the coupling position with the second short-circuited resonator (SR 2) 5 A horizontal partial line length l representing the coupling of the second terminal short-circuited line segment (ST 2) and the second terminal short-circuited resonator (SR 2) 6 Represents the wire length, w, of the first terminal short-circuit section (ST 1) from the crossover finger structure (CF) access position to the first metalized via (S1) 1 Represents the line width, w, of the first short-circuited resonator (SR 1) 2 Represents the line width, w, of the second terminal short-circuited section (ST 2) 3 Indicating crossed finger structuresLine width of (CF), g 1 Denotes a gap width g between the first and second short-circuited resonators (SR 1, SR 2) 2 The width of a gap between one end of the first terminal short-circuit section (ST 1) and the first terminal short-circuit resonator (SR 1) is shown, and d is the diameter of the second metalized through hole (S2); when l is 1 ＝5.80mm、l 2 ＝23.58mm、l 3 ＝2.00mm、l 4 ＝11.20mm、l 5 ＝4.80mm、l 6 ＝31.56mm、w 1 ＝0.80mm、w 2 ＝0.60mm、w 3 ＝0.22mm、g 1 ＝1.65mm、g 2 =0.28mm and d =0.30mm, the minimum insertion loss in the quasi-band-pass frequency-selecting network band is 4.51dB, the group delay peak value is 11.25ns at 2.004GHz, five transmission zeros are respectively located at 0.54GHz, 1.87GHz, 2.09GHz, 2.63GHz and 3.61GHz, and at 2f 0 The inhibition at (d) was-28.79 dB.
7. The microstrip oscillator according to claim 1, wherein the quasi-bandpass frequency-selecting network has the structural parameters: l. the 1 A line length l representing the gap coupling of the first short-circuited resonator (SR 1) and the second short-circuited resonator (SR 2) 2 The line length of the first short-circuited resonator (SR 1) is shown, and the line length of the second short-circuited resonator (SR 2) is also shown, | 3 Line length, l, representing cross finger structure (CF) 4 Denotes a line length l from the end of the second short-circuited stub (ST 2) to the coupling position with the second short-circuited resonator (SR 2) 5 A horizontal partial line length l representing the coupling of the second terminal short-circuited line segment (ST 2) and the second terminal short-circuited resonator (SR 2) 6 Represents the wire length, w, of the first terminal short-circuit stub (ST 1) from the crossover finger structure (CF) access position to the first metallized via (S1) 1 Represents the line width, w, of the first short-circuited resonator (SR 1) 2 Represents the line width, w, of the second terminal short-circuited section (ST 2) 3 Line width, g, of the interdigitated finger structure (CF) 1 Denotes a gap width g between the first and second short-circuited resonators (SR 1, SR 2) 2 Indicates the width of the gap between one end of the first terminal short-circuited stub (ST 1) and the first terminal short-circuited resonator (SR 1),d represents the diameter of the second metallized via (S2); when l is 1 ＝5.80mm、l 2 ＝23.58mm、l 3 ＝2.00mm、l 4 ＝11.20mm、l 5 ＝4.80mm、l 6 ＝31.56mm、w 1 ＝0.80mm、w 2 ＝0.60mm、w 3 ＝0.22mm、g 1 ＝1.65mm、g 2 =0.28mm and d =0.30mm; when the DC bias voltage is 2.7V and the power current is 12mA, the oscillation frequency of the micro-strip oscillator is 1.976GHz, the output power is 9.61dBm, the phase noise is-129.85 dBc/Hz @100kHz, the out-of-band second harmonic suppression reaches 48.21dBc, and the quality factor of the oscillator at the position deviated from the carrier wave by 100kHz is-200.66 dBc/Hz.
CN202210837286.3A 2022-07-15 2022-07-15 Ultra-low phase noise microstrip oscillator Pending CN115208317A (en)

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