CN115189615A - SVPWM control device of brushless DC motor - Google Patents
SVPWM control device of brushless DC motor Download PDFInfo
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
- H02P25/024—Synchronous motors controlled by supply frequency
- H02P25/026—Synchronous motors controlled by supply frequency thereby detecting the rotor position
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- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
- H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
- H02P2207/055—Surface mounted magnet motors
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P2209/00—Indexing scheme relating to controlling arrangements characterised by the waveform of the supplied voltage or current
- H02P2209/11—Sinusoidal waveform
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Abstract
The invention discloses an SVPWM control device of a brushless DC motor, which comprises a main control chip, a DC voltage and bus current sampling circuit, a switching power supply, a DC power supply, a driving circuit, a main inverter circuit and a Hall position detection circuit, wherein the main control chip is connected with the main control chip; the main control chip judges the space voltage vector U according to the collected signals ref Determining two adjacent basic voltage vectors in the sector, and then determining a space voltage vector U ref The action time and the duty ratio of two adjacent voltage vectors in the sector finally determine a space voltage vector U ref And synthesizing the SVPWM signals and outputting the SVPWM signals to the driving circuit, controlling the main inverter circuit to be switched on and off at corresponding moments, and generating sine waves to control the brushless motor to continuously operate. The invention adopts a sine wave control mode, the waveform is very continuous and smooth, the torque fluctuation is small, and the actual rotation is realizedThe control mode is kept in a mute state, and is more suitable for being applied to harsher occasion environments than a square wave control mode with large fluctuation and noise. The invention has high position detection precision and is more practical.
Description
Technical Field
The invention relates to the technical field of motor control, in particular to an SVPWM control device of a brushless direct current motor.
Background
An electric motor is a machine for converting electric energy into mechanical energy, and a brushed direct current motor is used as energy conversion equipment in the past, but the brushed motor has many defects, such as large loss, large heat generation, low efficiency and the like. With the development of power electronics and other technologies in recent years, a brushless motor gradually becomes a mainstream, a series of operations of phase change and the like can be completed only by selectively conducting an inverter module of the brushless motor by using a power electronic control technology at the phase change moment, the motor service life is not short due to the abrasion of an electric brush and a commutator of the brushless motor, and the brushless direct current motor has the corresponding mechanical characteristics similar to those of a direct current motor under the support of an intelligent control algorithm and the phase change technology, has good performance and high efficiency, and is widely applied to the fields of aerospace, automobiles, household appliances and the like.
The general brushless direct current motor carries out position judgment according to a position sensor arranged on the body of the brushless direct current motor or generates target magnetic fields in six specific directions through conversion of other algorithms, so that the permanent magnet of the rotor magnetic field is close to each other in the specific direction, and the specific direction is changed into the next direction for the next rotation of the permanent magnet of the rotor after the rotor rotates for a certain angle. This control mode we call square wave control. Although the square wave control method has a simple algorithm and can enable the brushless motor to normally rotate, generally, only 6 kinds of generated magnetic fields in fixed directions are used, which causes large torque ripple and large noise in the current commutation process, so that the square wave drive cannot be applied to certain occasions with harsh requirements on the environment. The other is called sine wave drive, the rotor is dragged to rotate by the drive mode that the direction of the generated magnetic field is almost right-angled to the current magnetic field of the rotor, so that high-efficiency drive is achieved, the method is small in fluctuation and noise, and is more suitable for occasions with harsh environments and better in future development prospect.
In recent years, a sinusoidal wave vector control method of a brushless dc motor is under study, and the most important problem to be solved is to improve the position detection accuracy as much as possible under the condition of only adopting a simple hall detector, and a general solution is as follows: the method of indirectly calculating the rotation angle of the motor by measuring time, assuming that the previous hall signal period is the same as the current hall signal period, is inaccurate when the speed is low and the motor is in a speed change state, thereby limiting the practicability.
Therefore, it is an urgent need to solve the problem for those skilled in the art to provide an SVPWM control apparatus and method for a brushless dc motor capable of improving position detection accuracy.
Disclosure of Invention
In view of the deficiencies of the prior art, the present invention aims to provide an SVPWM control apparatus and method for a brushless dc motor.
In order to achieve the purpose, the invention adopts the following technical scheme:
the SVPWM control device of the brushless DC motor comprises a main control chip, a DC voltage and bus current sampling circuit, a switching power supply, a DC power supply, a driving circuit, a main inverter circuit and a Hall position detection circuit; the direct current power supply is respectively connected with the main inverter circuit and the switching power supply; the switching power supply is respectively connected with the direct-current voltage and bus current sampling circuit, the main control chip and the driving circuit; the main control chip is connected with the direct-current voltage and bus current sampling circuit to obtain direct-current voltage and three-phase current signals of the motor; the input end of the driving circuit is connected with the output end of the main control chip, and the output end of the driving circuit is connected with the main inverter circuit; the main inverter circuit and the Hall position detection circuit are respectively connected with the brushless direct current motor; the main control chip is connected with the output end of the Hall position detection circuit to obtain a detection signal of the Hall sensor; the main control chip judges the space voltage vector U according to the collected signals ref The sector is located, so as to determine two adjacent basic voltage vectors in the sector, and then determine a space voltage vector U ref The action time and the duty ratio of two adjacent voltage vectors in the sector finally determine a space voltage vector U ref And synthesizing the SVPWM signals and outputting the SVPWM signals to the driving circuit, and further controlling the main inverter circuit to be switched on and off at corresponding moments to generate sine waves to control the brushless motor to continuously operate.
Preferably, the brushless motor is a 4-pole brushless motor.
Preferably, the brushless motor is a surface-mounted permanent magnet synchronous motor.
Preferably, in the d-q coordinate system, the electromagnetic torque equation of the surface-mounted permanent magnet synchronous motor is as follows:
T e ＝n p ψ f i q
wherein, T e Is an electromagnetic torque, n p Is the pole pair number psi of the motor f Is a permanent magnet flux linkage i q Respectively, the components of the resultant current vector in the q-axis.
Preferably, the SVPWM is a 7-segment SVPWM.
Preferably, the main inverter circuit is a three-phase inverter.
The invention also aims to disclose a sine wave control method of the brushless direct current motor of the device, which comprises the following steps:
(1) In the current loop control, the voltage u under the two-phase static coordinate system of the motor is obtained α And u β ，
(2) According to voltage u α And u β Calculating and judging space voltage vector U ref The sector in which the cell is located;
(3) Determining a space voltage vector U ref Action time T of two adjacent voltage vectors in the located sector x 、T y And a switching time t aon 、t bon 、t con And generating six paths of complementary SVPWM signals and outputting the signals to a driving circuit, and further controlling a main inverter circuit to be switched on and off at corresponding time to generate sine waves to control the brushless motor to continuously operate.
Compared with the prior art, the invention has the following beneficial effects:
(1) The invention adopts a sine wave control mode, the formed waveforms are basically continuous and smooth, the torque fluctuation is very small, the rotation is basically kept in a mute state during the actual rotation, and the invention is more suitable for being applied to harsher occasion environments than a square wave control mode with large fluctuation and noise.
(2) The sector judgment is obtained by converting the relation between phase currents, and compared with the Hall element detection signal, the current position is judged more accurately.
Drawings
Fig. 1 is a schematic circuit diagram of an SVPWM control apparatus for a brushless dc motor according to an embodiment of the present invention;
FIG. 2 is a schematic diagram of a method for mounting a Hall sensor of a brushless motor according to an embodiment of the present invention;
FIG. 3 is a schematic diagram of an equivalent model of a three-phase AC winding according to an embodiment of the present invention;
FIG. 4 is a schematic diagram of an equivalent model of a two-phase AC winding according to an embodiment of the present invention;
FIG. 5 is a schematic diagram of an equivalent model of a rotating DC winding according to an embodiment of the present invention;
FIG. 6 is a schematic diagram of a three-phase inverter topology employed in embodiments of the present invention;
FIG. 7 is a schematic diagram of space voltage vector sector division according to an embodiment of the present invention;
FIG. 8 is a three-phase waveform of a first sector according to an embodiment of the present invention;
FIG. 9 is a flowchart of an SVPWM algorithm implementation procedure according to an embodiment of the present invention;
FIG. 10 shows a surface-mounted permanent magnet synchronous motor in the embodiment of the present invention at i d Vector control block diagram of =0;
FIG. 11 is a schematic diagram of an overall model of a square wave driving system according to an embodiment of the present invention;
FIG. 12 is a schematic diagram of a simulation setup of a pulse generation module according to an embodiment of the present invention;
FIG. 13 is a schematic diagram of an overall model of a sine wave drive system according to an embodiment of the present invention;
FIG. 14 is a diagram of actual phase currents of a brushless motor in an embodiment of the present invention;
FIG. 15 is a three-phase current diagram of a brushless motor at 1000rpm for a simulation experiment in accordance with an embodiment of the present invention;
FIG. 16 is a waveform of a simulation experiment sector decision in an embodiment of the present invention;
FIG. 17 is a waveform diagram of the time distribution of the basic voltage vector in the simulation experiment in the embodiment of the present invention,
FIG. 18 is a graph of a simulated experimental motor torque waveform in an embodiment of the present invention;
FIG. 19 is a waveform diagram of the motor rotation speed of the simulation experiment in the embodiment of the present invention;
fig. 20 is a waveform diagram of motor torque of a simulation experiment square wave control system in a comparative example of the present invention.
Detailed Description
The present invention is further described with reference to the following examples, which should not be construed as limiting the scope of the invention.
Referring to fig. 1, the SVPWM control apparatus of the brushless DC motor includes a main control chip, a DC voltage and bus current sampling circuit, a switching POWER supply, a DC POWER supply (DC POWER), a driving circuit, a main inverter circuit, and a hall position detecting circuit; the direct current power supply is respectively connected with the main inverter circuit and the switching power supply; the switching power supply is respectively connected with the direct-current voltage and bus current sampling circuit, the main control chip and the driving circuit; the main control chip is connected with the direct-current voltage and bus current sampling circuit to obtain direct-current voltage and three-phase current signals of the motor; the input end of the driving circuit is connected with the output end of the main control chip, and the output end of the driving circuit is connected with the main inverter circuit; the main inverter circuit and the Hall position detection circuit are respectively connected with the brushless direct current motor; the main control chip is connected with the output end of the Hall position detection circuit to obtain a detection signal of the Hall sensor; the main control chip judges the space voltage vector U according to the collected signals ref The sector is located, so as to determine two adjacent basic voltage vectors in the sector, and then determine a space voltage vector U ref The action time and the duty ratio of two adjacent voltage vectors in the sector finally determine a space voltage vector U ref And synthesizing the SVPWM signals and outputting the SVPWM signals to a driving circuit, and further controlling a main inverter circuit to be switched on and off at corresponding moments to generate sine waves to control the brushless motor to continuously operate.
The following describes the implementation of the SVPWM control apparatus and the control method for the brushless dc motor in detail:
1. basic structure of brushless DC motor
As shown in fig. 2, the brushless motor of the present embodiment employs three hall sensors (hall 1, hall 2, hall 3) installed around the brushless motor with an electrical angle difference of 120 degrees. Therefore, the three Hall sensors are arranged, and the position of the magnetic field of the motor rotor can be effectively reflected. The brushless motor used in the present embodiment is a 4-pole pair brushless motor, and for an n-pole pair brushless motor, n × mechanical angle = electrical angle, and the mechanical angle is the actual rotation angle of the rotor.
According to the installation position and the principle that the Hall signals jump when the magnetic densities of the Hall sensors are balanced, the position of the current magnetic field can be judged after the three Hall sensor detection signals with the electrical angle difference of 120 degrees are converted. For example, when the signals of hall 1, hall 2 and hall 3 are 100, the magnetic field at this time is in the current sector, and each sector occupies 60 degrees, and if the three hall signals are detected to repeatedly circulate in the order of 100 → 110 → 010 → 011 → 001 → 101 → back to 100, the motor is rotating reversely; similarly, if the hall signals are cycled in the reverse order, this indicates that the motor is rotating in the forward direction. Each cycle represents that the magnetic field has rotated one turn, and the hall signal returns to the hall signal that was six times earlier, representing that the magnetic field direction returned to the original sector.
The three-phase current of the brushless motor is constantly changing, so that for analysis and calculation, on the premise of ensuring the two coordinate transformation basic rules of instant power identity and assignment identity, a three-phase static coordinate system (abc coordinate) of the brushless motor can be equivalently transformed into other coordinate systems which are convenient to analyze and utilize, such as a two-phase static coordinate system (alpha-beta coordinate) and a two-phase rotating coordinate system (dq coordinate), for convenience.
2. Coordinate system equivalent transformation
(1) Three-phase stationary coordinate system
The three-phase stationary coordinate system A, B, C and the three-coordinate axis A, B, C represent the directions of the magnetomotive forces of the three-phase alternating-current windings after being respectively electrified with currents of certain magnitudes. The three-phase windings are spatially separated from each other by 120 degrees of spatial angle, so that in the three-phase stationary coordinate system is a planar coordinate system of three coordinate axes separated from each other by 120 degrees, as shown in fig. 3.
(2) Two-phase stationary coordinate system
If the three-phase winding in the three-phase static coordinate system is input by a symmetrical sine wave, the direction of the synthesized magnetic field is a circle with the center of the circle at the origin in a plan view, and if a two-phase static coordinate system exists, the track on the plane of the coordinate system is the same as the track on the three-phase static coordinate system by input with a certain change, the center of the circle is at the origin, the radius is equal, and the motion angle is consistent, the principle of conversion equivalence can be considered to be met, namely the two coordinate systems can be mutually rotated. As shown in fig. 4, the coordinate axis α axis in the two-phase stationary coordinate system coincides with the a axis in the three-phase stationary coordinate system, and the coordinate axis β axis in the two-phase stationary coordinate system is perpendicular to the α axis by 90 degrees and is behind the α axis by 90 degrees in the counterclockwise direction. This transformation is called the Clark transformation.
The transformation process is represented by equation (2.1):
according to the principle of invariable total electromagnetic power, the method comprises the following steps:
N 2 i a ＝N 3 i A +N 3 i B cos120°+N 3 i c cos((-120°) (2.2)
N 2 i β ＝0+N 3 i B sin120°+N 3 i c sin(-120°) (2.3)
N 2 i 0 ＝KN 3 i A +KN 3 i B +KN 3 i c (2.4)
wherein N is 2 And N 3 The number of windings per phase on two-phase and three-phase stationary coordinate systems. Conversion to matrix form:
therefore, the method comprises the following steps:
taking down to obtain:
this has the following:
so that it is possible to obtain:
substitution of formula (2.6) to obtain the complete Clark transformation matrix:
(3) Two phase rotating coordinate system
After the three-phase static coordinate system is converted into the two-phase static coordinate system, part of calculation and analysis can be simplified, but the synthetic magnetic fields in the two coordinate systems still carry out continuous circular motion around the origin, and because the angles change all the time, the analysis of the change amount in the two coordinate systems is still difficult. However, if we use the principle of physical relativity and assume that the coordinate axis can rotate simultaneously with the magnetic field vector making circular motion, the dynamic change of the motion of the direction of the synthetic magnetic field can be realized, and the coordinate in the rotating coordinate system is a fixed value, which can be realized only on the premise that the rotation speed of the rotating coordinate system is consistent with the rotation speed of the synthetic magnetic field with a circular track. In this way, the analysis and calculation of the system can be further simplified. Because the conversion process of ensuring the rotation of the magnetic field and simplifying the analysis calculation amount exists, the winding input in the rotating coordinate system only needs to input direct current, the effect is similar to that of a direct current motor, the conversion is called Park conversion, and according to the difference between a graph 4 and a graph 5, the Park conversion formula is a formula (2.11), and the formula (2.12) is the inverse conversion of the Park conversion.
3.SVPWM principle and algorithm implementation
(1) SVPWM principle
The vector control is to finally convert variables in the three-phase coordinate system into variables under the two-phase rotating coordinate by means of a coordinate transformation formula, so that the SVPWM control can be realized in a mode of simulating direct current motor control.
The current control method adopted by the embodiment is rotor magnetic field orientation control (i) d =0 control): i.e. i d The =0 control is a relatively simple one of all vector control methods. The excitation component of the stator armature current is always 0 in the control process, and the control method is equivalent to a direct-axis open circuit. Namely, the direct-axis current in the two-phase rotating coordinate is kept to be zero, and the stator current of a Permanent Magnet Synchronous Motor (PMSM) only has a quadrature-axis component, so that the control of the PMSM is as simple as the control of a direct-current motor. For surface-mounted PMSM, L of the motor d ≈L q At the moment, the torque equation of the motor can be simplified into the formula (3.11), the direct-axis current and the stator winding are completely decoupled, the direction of the quadrature axis is coincident with the direction of the synthetic magnetic potential of the stator and is just orthogonal to the magnetic potential of the rotor, and under the condition, the quadrature-axis current is only controlled, so that a good effect can be obtained. When the motor adopts the control mode, the minimum torque and the maximum rotating speed which are required by the motor are limited due to the maximum voltage allowed by the inverter, and if the power output by the inverter is kept constant, the electromagnetic torque output by the motor is reduced along with the increase of the output voltage of the inverter, so that higher torque can be obtainedThe rotational speed of (2).
Cause i d The control method of =0 is easy to realize in engineering, flexible to control, simple in algorithm, hard in mechanical characteristic and good in torque characteristic; and for the surface-mounted PMSM, only electromagnetic torque can be used, and good control performance can be obtained, so the design adopts the control mode.
SVPWM utilizes the voltage space vector who changes according to certain law alternation to realize circular rotating magnetic field, and this circular rotating magnetic field amplitude is invariable to reach the purpose that the motor can produce invariable electromagnetic torque.
Setting the amplitude of the phase voltage to U m Power supply frequency f, three voltage vectors U A (t)、U B (t)、U C (t), equal in magnitude, 120 degrees out of phase, then have:
the three-phase space voltage vector can be expressed as follows:
it can be seen from equation (3.2) that U (t) is a space rotation voltage vector having a magnitude 1.5 times the magnitude of the phase voltage and rotating at a constant speed in the counterclockwise direction at an angular velocity ω =2 π f. That is, when three-phase sinusoidal voltage is applied to the stator winding, the air-gap flux rotates at a constant angular velocity and the trajectory is circular because the voltage-synthesized space vector is a constant-amplitude rotation vector.
The basic principle of SVPWM is that eight voltage basic vectors are combined by different combination modes of six switching tubes, and any space voltage vector with required amplitude and direction can be combined by controlling the switch conduction time and effectively combining the eight vectors.
The three-phase inverter topology structure is shown in fig. 6, and the dc bus voltage is U dc The three-phase voltage output by the inverter is U A 、 U B 、U C 。
Taking an a-phase bridge arm as an example, the MOS transistors VT1 and VT4 on the bridge arm where the a-phase bridge arm is located cannot operate simultaneously, so in order to study the output voltage vectors of the three-phase voltage inverter under different switch combinations, a switch function is defined:
(S a ,S b ,S c ) There are 8 possible combinations that can be included: u shape 0 (000)、U 1 (001)、U 2 (010)、U 3 (011)、U 4 (100)、U 5 (101)、U 6 (110)、U 7 (111). Combined U 1 ～U 6 I.e. the mentioned basic voltage vector; according to the definition of the status identifier, U 0 And U 7 The lower three tubes or the upper three tubes of the three bridge arms are simultaneously conducted, and the two conduction states are meaningless and are zero vectors. Six basic voltage vectors divide the flux linkage circle equally into 6 regions (sectors), defining U 4 (100) The direction is the right horizontal direction as shown in fig. 7.
Taking a switch combination as an example, assume (S) a ,S b ,S c ) = (001), then:
U AB ＝U dc ,U BC ＝0,U CA ＝-U dc
U AN -U BN ＝U dc ,U AN -U CN ＝U dc
U AN +U BN +U CN ＝0
solving the equations the above equations can be obtained
The voltage magnitude of the three-phase winding in other switch states can be calculated by the same method. Or it may be calculated as: motor side three-phase winding input voltage vector [ U ] A U B U C ] T And the switch state vector [ S a S b S c ] T The relationship between can be summarized as:
in the formula of U dc The relationship between the dc-side power supply voltage and the dc-side power supply voltage is shown in table 1.
TABLE 1 relationship between switching states, phase voltages, line voltages and vector signs
U in the above table A 、U B 、U C The voltage vectors are respectively synthesized into a basic voltage vector U 1 ～U 6 The following results are obtained: the amplitudes of the six proper non-zero voltages are equal, and the module lengths are all 2/3U dc . Two adjacent basic voltage vectors are spaced by 60 degrees, and two zero vectors are located at the center and have zero amplitude. The magnitude and distribution of the eight space voltage vectors are shown in fig. 7. The rotating voltage is generated by different combinations of the eight space vectors, and the voltage U needed by the combination of two adjacent non-zero vectors and one zero vector is commonly used ref 。
As can be seen from FIG. 7, the whole circular region is divided into six sectors with equal areas, and each sector is divided by two adjacent non-zero voltage vectors when the voltage vector U is used as a boundary ref When turning to one of the sectors, the voltage vector U can be synthesized by the space voltage vector and the zero vector adjacent to the sector ref . In the actual control process, the resultant voltage vector U is limited by the switching frequency ref Is a polygon that approximates a circle. When the frequency rises to a certain extent, the shape will look like a circle.
(2) Specific algorithm implementation of SVPWM
The SVPWM function can be realized by three steps: (1) First, a composite space voltage vector is determinedQuantity U ref To which sector, thereby determining two adjacent basic voltage vectors within the sector; (2) Then, a space voltage vector U is determined ref The action time and the duty ratio of two adjacent voltage vectors in the sector; (3) And finally, determining a switching point of the space voltage vector, and synthesizing an SVPWM signal.
Now with the voltage vector U ref The SVPWM implementation is illustrated in the first sector as an example, and the voltage vector U is synthesized by two voltage vectors adjacent to the first sector ref As shown in fig. 7, a voltage vector U ref The expression in two-phase stationary coordinates is: u shape ref ＝u α +ju β (although not so simple synthesis). Note here in particular u α 、u β Is the voltage u under a two-phase static coordinate system obtained by inverse Park transformation α 、u β 。
1) Sector judgment
Suppose U ref In the I region, its equivalence condition in the sector can be expressed as: 0 ° < arctan (u) β /u α ) < 60 °, equivalently expressed by geometric calculations as:
u β is greater than 0. Similarly, the determination conditions of other sectors can be obtained, as shown in table 2.
TABLE 2 u α 、u β Conditions to be met in different sectors
From Table 2, it can be seen that β 、
The signs of the three terms determine the space voltage vector U ref In which sector, the three equations are defined:
to facilitate representation of the resultant voltage vector U ref The relation between the located sector and the formula (3.4) defines three parameters N a 、N b 、 N c The following relationship is satisfied:
when a is more than 0, taking N a =1, otherwise take N a ＝0；
When b is greater than 0, taking N b =1, otherwise take N b ＝0；
When c is greater than 0, taking N c =1, otherwise take N c ＝0。
From which N can be seen a 、N b 、N c There are eight combinations, but N can be obtained by the formula (3.4) a 、N b 、N c It is not possible to have 1's at the same time or 0's at the same time, so there are only six combinations in practice, and different combinations correspond to different sectors, each combination corresponding to a unique one of the sectors. So that available N a 、N b 、N c To determine U ref The sector in which it is located. To distinguish these different combinations, a composite voltage vector U is defined ref The number corresponding to the sector number is N sec ：
N sec ＝N a +2N b +4N c (3.5)
Then N is sec Same voltage vector U ref The sectors determined have the following relationship, as shown in table 3:
TABLE 3N sec Table of values and corresponding sectors
2) Calculation of action time of voltage vector and duty ratio thereof
Synthesizing any voltage vector in each sector according to the principle of volt-second balance, U can be seen from FIG. 7 ref Should be formed by a non-zero voltage vector U 4 、U 6 And zero voltage vector U 0 、U 7 To synthesize, the following formula：
TU ref ＝T 1 U 4 +T 2 U 6 +T 0 U 0/7
T＝T 1 +T 2 +T 0 (3.6)
Wherein T is 1 、T 2 、T 0 Respectively corresponding to voltage vectors U 4 、U 6 、U 0/7 The on-time within one cycle; t is the switching period, which can be obtained from FIG. 7:
the amplitude of each voltage vector effective value is known to be 2/3U dc From equation (3.7), two adjacent voltage vectors U in the first sector can be obtained 4 、U 6 The respective on-times are as follows:
similarly, the action time of each other adjacent basic voltage vector in the corresponding sector is calculated, and the working time of the adjacent basic voltage space vector in each sector is shown in table 4.
TABLE 4 working time of neighboring elementary voltage space vectors in each sector (U in the table) d Is namely U dc )
In a PWM period, setting the acting time of each basic voltage vector in the corresponding sector as T x 、T y It is apparent that the duty ratio of each basic voltage vector acting on the corresponding sector can be represented as T x /T、T y /T。
As can be seen from Table 4, the duty cycle of the adjacent fundamental voltage vectors, expressed by the general formula, is:
thus, the duty cycle T of the adjacent fundamental voltage space vector action x /T、T y T and its action time T x 、T y Specifically, as shown in table 5:
TABLE 5 Duty ratio of adjacent basis voltage space vector contributions T x /T、T y T and its action time T x 、T y
| Sector number | I | II | III | IV | V | VI |
| Duty cycle T x /T | K 3 | K 3 | -K 1 | -K 1 | -K 2 | -K 3 |
| Time of action T x | K 3 T | K 3 T | -K 1 T | -K 1 T | -K 2 T | -K 3 T |
| Duty cycle T y /T | K 1 | K 2 | K 2 | -K 3 | -K 3 | K 1 |
| Time of action T y | K 1 T | K 2 T | K 2 T | -K 3 T | -K 3 T | K 1 T |
Wherein, when T x +T y When > T, T x ＝T x T/(T x +T y )，T y ＝T y T/(T x +T y )，T 0 =0; when T is x +T y When < T, T x 、T y Remains unchanged, T 0 ＝T-T x -T y 。
(3) Calculation of switching time and PWM signal synthesis
The calculation of the vector switching time is now defined as follows:
as previously mentioned, table 6 lists the handoff times for each sector.
TABLE 6 selection of handoff time for each sector
Given the voltage vector on-time for each sector, the generation of the PWM signal can be synthesized using a seven-segment SVPWM. The seven-segment SVPWM method is to reduce the turn-on and turn-off times of the switching device and prolong the service life of the device, so that the switching state of one phase can be changed to meet the requirement when the switching state is switched every time. Meanwhile, in order to reduce harmonic components of PWM, two zero vectors can be evenly distributed, so that the generated PWM waveform is symmetrical.
The most obvious characteristic of this method is that each period starts with a zero vector and ends with a zero vector, and only the state of one phase is changed each time, so that the energy consumption can be saved. Table 6 shows the SVPWM output for six sectors.
The situation in the I-th sector is shown in fig. 8.
1), 2), 3) implemented according to the aforementioned SVPWM algorithm, a flow chart of the program algorithm is shown in fig. 9. The method comprises the following steps: first, in the current loop control, u to be obtained α And u β Value (u here) α 、u β That is, the voltage in the two-phase stationary coordinate system obtained by inverse Park transformation in fig. 4) is calculated to find N a 、N b 、N c Then, the number N corresponding to the sector where the voltage is located can be known sec The sector in which the voltage is located is determined according to table 3. The action time T of two adjacent basic voltage vectors of the sector can be calculated by the formula (3.9) x 、T y Then, the switching time t is obtained from the formula (3.10) aon 、t bon 、t con And finally, generating six paths of complementary SVPWM signals and outputting the signals to a driving circuit, and further controlling the main inverter circuit to be switched on and off at corresponding moments to generate sine waves to control the brushless motor to continuously operate.
FIG. 10 shows a surface-mounted permanent magnet synchronous motor at i d The vector control block diagram of =0 adopts a current and speed double closed-loop speed regulating system, wherein the electromagnetic torque formula is as follows:
T e ＝n p ψ f i q (3.11)
wherein, T e Is an electromagnetic torque, n p For number of pole pairs, psi, of the motor f Is a permanent magnet flux linkage i q Respectively, the components of the resultant current vector in the q-axis. That is, in the d-q coordinate system, the electromagnetic torque of the surface-mounted PMSM is linearly related to the q-axis component current. Therefore, the electromagnetic torque T of the motor can be realized by controlling the torque current component of the q axis e Control of (2). Meanwhile, all variables are direct current variables in a d-q coordinate system.
The system shown in FIG. 10 employs two closed loop controls, one speed loop and one current loop, where the inner loop, which has a limiting effect on the performance of the system, is an important part of the torque control; the outer ring can make the speed finally stable on the expected value, and the effect of eliminating the interference of external factors is achieved by tracking the speed command. Given i of direct axis current d =0, output of speed outer loop ASR
A given quadrature current is determined. The control process of the current and speed double closed-loop speed regulating system is as follows:
(1) Current i flowing in two-phase winding of actual motor a And i b The value of (a) is measured by a current sensor, and the current in an actual coordinate system is subjected to Clark coordinate transformation to obtain the current i α 、i β The value is a value in two-phase stationary coordinates; will current i α 、i β Obtaining a current i after Park conversion d 、i q 。
(2) Will generate electricityThe difference value between the actual rotating speed and the given rotating speed of the machine is used as the input of ASR, and the output is
(3) System provisioning
Will be provided with
And i d The difference of,
And i q The difference value of the first and second voltage values is used as the input of a corresponding current PI regulator, and the direct-axis voltage u is respectively obtained through the output of the PI regulator d =0 and quadrature voltage u q The voltage u under the two-phase static coordinate system can be obtained by inverse Park transformation α 、u β 。
(4) Modulating the drive output with a voltage Space Vector (SVPWM) by first determining u at a certain time α And u β And synthesizing the sector where the vector is located, then respectively calculating the action time of the zero vector in the sector and the action time of the adjacent voltage vector to obtain six paths of signals of PWM (pulse-width modulation) modulation waves to control six power switching tubes of the voltage type inverter, obtaining the voltage required by the motor through the conduction and the disconnection of the switching tubes to drive the motor to normally operate, and finally finishing the vector control.
The motor rotating speed and position judgment and the motor rotor position estimation in the embodiment can adopt the prior art.
4. Comparison of simulation experiments
(1) Experiment platform
The hardware platform used in the present embodiment is an STM32 development board. The main chip is STM32F103ZET6, and is the highest chip of configuration in the STM32F1 common model. STM 32's function is various, has included this all kinds of timers that motor was last, corresponds modules such as input/output port, hall sensor.
The software platform used in this example is keilMDK. The design of the software part of the system is mainly based on a German RealView MDK development platform, and the whole system is written in C language except a starting code.
(2) Simulation program construction
The simulation platform of the embodiment is a matlab/simulink platform, can provide a dynamic modeling integrated environment, can form a complete simulation model only by dragging out the required elements in a library with a mouse and then carrying out corresponding connection, does not need to compile a large amount of program codes, and is simple and visual in the whole simulation program model diagram. The simulation software is used for building and simulating waveforms of the simple square wave control system and the sine wave control system, and the simulation waveforms of the simple square wave control system and the sine wave control system are compared. The simulation type used selects Continuous, i.e. Continuous mode, which is more intuitive than Discrete mode of Discrete, and the observed waveform is smooth and stable.
1) Simple square wave control system building
The simple square wave control system module has less content, and can complete simple square wave control driving operation only by the brushless direct current motor, the inverter circuit module and the pulse generating module, wherein the PWM module is contained in the pulse generating module. The overall model of the control system is shown in fig. 11.
The universal bridge module selects a three-bridge mode, and the internal structure of the universal bridge module is consistent with that of the universal bridge module shown in figure 2. The pulse generating module is used for judging the approximate position of the current rotor magnetic field according to the signal input of the current Hall sensor, generating a magnetic field direction with the maximum average torque according to the position and the relation of a table 7 by adopting a 120-degree conduction mode to drag the motor to rotate, performing phase change when the motor rotates by one sector, and controlling the duty ratio of the output pulse by adopting a PWM mode of using an upper bridge arm and a lower bridge arm. The specific simulation set-up of the module is shown in fig. 12.
The function module is used for converting the three Hall sensor detection signals into 6 PWM signals, and the conversion rule is carried out according to the relation of a table 7.
TABLE 7 electrified phase sequence under 120 degree conduction mode
In order to facilitate comparison with the sine wave-driven simulation waveform of the embodiment, the PWM on ratio value is 1 in the simulation screenshot, that is, the full on state, and the actual motor can operate in a normal PWM manner only by modifying the PWM input value of the corresponding upper or lower bridge arm to a proper value.
2) Sine wave control system construction
The simulation model of the sine wave control system has more building contents and comprises a brushless direct current motor, an inverter circuit module, a PI (proportion integration) adjusting module, a coordinate transformation module, an SVPWM (space vector pulse width modulation) module, a phase current sampling module and the like. In this embodiment, a brushless dc motor with 4 pairs of poles is adopted, and p =4 is also selected as a model parameter of the brushless dc motor in the simulation software. The relation between the mechanical angle and the electrical angle of the brushless motor with 4 pairs of poles is 4 times, wherein the angular speed omega and the angle theta displayed by the output port m are relative to the electrical angle, so when the real rotor position and the motor rotating speed are judged, the output signal needs to be multiplied by 4 times, the unit of the angular speed is rad/s, when the rotating speed of the motor converted per minute is measured, the output signal needs to be multiplied by a unit conversion coefficient, and the unit conversion coefficient is 60/(2 x pi), so that the real rotating speed of the motor per minute can be measured. The overall system model is shown in fig. 9.
(3) Simulation results and analysis
1) Simulation result of sine wave control system
Because the brushless motor used in the design is a 4-pair-pole motor, the rotor magnetic field needs to be converted before the motor rotating speed is detected:
actual motor speed (r/min) = Rotor speed wm (rad/s) × 4 (number of poles) × 30/pi (unit conversion)
In the simulation software of the present design, when θ =0 of the dc brushless motor, the rotor flux position lags the a phase position by 90 degrees, so the phase difference of 90 degrees is added to the simulation module before the rotor position is determined, which is specifically shown in the simulation general block diagram 13.
Fig. 14 is a diagram of actual phase current of the brushless motor, which is a waveform diagram of the overall simulation model fig. 13, and it can be seen that the set rotation speed at this time is set to 500 rpm. FIG. 15 is a three-phase current diagram at 1000 rpm. When the set value is changed to 1000rpm, namely the rotating speed is doubled, the obtained three-phase current exchange frequency after the motor stably operates is doubled as well.
As shown in fig. 16, the sector decision waveform diagram is a cyclic waveform diagram of 1 → 2 → 3 → 4 → 5 → 6 and then 1, the sector division diagram 7 according to the embodiment can determine that the brushless motor rotates counterclockwise in the simulation operation, and the sector decision module also provides a selection function for the subsequent modules, so that each sector can correctly select the input terminal to satisfy the modulation of the sine wave.
Fig. 17 is a waveform diagram of the basic voltage vector acting time distribution, when the motor rotates at a constant speed stably, the two quantities are distributed in such a way that the acting time of one quantity changes from 0 to the maximum value at the end of any sector, and the other quantity changes from the maximum value to 0 at the same time, and finally a circular flux linkage is formed, so that the motor can be judged to operate at a constant speed circularly, and the motor is in line with the expectation of the embodiment.
Fig. 18 is a schematic diagram showing a simulation result of the motor torque, and fig. 19 is a schematic diagram showing a simulation result of the motor rotation speed. After smooth running, the motor torque waveform only fluctuates between 2.93 and 3.07, the fluctuation amplitude is small, and the rotating speed waveform is also kept to be almost a straight line at 500 revolutions per minute. The brushless motor stably and continuously rotates anticlockwise at the speed of 500 revolutions per minute, and the sine wave design simulation is successfully realized.
2) Advantages over Square wave control
The control mode of square wave drive can only generate magnetic fields in 6 specific directions to drag the rotor to rotate, and the fluctuation is inevitably larger than that generated by the sine wave control mode, and the simple square wave control system and the sine wave control system built in the simulation experiment are subjected to waveform comparison. For convenience of comparison, the PWM pulse width modulation simulated by the square wave control system is set to 1, i.e. the full conduction mode, although the rotation speed of the motor is fast, the fluctuation condition of each waveform of the square wave control system can be analyzed only by full conduction, and the observed content is mainly the fluctuation ratio of the waveform, so the magnitude of the rotation speed does not affect the analysis of the advantages and disadvantages of the two control modes.
Fig. 20 is a waveform diagram of the motor torque of the square wave control system, which is better compared with a waveform diagram of the sine wave driving mode, and the unit length of the horizontal axis and the vertical axis of the waveform diagram is the same as that of fig. 18, and compared with the waveform diagram, it is found that in the same two control modes with the torque set value of 3N · m, the motor torque of the sine wave control mode only fluctuates between 2.93 and 3.07, the fluctuation amplitude is small, and the motor torque of the square wave driving mode fluctuates between 1.9 and 4.4, the fluctuation amplitude is large, which shows the advantages that the sine wave control system can smoothly run, the noise is small, the rotating speed is uniform compared with the square wave control system, and simultaneously, the correctness of the design simulation is verified.
The technical solutions provided by the embodiments of the present invention are described in detail above, and the principles and embodiments of the present invention are explained herein by using specific examples, and the descriptions of the embodiments are only used to help understanding the principles of the embodiments of the present invention; meanwhile, for a person skilled in the art, according to the embodiments of the present invention, there may be variations in the specific implementation manners and application ranges, and in summary, the content of the present description should not be construed as a limitation to the present invention.
Claims (7)
1. The SVPWM control device of the brushless DC motor is characterized by comprising a main control chip, a DC voltage and bus current sampling circuit, a switching power supply, a DC power supply, a driving circuit, a main inverter circuit and a Hall position detection circuit; the direct current power supply is respectively connected with the main inverter circuit and the switching power supply; the switching power supply is respectively connected with the direct-current voltage and bus current sampling circuit, the main control chip and the driving circuit; the main control chip is connected with the direct-current voltage and bus current sampling circuit to obtain direct-current voltage and three-phase current signals of the motor; the input end of the driving circuit and the output end of the main control chipThe output end of the main inverter circuit is connected with the output end of the voltage regulator; the main inverter circuit and the Hall position detection circuit are respectively connected with the brushless direct current motor; the main control chip is connected with the output end of the Hall position detection circuit to obtain a detection signal of the Hall sensor; the main control chip judges the space voltage vector U according to the collected signals ref The sector is located, so as to determine two adjacent basic voltage vectors in the sector, and then determine a space voltage vector U ref The action time and the duty ratio of two adjacent voltage vectors in the sector finally determine a space voltage vector U ref And synthesizing the SVPWM signals and outputting the SVPWM signals to the driving circuit, and further controlling the main inverter circuit to be switched on and off at corresponding moments to generate sine waves to control the brushless motor to continuously operate.
2. The SVPWM control apparatus of claim 1, wherein said brushless motor is a 4-pole brushless motor.
3. The SVPWM control apparatus of claim 1, wherein said brushless DC motor is a surface-mounted permanent magnet synchronous motor.
4. The SVPWM control apparatus of brushless DC motor according to claim 3, wherein the electromagnetic torque equation of the surface-mounted permanent magnet synchronous motor in d-q coordinate system is as follows:
T e ＝n p ψ f i q
wherein, T e Is an electromagnetic torque, n p For number of pole pairs, psi, of the motor f Is a permanent magnet flux linkage i q Respectively, the components of the resultant current vector in the q-axis.
5. The SVPWM control apparatus of claim 1, wherein said SVPWM is a 7-segment SVPWM.
6. The SVPWM control apparatus for a brushless dc motor according to claim 1, wherein the main inverter circuit is a three-phase inverter.
7. An SVPWM control method of a brushless dc motor based on the apparatus according to any one of claims 1 to 6, comprising the steps of:
(1) In the current loop control, the voltage u under the two-phase static coordinate system of the motor is obtained α And u β ，
(2) According to voltage u α And u β Calculating and judging space voltage vector U ref The sector in which the cell is located;
(3) Determining a space voltage vector U ref Action time T of two adjacent voltage vectors in the located sector x 、T y And a switching time t aon 、t bon 、t con And generating six paths of complementary SVPWM signals and outputting the signals to a driving circuit, and further controlling a main inverter circuit to be switched on and off at corresponding time to generate sine waves to control the brushless motor to continuously operate.
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