CN115133834A - Two-degree-of-freedom harmonic current control method of high-speed double three-phase permanent magnet synchronous motor - Google Patents

Two-degree-of-freedom harmonic current control method of high-speed double three-phase permanent magnet synchronous motor Download PDF

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CN115133834A
CN115133834A CN202210919138.6A CN202210919138A CN115133834A CN 115133834 A CN115133834 A CN 115133834A CN 202210919138 A CN202210919138 A CN 202210919138A CN 115133834 A CN115133834 A CN 115133834A
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CN115133834B (en
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杨贵杰
鲁光旭
苏健勇
刘凤扬
王学鹏
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Harbin Institute of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
    • Y02E40/40—Arrangements for reducing harmonics

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Abstract

A two-degree-of-freedom harmonic current control method for a high-speed double three-phase permanent magnet synchronous motor relates to the technical field of motor control. The invention aims to solve the problem that the harmonic wave is difficult to inhibit in the application occasions with wide speed regulation range, large load disturbance and frequent change of operation working conditions in the conventional harmonic wave inhibition technology. The invention transforms a phase voltage equation of a motor to obtain a voltage equation under a fundamental wave synchronous rotating coordinate system and a harmonic anti-synchronous rotating coordinate system; converting the signal into a motor transfer function under a complex frequency domain, and setting the signal into a generalized second-order system; obtaining a gain coefficient in a transfer function of the controller and then dispersing; acquiring phase current of a motor, and obtaining current under a fundamental wave synchronous rotating coordinate system and a harmonic anti-synchronous rotating coordinate system through coordinate transformation; calculating the voltage under a fundamental wave static coordinate system and a harmonic static coordinate system; and carrying out SVPWM modulation on the voltage under the fundamental wave static coordinate system and the harmonic static coordinate system, and then controlling the high-speed double three-phase permanent magnet synchronous motor.

Description

Two-degree-of-freedom harmonic current control method of high-speed double three-phase permanent magnet synchronous motor
Technical Field
The invention belongs to the technical field of motor control, and particularly relates to control over two-degree-of-freedom harmonic current.
Background
The high-speed double-three-phase permanent magnet synchronous motor has the advantages of a high-speed motor and a multi-phase motor, has the advantages of high efficiency, high power density, high-speed direct drive and the like, and is widely applied to aviation actuators and aviation starting systems. The high speed and multiple phase of the motor improve the power density and power level of the motor system, but also increase the harmonic problem of the motor system. The harmonic current of the high-speed multi-phase motor is difficult to suppress compared with the normal-speed motor. On the one hand, the motor fundamental frequency is improved due to high speed, the carrier ratio is reduced, and meanwhile, in the design process of the body, the inductance of a high-speed motor winding is smaller due to the constraint of the outer diameter of a rotor and the electric load. On the other hand, the multi-phase can increase the dimension of current control, the harmonic planar inductance of the double three-phase motor is composed of leakage inductance, and the harmonic planar inductance is smaller than the fundamental planar inductance. For the above reasons, current harmonics are a common problem in high speed dual three-phase permanent magnet synchronous motor systems. The current harmonic wave occupies the current capacity of the inverter, increases the loss of the motor and increases the heat dissipation burden of the motor system. Therefore, a harmonic suppression strategy is required to be adopted in a high-speed double three-phase motor system to meet the requirements of the transmission field with high power density, high power and high efficiency.
In the active harmonic suppression technology, the multi-proportion resonance controller and the harmonic voltage disturbance observer have obvious advantages in harmonic suppression effect and calculation amount, and are widely applied. Both are to add a resonance pole into a current closed loop through a controller, and generate infinite closed loop gain at a resonance frequency to suppress current harmonics of a corresponding frequency. The resonance frequency used in the controller is obtained by rotating speed feedback calculation, and because the rotating speed calculation has time delay, the calculation result generally needs filtering processing, and the instantaneous rotating speed and the resonance frequency cannot be accurately obtained, so that the deviation of the resonance frequency is inevitable in the control process. A shift in the resonance frequency results in a significant reduction in the harmonic suppression effect. In the application occasions of the aviation actuator, such as wide speed regulation range, large load disturbance and frequent change of operation conditions, the good harmonic suppression effect is difficult to ensure.
Disclosure of Invention
The invention provides a two-degree-of-freedom harmonic current control method of a high-speed double three-phase permanent magnet synchronous motor, aiming at solving the problem that the prior harmonic suppression technology is difficult to ensure good harmonic suppression effect in the application occasions with wide speed regulation range, large load disturbance and frequent change of operation working conditions.
The method comprises the following steps: carrying out vector space decoupling transformation on a phase voltage equation of the motor, and then respectively carrying out park transformation and park inverse transformation on a transformation result to respectively obtain a voltage equation under a fundamental wave synchronous rotating coordinate system and a voltage equation under a harmonic wave anti-synchronous rotating coordinate system;
step two: respectively carrying out Laplace transform on the two voltage equations obtained in the step one in a complex frequency domain to obtain motor transfer functions in the two complex frequency domains;
step three: respectively setting two motor transfer functions into standard generalized second-order systems according to a zero-pole cancellation principle, and further obtaining controller transfer functions of the two generalized second-order systems;
step four: selecting the adjusting time t according to the actual performance requirement of the motor control system s And a second-order system damping coefficient zeta, and then a gain coefficient in a transfer function of the controller is obtained;
step five: discretizing the controller transfer function with the gain coefficient to obtain two discrete controller transfer functions;
step six: collecting phase current of a motor, carrying out vector space decoupling transformation on the phase current to obtain current under a fundamental wave static coordinate system and current under a harmonic static coordinate system, then carrying out park transformation on the current under the fundamental wave static coordinate system to obtain current under a fundamental wave synchronous rotating coordinate system, and carrying out park inverse transformation on the current under the harmonic static coordinate system to obtain current under a harmonic anti-synchronous rotating coordinate system;
step seven: respectively inputting a given current instruction, current under a fundamental wave synchronous rotation coordinate system and current under a harmonic anti-synchronous rotation coordinate system into the discrete controllers, and then respectively carrying out park transformation and park inverse transformation with angle compensation on the two output voltages to obtain a fundamental wave static coordinate system and voltage under the harmonic static coordinate system;
step eight: and carrying out SVPWM modulation on the voltage under the fundamental wave static coordinate system and the harmonic static coordinate system to obtain a PWM signal of the six-phase inverter, and controlling the high-speed double three-phase permanent magnet synchronous motor by using the PWM signal.
In a further step one, a voltage equation in the fundamental wave synchronous rotation coordinate system is as follows:
Figure BDA0003774984730000021
the voltage equation under the harmonic anti-synchronous rotating coordinate system is as follows:
Figure BDA0003774984730000022
wherein u is dq ＝u d +ju q ，u d And u q The voltages of the d axis and the q axis under the fundamental wave synchronous rotation coordinate system are respectively,
i dq ＝i d +ji q ，i d and i q Are d-axis current and q-axis current under a fundamental wave synchronous rotating coordinate system respectively,
L dq ＝L d +jL q ，L d and L q Are respectively the d-axis inductance and the q-axis inductance under the fundamental wave synchronous rotating coordinate system,
Figure BDA0003774984730000023
and
Figure BDA0003774984730000024
respectively harmonic anti-synchronizationThe voltages of the z1 axis and the z2 axis under the rotating coordinate system,
Figure BDA0003774984730000025
and
Figure BDA0003774984730000026
the current of the z1 axis and the z2 axis under the harmonic anti-synchronous rotating coordinate system respectively,
L z1z2 ＝L z1 +jL z2 ，L z1 and L z2 The inductances of the z1 axis and the z2 axis under the harmonic anti-synchronous rotating coordinate system respectively,
j is an imaginary unit, R s Is stator resistance, ω e For electrical angular frequency, # f Is a permanent magnetic linkage.
In the second step, the motor transfer functions in the two complex frequency domains are respectively the motor transfer functions in the fundamental wave synchronous rotation coordinate system
Figure BDA0003774984730000031
Transfer function of motor under harmonic anti-synchronous rotating coordinate system
Figure BDA0003774984730000032
The expressions for the two motor transfer functions are respectively as follows:
Figure BDA0003774984730000033
Figure BDA0003774984730000034
wherein s is a laplace operator.
Further, the controller transfer functions of two generalized second-order systems in the third step are controller transfer functions under the fundamental wave synchronous rotation coordinate system respectively
Figure BDA0003774984730000035
Transfer function of controller under harmonic anti-synchronous rotating coordinate system
Figure BDA0003774984730000036
The expressions for the two controller transfer functions are respectively as follows:
Figure BDA0003774984730000037
Figure BDA0003774984730000038
wherein the content of the first and second substances,
Figure BDA0003774984730000039
and
Figure BDA00037749847300000310
two gain coefficients under the fundamental wave synchronous rotation coordinate system,
Figure BDA00037749847300000311
and
Figure BDA00037749847300000312
two gain coefficients, omega, in harmonic anti-synchronous rotating coordinate system 1dq And ω 1zr The harmonic synchronous rotating coordinate system and the harmonic anti-synchronous rotating coordinate system are respectively the resonance frequency under the fundamental wave synchronous rotating coordinate system and the harmonic anti-synchronous rotating coordinate system.
Further, in the fourth step, when the allowable error band for the harmonic control is 2%, the gain coefficient in the controller transfer function is obtained by the following formula:
Figure BDA00037749847300000313
wherein when ω is 1 ＝ω 1dq When the temperature of the water is higher than the set temperature,
Figure BDA00037749847300000314
when ω is 1 ＝ω 1zr When the temperature of the water is higher than the set temperature,
Figure BDA00037749847300000315
in the fourth step, the time t is adjusted s The value range of (1) is 0.5 ms-20 ms, and the value range of the second-order system damping coefficient zeta is 0.350-0.707.
Further, in the fifth step, a specific method for discretizing the controller transfer function for which the gain coefficient is determined is as follows:
adapting the controller transfer function for which the gain factor is determined to a form of a sum of polynomials:
Figure BDA0003774984730000041
Figure BDA0003774984730000042
wherein the content of the first and second substances,
Figure BDA0003774984730000043
when ω is 1 ＝ω 1dq When the utility model is used, the water is discharged,
Figure BDA0003774984730000044
when ω is 1 ＝ω 1zr When the temperature of the water is higher than the set temperature,
Figure BDA0003774984730000045
then are respectively paired
Figure BDA0003774984730000046
And Q c (ω 1 ,K GI2 ) Discretizing, wherein expressions after discretization are respectively as follows:
Figure BDA0003774984730000047
Figure BDA0003774984730000048
Figure BDA0003774984730000049
wherein σ 1 ＝T s ω 1 ，σ 2 ＝K d T s ω 1 ，σ 3 ＝K GI2 T s ω 1 ，
Figure BDA00037749847300000410
T s To sample time, K d For the delay compensation parameter, T d ＝K d T s 。
Further, after the eighth step, a resonance frequency shift experiment of the high-speed double three-phase permanent magnet synchronous motor under the steady state condition is performed, and the specific method of the resonance frequency shift experiment is as follows:
under the conditions of 50% rated rotation speed and 50% rated torque, the rotation speed of the motor is kept unchanged, and the resonance frequency omega of the motor is adjusted within the range of +/-10% 1 And testing whether the phase current harmonic wave of the motor is restrained, if so, finishing the harmonic wave current control, and if not, returning to the step four.
The invention realizes the following beneficial effects through the method:
1. the invention sets the response process of the harmonic current into a generalized type II system, the harmonic amplitude response is consistent with that of a typical type II system, the parameter meaning is clear, and the adjustment is convenient.
2. The invention has the capability of resisting the resonance frequency disturbance, and can still ensure good harmonic suppression effect even if the resonance frequency has certain deviation.
3. The controller structure of the invention realizes the decoupling of the motor model, improves the dynamic response effect of the motor system and improves the stability of the system.
4. The trigonometric function operation in the invention can be repeatedly used, the calculated amount is small, and the method can be conveniently expanded to higher-order harmonic suppression.
5. The method adopts the SVPWM method capable of realizing even harmonic elimination, reduces the common mode leakage current and winding circulation current of the high-speed double three-phase motor, and the odd harmonic added by the SVPWM strategy for even harmonic elimination can be inhibited by the method.
In conclusion, the parameter design process is determined, the resistance of the algorithm to frequency deviation is enhanced, and a better harmonic suppression effect can be still realized when the resonant frequency deviates.
Drawings
FIG. 1 is a flow chart of a two-degree-of-freedom harmonic current control method of a high-speed double three-phase permanent magnet synchronous motor according to the invention;
FIG. 2 is a functional block diagram of a dq plane current controller;
FIG. 3 is a functional block diagram of a z1z2r planar current controller;
FIG. 4 is a structural block diagram of a control system of a dual three-phase permanent magnet synchronous motor;
FIG. 5 is a graph of closed loop harmonic admittance in the z1z2r plane;
FIG. 6 is a waveform diagram of an experimental resonant frequency sweep;
FIG. 7 is an experimental waveform diagram of the start-up and steady-state effects of the harmonic suppression algorithm.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention. It should be noted that the embodiments and features of the embodiments of the present invention may be combined with each other without conflict.
The first embodiment is as follows: specifically describing the present embodiment with reference to fig. 1 to 7, the method for controlling two-degree-of-freedom harmonic current of a high-speed dual three-phase permanent magnet synchronous motor according to the present embodiment includes the following steps:
the method comprises the following steps: and carrying out vector space decoupling transformation on a phase voltage equation of the motor, and then respectively carrying out park transformation and park inverse transformation on the transformation result to respectively obtain a voltage equation under a fundamental wave synchronous rotating coordinate system (dq plane) and a voltage equation under a harmonic anti-synchronous rotating coordinate system (z1z2r plane).
The voltage equation (complex vector form) under the fundamental wave synchronous rotation coordinate system is as follows:
Figure BDA0003774984730000051
the voltage equation (complex vector form) under the harmonic anti-synchronous rotating coordinate system is as follows:
Figure BDA0003774984730000052
wherein u is dq ＝u d +ju q ，u d And u q The voltages of the d axis and the q axis under the fundamental wave synchronous rotation coordinate system are respectively,
i dq ＝i d +ji q ，i d and i q Are d-axis current and q-axis current under a fundamental wave synchronous rotating coordinate system respectively,
L dq ＝L d +jL q ，L d and L q Are respectively the d-axis inductance and the q-axis inductance under the fundamental wave synchronous rotating coordinate system,
Figure BDA0003774984730000061
and
Figure BDA0003774984730000062
the voltages of the z1 axis and the z2 axis under the harmonic anti-synchronous rotating coordinate system respectively,
Figure BDA0003774984730000063
and
Figure BDA0003774984730000064
the current of the z1 axis and the z2 axis under the harmonic anti-synchronous rotating coordinate system respectively,
L z1z2 ＝L z1 +jL z2 ，L z1 and L z2 The inductances of the z1 axis and the z2 axis under the harmonic anti-synchronous rotating coordinate system respectively,
j is an imaginary unit, R s Is stator resistance, ω e For electrical angular frequency, # f Is a permanent magnetic linkage.
Step two: due to permanent magnetic potential j omega e ψ f Under the steady state condition, the two voltage equations obtained in the step one are subjected to Laplace transform respectively to obtain motor transfer functions under two complex frequency domains, wherein the motor transfer functions under the two complex frequency domains are respectively motor transfer functions under a fundamental wave synchronous rotating coordinate system
Figure BDA0003774984730000065
Transfer function of motor under harmonic anti-synchronous rotating coordinate system
Figure BDA0003774984730000066
The expressions for the two motor transfer functions are respectively as follows:
Figure BDA0003774984730000067
Figure BDA0003774984730000068
wherein s is a laplace operator.
Step three: respectively setting two motor transfer functions into standard generalized second-order systems according to a zero-pole cancellation principle to further obtain controller transfer functions of the two generalized second-order systems, wherein the controller transfer functions of the two generalized second-order systems are respectively controller transfer functions under a fundamental wave synchronous rotating coordinate system
Figure BDA0003774984730000069
Transfer function of controller under harmonic anti-synchronous rotating coordinate system
Figure BDA00037749847300000610
The expressions for the two controller transfer functions are respectively as follows:
Figure BDA00037749847300000611
Figure BDA00037749847300000612
wherein, the first and the second end of the pipe are connected with each other,
Figure BDA00037749847300000613
and
Figure BDA00037749847300000614
two gain coefficients under the fundamental wave synchronous rotation coordinate system,
Figure BDA00037749847300000615
and
Figure BDA00037749847300000616
two gain coefficients, omega, in harmonic anti-synchronous rotating coordinate system 1dq And ω 1zr Respectively the resonance frequency omega in the fundamental wave synchronous rotating coordinate system and the harmonic anti-synchronous rotating coordinate system 1dq ＝12ω e ，ω 1zr ＝6ω e 。
As shown in fig. 2 and 3, the dq motor controller is responsible for dq dc component control, and the dq plane harmonic current controller is responsible for the suppression of the 11th and 13th harmonics. The two controllers are in a parallel configuration. The resonant frequencies required by the dq plane harmonic controller and the z1z2r plane are calculated from the angle.
Step four: when the adjustment is selected according to the actual performance requirement of the motor control systemTime t s And a second-order system damping coefficient zeta, and then when the allowable error band of the motor control system is 2%, a gain coefficient in the transfer function of the controller is obtained by the following formula:
Figure BDA0003774984730000071
wherein when ω is 1 ＝ω 1dq When the temperature of the water is higher than the set temperature,
Figure BDA0003774984730000072
when ω is 1 ＝ω 1zr When the utility model is used, the water is discharged,
Figure BDA0003774984730000073
adjusting the time t s The convergence speed of the harmonic wave in the dynamic process is determined, and meanwhile, the capability of the harmonic wave suppression algorithm for resisting the resonance frequency deviation is also influenced. The second-order system damping coefficient zeta influences the capability of the harmonic suppression algorithm in resisting the deviation of the resonant frequency, and the effect of the algorithm in resisting the deviation of the resonant frequency can be enhanced by properly reducing the damping. When the modulation ratio is more than 30, the time t is adjusted under the influence of stability and harmonic suppression effect s The value range of (1) is 0.5 ms-20 ms, and the value range of the second-order system damping coefficient zeta is 0.350-0.707.
Step five: in continuous systems, the controller transfer function, which determines the gain factor, is adapted to the form of a polynomial sum:
Figure BDA0003774984730000074
Figure BDA0003774984730000075
wherein the content of the first and second substances,
Figure BDA0003774984730000076
when ω is 1 ＝ω 1dq When the utility model is used, the water is discharged,
Figure BDA0003774984730000077
when omega 1 ＝ω 1zr When the temperature of the water is higher than the set temperature,
Figure BDA0003774984730000078
for ensuring accuracy of resonant frequency and stability of system, P pair 1 c (ω 1 ) Adopting a time invariant method with delay compensation to disperse, wherein the expression after dispersion is as follows:
Figure BDA0003774984730000079
for is to
Figure BDA00037749847300000710
Dispersing by a bilinear transformation method with frequency pre-warping and delay compensation, wherein the expression after dispersion is as follows:
Figure BDA0003774984730000081
to Q c (ω 1 ,K GI2 ) The forward integrator is discretized by a zero-order retainer, the feedback integrator is discretized by an impact response time-invariant method, and the discretized expression is as follows:
Figure BDA0003774984730000082
in the above formula, σ 1 ＝T s ω 1 ，σ 2 ＝K d T s ω 1 ，σ 3 ＝K GI2 T s ω 1 ，
Figure BDA0003774984730000083
T s To sample time, K d For the delay compensation parameter, T d ＝K d T s 。
Two discrete controller transfer functions are finally obtained.
Step six: the method comprises the steps of collecting phase currents of a motor, carrying out vector space decoupling transformation on the phase currents to obtain currents under a fundamental wave static coordinate system and currents under a harmonic static coordinate system (alpha beta plane), then carrying out park transformation on the currents under the fundamental wave static coordinate system to obtain currents under a fundamental wave synchronous rotating coordinate system (z1z2 plane), and carrying out park inverse transformation on the currents under the harmonic static coordinate system to obtain currents under a harmonic anti-synchronous rotating coordinate system.
Step seven: and inputting a preset current command, a current under a fundamental wave synchronous rotating coordinate system and a current under a harmonic anti-synchronous rotating coordinate system into the discrete controller to obtain an output voltage, and then carrying out park conversion with angle compensation on the output voltage to obtain a voltage under a fundamental wave static coordinate system.
Step eight: and carrying out SVPWM modulation on the voltage under the fundamental wave static coordinate system to obtain a PWM signal of the six-phase inverter, and controlling the high-speed double three-phase permanent magnet synchronous motor by using the PWM signal.
Step nine: the method comprises the following steps of carrying out a resonance frequency deviation experiment of a high-speed double three-phase permanent magnet synchronous motor under a steady state condition, wherein the specific method of the resonance frequency deviation experiment comprises the following steps:
under the conditions of 50% rated rotation speed and 50% rated torque, the resonance frequency omega of the motor is adjusted within the range of +/-10% while keeping the rotation speed of the motor unchanged 1 And testing whether the phase current harmonic wave of the motor is restrained, if so, finishing the harmonic wave current control, and if not, returning to the step four.
In practical experiments, the specific parameters are set as follows: rated rotation speed 10000rpm, rated torque 4.1Nm, d-axis inductance 154.4uH, q-axis inductance 154.4uH, harmonic inductance 43.8uH, stator resistance 25.3m omega, switching frequency 50kHz, and control frequency 25 kHz. The controller parameters are as follows: t is t s ＝10ms，ζ＝0.5，K d ＝0.75，T s The generalized integrator parameter is a function of the resonant frequency that varies in real time with the motor speed, 40 mus. The steady state frequency offset experimental conditions were: rotating speed of 7500rpm, load rotatingMoment 2Nm, the resonance frequency transitions from 0.9 times true to 1.1 times true.
A closed loop harmonic admittance plot is plotted according to the above parameters, as shown in fig. 5, from which it can be seen that infinite harmonic impedance suppression harmonics are provided at the resonance poles. Under the condition that the resonant frequency deviates by 10%, the harmonic admittance is about-12 dB, and the harmonic suppression capability can still be provided, which indicates that the parameter selection is reasonable. The harmonic suppression effect is shown in fig. 6. The experimental result shows that the harmonic suppression effect is good within the frequency deviation range of +/-5%, and the parameters meet the requirements. As shown in FIG. 7, the harmonic amplitude decreases rapidly until it converges to zero, the convergence time and t being set s And the harmonic plane current is not influenced when the load suddenly changes, and the decoupling effect of the complex controller is good.
In conclusion, the embodiment realizes the decoupling control of the harmonic current loop, and can achieve a better harmonic suppression effect under the condition of inaccurate resonant frequency.
Although the invention herein has been described with reference to particular embodiments, it is to be understood that these embodiments are merely illustrative of the principles and applications of the present invention. It is therefore to be understood that numerous modifications may be made to the illustrative embodiments and that other arrangements may be devised without departing from the spirit and scope of the present invention as defined by the appended claims. It should be understood that various dependent claims and the features described herein may be combined in ways different from those described in the original claims. It is also to be understood that features described in connection with individual embodiments may be used in other described embodiments.

Claims (8)

1. The two-degree-of-freedom harmonic current control method of the high-speed double three-phase permanent magnet synchronous motor is characterized by comprising the following steps of:
the method comprises the following steps: carrying out vector space decoupling transformation on a phase voltage equation of the motor, and then respectively carrying out park transformation and park inverse transformation on a transformation result to respectively obtain a voltage equation under a fundamental wave synchronous rotating coordinate system and a voltage equation under a harmonic wave anti-synchronous rotating coordinate system;
step two: respectively carrying out Laplace transform on the two voltage equations obtained in the step one in a complex frequency domain to obtain motor transfer functions in the two complex frequency domains;
step three: respectively setting two motor transfer functions into standard generalized second-order systems according to a zero-pole cancellation principle, and further obtaining controller transfer functions of the two generalized second-order systems;
step four: selecting the adjusting time t according to the actual performance requirement of the motor control system s And a second-order system damping coefficient zeta, and then a gain coefficient in a transfer function of the controller is obtained;
step five: discretizing the controller transfer function with the determined gain coefficient to obtain two discrete controller transfer functions;
step six: collecting phase current of a motor, carrying out vector space decoupling transformation on the phase current to obtain current under a fundamental wave static coordinate system and current under a harmonic static coordinate system, then carrying out park transformation on the current under the fundamental wave static coordinate system to obtain current under a fundamental wave synchronous rotating coordinate system, and carrying out park inverse transformation on the current under the harmonic static coordinate system to obtain current under a harmonic anti-synchronous rotating coordinate system;
step seven: respectively inputting a given current instruction, current under a fundamental wave synchronous rotating coordinate system and current under a harmonic anti-synchronous rotating coordinate system into the discrete controllers, and then respectively carrying out park transformation and park inverse transformation with angle compensation on the two output voltages to obtain a fundamental wave static coordinate system and voltage under the harmonic static coordinate system;
step eight: and carrying out SVPWM modulation on the voltage under the fundamental wave static coordinate system and the harmonic static coordinate system to obtain a PWM signal of the six-phase inverter, and controlling the high-speed double three-phase permanent magnet synchronous motor by using the PWM signal.
2. The two-degree-of-freedom harmonic current control method of the high-speed double-three-phase permanent magnet synchronous motor according to claim 1, wherein in the step one, a voltage equation under a fundamental wave synchronous rotation coordinate system is as follows:
Figure FDA0003774984720000011
the voltage equation under the harmonic anti-synchronous rotating coordinate system is as follows:
Figure FDA0003774984720000012
wherein u is dq ＝u d +ju q ，u d And u q The voltages of the d axis and the q axis under the fundamental wave synchronous rotation coordinate system are respectively,
i dq ＝i d +ji q ，i d and i q Are d-axis current and q-axis current under a fundamental wave synchronous rotating coordinate system respectively,
L dq ＝L d +jL q ，L d and L q Are respectively the d-axis inductance and the q-axis inductance under the fundamental wave synchronous rotating coordinate system,
Figure FDA0003774984720000021
Figure FDA0003774984720000022
and
Figure FDA0003774984720000023
the voltages of the z1 axis and the z2 axis under the harmonic anti-synchronous rotating coordinate system respectively,
Figure FDA0003774984720000024
Figure FDA0003774984720000025
and
Figure FDA0003774984720000026
the current of the z1 axis and the z2 axis under the harmonic anti-synchronous rotating coordinate system respectively,
L z1z2 ＝L z1 +jL z2 ，L z1 and L z2 The inductances of the z1 axis and the z2 axis under the harmonic anti-synchronous rotating coordinate system respectively,
j is an imaginary unit, R s Is stator resistance, ω e For electrical angular frequency, # f Is a permanent magnetic linkage.
3. The two-degree-of-freedom harmonic current control method of the high-speed double-three-phase permanent magnet synchronous motor according to claim 2, wherein in the second step, the motor transfer functions in the two complex frequency domains are respectively the motor transfer functions in the fundamental wave synchronous rotation coordinate system
Figure FDA0003774984720000027
Transfer function of motor under harmonic anti-synchronous rotating coordinate system
Figure FDA0003774984720000028
The expressions for the two motor transfer functions are respectively as follows:
Figure FDA0003774984720000029
Figure FDA00037749847200000210
wherein s is a laplace operator.
4. The two-degree-of-freedom harmonic current control method of the high-speed double-three-phase permanent magnet synchronous motor according to claim 3, wherein the controller transfer functions of the two generalized second-order systems in step three are the controller transfer functions under a fundamental wave synchronous rotation coordinate system respectively
Figure FDA00037749847200000211
Transfer function of controller under harmonic anti-synchronous rotating coordinate system
Figure FDA00037749847200000212
The expressions for the two controller transfer functions are respectively as follows:
Figure FDA00037749847200000213
Figure FDA00037749847200000214
wherein the content of the first and second substances,
Figure FDA00037749847200000215
and
Figure FDA00037749847200000216
two gain coefficients under the fundamental wave synchronous rotation coordinate system,
Figure FDA00037749847200000217
and
Figure FDA00037749847200000218
two gain coefficients, omega, in harmonic anti-synchronous rotating coordinate system 1dq And ω 1zr The harmonic synchronous rotation coordinate system and the harmonic anti-synchronous rotation coordinate system are respectively the resonance frequency.
5. The method for controlling harmonic current with two degrees of freedom of a high-speed dual three-phase permanent magnet synchronous motor according to claim 4, wherein in step four, when the allowable error band of harmonic control is 2%, the gain coefficient in the transfer function of the controller is obtained by using the following formula:
Figure FDA0003774984720000031
wherein when ω is 1 ＝ω 1dq When the temperature of the water is higher than the set temperature,
Figure FDA0003774984720000032
when ω is 1 ＝ω 1zr When the temperature of the water is higher than the set temperature,
Figure FDA0003774984720000033
6. the two-degree-of-freedom harmonic current control method of the high-speed double three-phase permanent magnet synchronous motor according to claim 4 or 5, characterized in that in the fourth step, the time t is adjusted s The value range of (1) is 0.5 ms-20 ms, and the value range of the second-order system damping coefficient zeta is 0.350-0.707.
7. The two-degree-of-freedom harmonic current control method of the high-speed double-three-phase permanent magnet synchronous motor according to claim 4, wherein the concrete method for discretizing the controller transfer function with the gain coefficient determined in the step five is as follows:
adapting the controller transfer function for which the gain factor is determined to a form of a sum of polynomials:
Figure FDA0003774984720000034
Figure FDA0003774984720000035
wherein, the first and the second end of the pipe are connected with each other,
Figure FDA0003774984720000036
when ω is 1 ＝ω 1dq When the utility model is used, the water is discharged,
Figure FDA0003774984720000037
when ω is 1 ＝ω 1zr When the temperature of the water is higher than the set temperature,
Figure FDA0003774984720000038
then separately for P 1 c (ω 1 )、P 2 c (ω 1 ) And Q c (ω 1 ,K GI2 ) Discretizing, wherein expressions after discretization are respectively as follows:
Figure FDA0003774984720000039
Figure FDA00037749847200000310
Figure FDA00037749847200000311
wherein σ 1 ＝T s ω 1 ，σ 2 ＝K d T s ω 1 ，σ 3 ＝K GI2 T s ω 1 ，
Figure FDA00037749847200000312
T s To sample time, K d For the delay compensation parameter, T d ＝K d T s 。
8. The two-degree-of-freedom harmonic current control method of the high-speed double-three-phase permanent magnet synchronous motor according to claim 1, 2, 3, 4, 5 or 7, characterized in that after the step eight, a resonance frequency shift experiment under a steady state condition is performed on the high-speed double-three-phase permanent magnet synchronous motor, and the specific method of the resonance frequency shift experiment is as follows:
under the conditions of 50 percent of rated rotation speed and 50 percent of rated torque, the motor is kept rotatingThe resonance frequency omega is adjusted within the range of +/-10 percent without changing the speed 1 And testing whether the phase current harmonic wave of the motor is inhibited, if so, finishing the harmonic wave current control, otherwise, returning to the fourth step.
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