CN115097409A

CN115097409A - Doppler frequency shift self-adaptive matched filtering method based on control circuit

Info

Publication number: CN115097409A
Application number: CN202210692613.0A
Authority: CN
Inventors: 陈华; 赵乐
Original assignee: Shaanxi Changling Electronic Technology Co ltd
Current assignee: Shaanxi Changling Electronic Technology Co ltd
Priority date: 2022-06-17
Filing date: 2022-06-17
Publication date: 2022-09-23

Abstract

The invention discloses a Doppler frequency shift self-adaptive matched filtering method based on a control circuit, which mainly solves the problems of large loss of pulse compression peak or related peak amplitude and more occupied resources in Doppler frequency shift in the prior art. The scheme is as follows: the receiving end down-converts the received frequency band signal into a baseband signal, takes the sign bit of the baseband signal, and detects the edge jump of the sign bit through a designed control circuit: when the sign bit jumps along, the counter resets and starts cycle counting again, namely, sampling each bit of code element at the central moment of the counting value to obtain code element sampling points, entering a matched filter, generating code element clock enabling signals with i global clock periods, and when the signals are effective, entering the matched filter by the i sampling points of each bit of code element, and outputting a final pulse compression peak or a related peak. The pulse compression peak or related peak amplitude output by the invention has no loss, reduces the hardware resource occupation, realizes Doppler frequency shift self-adaptive matched filtering, and can be used for radar detection and communication navigation.

Description

Doppler frequency shift self-adaptive matched filtering method based on control circuit

Technical Field

The invention belongs to the technical field of signal processing, and particularly relates to a matched filtering method which can be used for radar detection and communication navigation.

Background

In radar detection, in order to solve the contradiction between the range and the range resolution of the radar, a pulse compression technology is adopted, namely a group of matched filters are adopted to compress pulses transmitted by the radar. In communication navigation, a spread spectrum technology is adopted to achieve a long range and strong anti-interference capability, and a matched filter is adopted in the spread spectrum technology to despread each bit of data.

The radar can obtain large transmission energy to reach a far action distance by transmitting a wide pulse, target detection is carried out, the received signal is processed by pulse compression to obtain a narrow pulse, and high distance resolution can be obtained. The pulse compression widely adopts a phase coding form signal, a single pulse is subjected to phase coding by using a pseudorandom sequence code at a transmitting end to form a baseband signal, namely a wide pulse with the width of T, the received baseband signal and a matching value in a matched filter are subjected to convolution processing at a receiving end, the wide pulse can be compressed into a narrow pulse, the matching value is the same prestored pseudorandom sequence code, and the narrow pulse is called a pulse compression peak. The time width T of the wide pulse is N.tau.tau is the code element bit width of the pseudo-random sequence code, N is the code length of the pseudo-random sequence code, namely the code element number, the wide pulse comprises N code elements with the width tau, the phase of each code element is taken according to different rules, if the phase is only taken to be 0 and pi, the wide pulse is a two-phase coded signal, otherwise, the wide pulse is a multiphase coded signal, and the two-phase coded signal is most widely applied. The phase encoded signal has an advantage of ensuring high range resolution and ranging accuracy, and a disadvantage of being sensitive to doppler shift.

Because of the relative motion of the radar and the target, the base band signal of the receiving end is added with Doppler frequency shift, which causes the code element bit width of the base band signal to change: when the Doppler frequency shift is positive, the frequency of the received signal is higher than the local carrier frequency, and the bit width of the code element is narrowed; when the Doppler frequency shift is negative, the frequency of the received signal is lower than the local carrier frequency, and the bit width of the code element is widened; if the bit width of the code element with the Doppler shift is tau ', the bit width difference of each code element with the Doppler shift and without the Doppler shift is tau' -tau. The matched filter prestores a group of matching value points according to the standard code element bit width, so that the base band signal with Doppler frequency shift has mismatch phenomenon with the matching value prestored in the matched filter: the larger the Doppler frequency shift is, the larger the bit width difference | tau' -tau | of each bit code element is, so that the number of malposition points is increased, and the mismatch phenomenon occurs; when the code length N is longer, the bit width difference N · | τ' - τ | of N consecutive code elements is larger, the number of misplaced points is larger, mismatch is aggravated, and the amplitude of a pulse compression peak output by the matched filter is rapidly reduced until no obvious peak exists.

In practical application, the Doppler frequency shift of a target is unknown, but the maximum range of the Doppler frequency shift is fixed, in order to overcome the influence of the Doppler frequency shift on the amplitude of a pulse compression peak, the existing method adopts a method of trial and selection of a plurality of branches, adopts a group of matched filters to work simultaneously in parallel, divides the maximum range of the Doppler frequency shift into a plurality of intervals, matched filters corresponding to a plurality of branches, each matched filter corresponding to a particular Doppler frequency, which matches the baseband signal at a particular doppler frequency, the greater the doppler shift the greater the number of branches requiring a matched filter, selecting the matched filter with the largest output pulse compression peak amplitude from the matched filters, wherein the highest point of the output pulse compression peak amplitude of the matched filter is N, the output signal-to-noise ratio is the largest, and the pulse compression of the signal is finished; however, in practice, most situations are that the doppler frequency shift of the target is not at a specific doppler frequency, but between adjacent specific doppler frequencies, which results in the loss of the amplitude of the pulse compression peak output from the matched filter, and the reduction of the amplitude directly affects the radar to find the target.

In spread spectrum communication, a transmitting end adopts a pseudorandom sequence code to perform phase coding on each transmitted bit of data to realize spread spectrum, and a receiving end needs to adopt the pseudorandom sequence code which is completely the same as that of the transmitting end to perform de-spreading on each received bit of data to recover each bit of data of the transmitting end. And despreading, carrying out correlation operation on the received data and the pseudorandom sequence code prestored in the matched filter to output a correlation peak, wherein the highest amplitude point of the correlation peak is N. The existence of the Doppler frequency shift can reduce the amplitude of the correlation peak, and when the amplitude of the correlation peak is lower than the threshold value of the decision threshold, the data cannot be recovered, and the data despreading fails.

It can be seen that, the despreading method of spread spectrum communication is similar to the pulse compression method of radar detection, and all needs to use a matched filter to complete the despreading method, but the pulse compression technique and the data despreading technique are both complex to implement, and have the following two problems:

first, there is a loss in the pulse compression peak amplitude or related peak amplitude.

In pulse compression, the maximum range f of the Doppler shift is shifted _dMAX Divided into M sections, each section covering a frequency of f _dMAX /M, each matched filter corresponding to a specific Doppler frequency, the specific Doppler frequency being f _dMAX The integer multiple of/M, when the Doppler frequency shift of the target is exactly at a specific Doppler frequency, the pulse compression peak amplitude is not lost, however, in most of the situations, the Doppler frequency shift of the target is not at the specific Doppler frequency, but in the interval of the adjacent specific Doppler frequency, the pulse compression peak amplitude is lost, and the reduction of the amplitude directly influences the radar to find the target.

In spread spectrum communication, the existence of Doppler frequency shift can reduce the amplitude of a correlation peak, and when the amplitude of the correlation peak is lower than a decision threshold value, data despreading fails.

Secondly, the calculation amount and the equipment amount are large.

In pulse compression, a method of trial and selection of a plurality of branches is adopted to overcome the influence of Doppler frequency shift on the amplitude of a phase coding pulse compression peak, a group of matched filters are required to work simultaneously in parallel, the matched filter with the largest pulse compression peak is selected as an output result, and the calculation amount and the equipment amount are large.

In spread spectrum communication, a matched filter needs to complete carrier synchronization before performing correlation operation between a pseudorandom sequence code and received data, and a receiving end needs to capture and track a carrier frequency by using an Automatic Frequency Control (AFC).

The above two disadvantages will cause the pulse compression peak amplitude/correlation peak amplitude to decrease under the scenario of large doppler shift or weak signal, until the pulse compression peak amplitude/correlation peak amplitude is lower than the threshold of decision threshold, which respectively causes that the radar cannot find the target and the data despreading fails.

Disclosure of Invention

The present invention aims to provide a doppler shift adaptive matched filtering method to overcome the above-mentioned shortcomings in the prior art, so as to reduce the loss of the amplitude of the output pulse compression peak or correlation peak, reduce the probability of radar missing target and data despreading failure, and improve the detection capability of radar and the speed of data despreading.

In order to achieve the purpose, the technical scheme of the invention comprises the following steps:

(1) designing a control circuit: the circuit comprises 4D triggers, 1 matched filter, 1 binary mode L counter, 1 pulse generator, 1 exclusive-OR gate and 1 AND gate;

(2) after the control circuit is powered on, setting the initial states of all the D triggers, the counter, the pulse generator and the matched filter to be low levels, and controlling the work of the D triggers, the counter, the matched filter and the pulse generator by a global clock CLK generated by a crystal oscillator;

(3) at a receiving end, down-converting a received frequency band signal into a baseband signal with a code length of N and a code element bit width of tau, taking a sign bit of the baseband signal, and shaping the signal waveform into a rectangular pulse signal;

(4) detecting edge jump of a baseband signal sign bit, generating a reset signal when the edge jump occurs each time, controlling a counter to reset by the reset signal, measuring each code element in the baseband signal sign bit by the counter in a sliding window mode, and restarting cycle counting;

(5) the sampling time is controlled by a counter to generate a latch pulse signal WENA, and a D trigger samples each bit of code element in the baseband signal sign bit for 1 time, namely each bit of code element obtains 1 code element sampling point;

(6) delaying the latch pulse signal by 1 global clock cycle to obtain a code element clock enable signal CLKEN, enabling the high level of the code element clock enable signal to last for i global clock cycles, and realizing sampling of i sampling points on each code element;

(7) outputting 1 sample point of each bit code element obtained in the step (5) to a code element sample point input end of a matched filter, outputting a code element clock enabling signal to a code element clock enabling end of the matched filter, and connecting a global clock to a clock end of the matched filter;

(8) according to the symbol clock enabling signal obtained in the step (6), performing the following operations on the matched filter accessed with the symbol clock enabling signal and the global clock in the step (7):

when the code element clock enable signal is in low level, the matched filter maintains the current state and does not act;

when the code element clock enabling signal is in a high level, the matched filter latches a newly input code element sampling point on the rising edge of the global clock, and performs sliding correlation on a pre-stored matching value per se and the latched code element sampling point, so that the duration of the pre-stored matching value is the same as the actual bit width of the code element sampling point and changes synchronously, the Doppler frequency shift is automatically adapted, and finally, a pulse compression peak or a correlation peak with the amplitude of i multiplied by N is output.

Compared with the prior art, the invention has the following advantages:

1) the method is automatically adaptive to Doppler frequency shift, has no loss of the output peak value of the matched filter, and is suitable for pulse compression and spread spectrum communication.

In the prior art, a baseband signal with Doppler frequency shift and a matching value in a matched filter are mismatched in an adjacent specific Doppler frequency interval, when the code length is longer, the mismatch is larger, the amplitude of a pulse compression peak or a related peak output by the matched filter is lost, and the reduction of the amplitude directly influences radar target finding and data de-spreading;

the invention adopts the counter, D trigger and other circuits, takes the edge jump as the starting time, the counter measures each code element in a sliding window mode, the D trigger is made to sample each code element in the baseband signal with Doppler shift at the central point of the counting value, not only the multiple sampling and the missing sampling of the code element sample point are avoided, but also the time length of the pre-stored matching value in the matching filter can be the same as the actual bit width of the code element sample point and synchronously changed after the code element clock enabling signal generated after the sampling enters the matching filter at the code element sample point, the Doppler shift can be automatically adapted, the pulse compression peak or the related peak amplitude value output by the matching filter is lossless, the signal to noise ratio is improved, and the invention is not only suitable for pulse compression, but also suitable for data de-spreading in spread spectrum communication.

2) The calculation amount is reduced, and hardware resources and time are saved.

In pulse compression, a group of matched filters are adopted in the prior art to work simultaneously in parallel, and the calculation amount and the equipment amount are large. In spread spectrum communication, in the prior art, an Automatic Frequency Control (AFC) is adopted at a receiving end to capture and track frequency;

the invention only adopts one matched filter to work in the pulse compression, thereby reducing the operation amount and saving the hardware resource; in spread spectrum communication, Automatic Frequency Control (AFC) is not needed for capturing and tracking the frequency, and a counter and a D trigger are only adopted for realizing synchronization, so that the speed of data despreading is increased, and the time is saved.

Drawings

FIG. 1 is a flow chart of an implementation of the present invention;

fig. 2 is a schematic diagram of a control circuit in the present invention.

Detailed Description

Embodiments of the present invention are described in detail below with reference to the accompanying drawings.

The invention is based on a control circuit, which is fed with the sign bit of the receiving end baseband signal and is driven by a global clock CLK to output a pulse compression peak or a correlation peak.

Referring to fig. 2, the control circuit includes 4D flip-flops, 1 matched filter, 1 binary analog L counter, 1 pulse generator, 1 exclusive or gate, and 1 and gate, wherein the first flip-flop D ₁ The output end of the voltage regulator is connected with the input end of an exclusive-OR gate; the output end of the exclusive-OR gate is connected with a second trigger D ₂ A data input terminal of, a second flip-flop D ₂ The output end of the counter is connected to the asynchronous reset end of the counter; the output end of the counter is connected with the input end of an AND gate, and the output ends of the AND gates are respectively connected with a third trigger D ₃ Enable terminal ena, fourth flip-flop D ₄ A data input terminal of, a fourth flip-flop D ₄ The output end of the pulse generator is connected with the data input end of the pulse generatorThe output end of the pulse generator is connected with the code element clock enabling end of the matched filter, and a third trigger D ₃ The output terminal of which is connected to the symbol sample input terminal of the matched filter.

Referring to fig. 1, the implementation steps of the present embodiment of performing doppler shift adaptive matched filtering based on the control circuit in the present embodiment are as follows:

step 1, at the receiving end, a control circuit is driven by a global clock CLK.

After the control circuit is powered on, setting the initial states of all the D triggers, the counters, the pulse generators and the matched filters to be low levels; generating a global clock CLK with a period of t by an external crystal oscillator, and outputting the global clock CLK to clock ends of all D triggers, counters, matched filters and pulse generators; under the synchronization of the global clock, the D flip-flop, the matched filter, the counter and the pulse generator inside the control circuit all start to work.

And 2, at a receiving end, taking the sign bit of the baseband signal, detecting the edge jump of the baseband signal, generating a reset signal, and using the reset signal to control the counter to reset so as to restart the cycle counting.

2.1) sampling and quantizing the received frequency band signal, and performing digital down-conversion to obtain a baseband signal a (t), wherein the code length of the baseband signal is N, the bit width of each bit is L times of the global clock period, i.e. τ ═ L × t, L is the count value of one cycle of the counter, and the sign bit af (t) in the baseband signal is taken;

2.2) using 2D flip-flops and 1 exclusive-OR gate in the control circuit to detect the edge transition of the baseband signal sign bit aF (t), wherein the detection of the edge transition takes 2 global clock cycles:

a first D flip-flop D for simultaneously inputting the sign bits aF (t) of the baseband signals into the control circuit ₁ A first input of an xor gate, a first D flip-flop D ₁ The output terminal of which is connected to the second input terminal of the xor gate, the data at the first input terminal of the xor gate always being the old data I of the sign bit af (t) each time the global clock CLK is triggered ₀ The data at the second input end is always the new data I of the sign bit aF (t) ₁ When detectingThe data output by the two D flip-flops is different, i.e. I ₀ ≠I ₁ The sign bit af (t) has an edge transition, and a second D flip-flop D connected to the xor gate ₂ Will output high level, will take 2 global clock cycles to produce edge jump to output high pulse;

2.3) triggering the second D flip-flop D ₂ The output high level is used as a reset signal to reset the counter, when the reset signal changes to low level, the counter can measure each bit code element in the baseband signal sign bit in a sliding window mode, and cycle counting is carried out from 0 to L-1 according to the beat of the global clock.

The counter in this example counts cyclically from 0 to 15 in beats of the global clock.

And 3, controlling the sampling time by a counter to generate a latch pulse signal WENA, and sampling each bit of code element by a D flip-flop for 1 time.

3.1) outputting a first counting bit r0 of the counter in an inverted way, outputting a second counting bit r1 normally, outputting a third counting bit r2 normally and outputting a fourth counting bit r3 in an inverted way, wherein the four counting bits are simultaneously connected to the input end of an AND gate;

3.2) according to the principle that the middle point value L/2 time counted by the counter corresponds to the center of each bit code element in the baseband signal sign bit, considering 2 global clock cycles consumed for detecting the transition of the baseband signal sign bit edge, taking the counting value L/2-2 time of the counter as the sampling time, and outputting a latch pulse signal WENA through an AND gate at the output value of the counter at the time; when the latch pulse signal is active, the third flip-flop D ₃ Sampling each code element in the baseband signal sign bit for 1 time, namely obtaining 1 code element sampling point for each code element, and latching the sampling points to avoid multi-sampling and missing sampling of each code element in the baseband signal sign bit.

In this example, the latch pulse signal is active and the third flip-flop D is enabled every time the count value z of the counter is (L/2) -2-6 ₃ And latching the sign bit of the baseband signal to obtain a code element sampling point.

And step 4, outputting a code element clock enable signal CLKEN.

Fourth flip-flop D ₄ Will latch the pulseThe signal is delayed by 1 global clock period, and a pulse generator connected with the signal is controlled to output a code element clock enabling signal CLKEN, the enabling signal is effective when the enabling signal is at a high level, the pulse generator enables the code element clock enabling signal to last for the time of i global clock periods, and sampling of i sampling points is achieved for each bit of code elements, wherein i is a positive integer greater than or equal to 2 and cannot be greater than a count value L of one cycle of a counter.

In this example, the high level duration of the symbol clock enable signal is 4 global clock cycles.

And 5, controlling the output of the matched filter by the code element clock enabling signal and the global clock.

Determining the working state of the matched filter according to the level of the symbol clock enable signal and the rising edge of the global clock CLK:

when the code element clock enabling signal is in low level, the matched filter maintains the current state and does not act;

when the code element clock enable signal is high level, at the rising edge of the global clock, i sample points of each code element enter a matched filter, the matched filter latches the sample points, the stored code element sample points move forward, and the code element sample points latched earliest are shifted out;

the matching filter performs sliding correlation on the existing code element sampling points and the pre-stored matching values, so that the duration of the pre-stored matching values is the same as the actual bit width of the code element sampling points and changes synchronously, thereby realizing automatic adaption to Doppler frequency shift. In this example, each bit symbol latches 4 sampling points, the code length of the baseband signal is 127, the amplitude of the pulse compression peak or the correlation peak output by the matched filter is i × N-4 × 127-508, the peak amplitude is lossless, the signal-to-noise ratio is increased by 4 times, and the doppler shift adaptive matched filtering can be realized.

The foregoing description is only an example of the present invention and should not be construed as limiting the invention, as it will be apparent to those skilled in the art that various modifications and variations in form and detail can be made without departing from the principle and structure of the invention after understanding the present disclosure and the principles, but such modifications and variations are considered to be within the scope of the appended claims.

Claims

1. A Doppler frequency shift adaptive matched filtering method based on a control circuit is characterized by comprising the following steps:

(1) designing a control circuit: the circuit comprises 4D triggers, 1 matched filter, 1 two-input mode L counter, 1 pulse generator, 1 exclusive-OR gate and 1 AND gate;

(4) detecting edge jump of a baseband signal sign bit, generating a reset signal when the edge jumps each time, controlling a counter to reset by the reset signal, measuring each code element in the baseband signal sign bit by the counter in a sliding window mode, and restarting cycle counting;

(5) the counter controls the sampling time to generate a latch pulse signal WENA, and a D trigger samples each bit of code elements in the baseband signal sign bit for 1 time, namely each bit of code elements obtains 1 code element sampling point;

2. The method of claim 1, wherein: the working process of the control circuit is as follows:

under a global clock CLK, the received baseband signal sign bit passes through a first D flip-flop D in turn ₁ An exclusive-or gate and a second D trigger D ₂ Generating a reset signal, wherein the reset signal controls the counter to reset through an asynchronous reset end of the counter so as to restart counting;

the output value of the counter passes through an AND gate, and a latch pulse signal WENA is output and passes through a fourth D trigger D ₄ And a pulse generator to output a symbol clock enable signal CLKEN;

the received baseband signal sign bit and the latch pulse signal output by the AND gate pass through a third D trigger D ₃ And then 1 symbol sample is output, the symbol sample is sent to a matched filter when a symbol clock enabling signal output by the pulse generator is effective, and the output of the matched filter is controlled by the symbol clock enabling signal and the global clock CLK.

3. The method of claim 1, wherein: (5) the sampling time is controlled by a counter to generate a latch pulse signal WENA, 2 global clock cycles consumed for detecting the transition of the sign bit edge of the baseband signal are considered according to the principle that the midpoint value L/2 time counted by the counter corresponds to the center of each bit code element in the sign bit of the baseband signal, the count value L/2-2 time of the counter is taken as the sampling time, and the latch pulse signal WENA is output by an AND gate at the output value of the counter at the time; when the latch pulse signal WENA is effective, a D trigger samples each bit code element in the baseband signal sign bit for 1 time, so that the multiple sampling and the missing sampling of each bit code element in the baseband signal sign bit are avoided, wherein L is a counting value of the counter which circulates once and is not less than 4.