CN115051570A - Controller for an inductor-capacitor resonant converter and method of operating the same - Google Patents
Controller for an inductor-capacitor resonant converter and method of operating the same Download PDFInfo
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- CN115051570A CN115051570A CN202110258340.4A CN202110258340A CN115051570A CN 115051570 A CN115051570 A CN 115051570A CN 202110258340 A CN202110258340 A CN 202110258340A CN 115051570 A CN115051570 A CN 115051570A
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
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Abstract
The invention discloses a controller applied to the primary side of an inductance-capacitance resonant converter and an operation method thereof. The controller includes a common mode voltage generating circuit and a control signal generating circuit. The common mode voltage generating circuit generates a common mode voltage. The control signal generating circuit generates an upper bridge switch control signal and a lower bridge switch control signal which respectively control an upper bridge switch and a lower bridge switch on the primary side of the inductance-capacitance resonant converter according to the compensation voltage, the sensing voltage and the common mode voltage. Therefore, because the controller controls the lc-lc converter using a current mode and the on-time of the upper bridge switch control signal is equal to the on-time of the lower bridge switch control signal, the controller may enable the lc-lc converter to have soft switching characteristics, low switching loss, and high conversion efficiency.
Description
Technical Field
The present invention relates to a controller for an inductor-capacitor resonant converter and a method for operating the same, and more particularly, to a controller for controlling an inductor-capacitor resonant converter using a current mode control method and a method for operating the same.
Background
In the prior art, a symmetric inductor-capacitor (LLC) power converter is a resonant circuit that makes the dual output voltages on the secondary side of the inductor-capacitor power converter constant by controlling the frequencies (frequency adjustments) of two power switches on the primary side of the inductor-capacitor power converter, where the inductor-capacitor power converter can make the inductor-capacitor power converter have the advantages of low switching loss and high conversion efficiency through soft switching characteristics.
However, when the lc-lc power converter is controlled in voltage mode, the transient response of the lc-lc power converter is slow, which makes the lc-lc power converter lose the above advantages. Therefore, how to improve the control method of the inductor-capacitor power converter becomes an important issue for designers of the inductor-capacitor power converter.
Disclosure of Invention
An embodiment of the present invention discloses a primary side (primary side) controller applied to an inductor-inductor-capacitor (LLC) resonant converter. The controller comprises a common mode voltage generating circuit and a control signal generating circuit. The common mode voltage generating circuit is used for generating a common mode voltage. The control signal generating circuit is configured to generate an upper bridge switch control signal and a lower bridge switch control signal according to a compensation voltage related to an output voltage of the lc-lc converter, a sensing voltage related to an input voltage of the lc-lc converter, and the common mode voltage, where the upper bridge switch control signal and the lower bridge switch control signal respectively control an upper bridge switch and a lower bridge switch on a primary side of the lc-lc converter.
Another embodiment of the present invention discloses an operating method of a controller applied to a primary side of an inductor-capacitor resonant converter, wherein the controller includes a common mode voltage generating circuit, a compensation voltage generating circuit and a control signal generating circuit. The operation method comprises the steps that the compensation voltage generation circuit generates a compensation voltage to the control signal generation circuit according to the output voltage of the inductance-capacitance resonance converter; the common mode voltage generating circuit generates a common mode voltage to the control signal generating circuit; the control signal generating circuit generates an upper bridge switch control signal and a lower bridge switch control signal according to the compensation voltage, a sensing voltage related to an input voltage of the inductor-capacitor resonant converter and the common mode voltage, wherein the upper bridge switch control signal and the lower bridge switch control signal respectively control an upper bridge switch and a lower bridge switch on a primary side of the inductor-capacitor resonant converter.
The invention provides a controller applied to an inductance-capacitance resonant converter and an operation method thereof. The controller and the operating method generate a common mode voltage by using a common mode voltage generating circuit, generate a compensation voltage according to an output voltage of the inductor-capacitor resonant converter by using a compensation voltage generating circuit, and generate an upper bridge switch control signal and a lower bridge switch control signal according to the compensation voltage, a sensing voltage related to an input voltage of the inductor-capacitor resonant converter and the common mode voltage by using a control signal generating circuit, wherein the upper bridge switch control signal and the lower bridge switch control signal respectively control an upper bridge switch and a lower bridge switch on a primary side of the inductor-capacitor resonant converter. Therefore, compared to the prior art, because the controller controls the lc-lc converter in a current mode, and the on-time of the upper bridge switch control signal is equal to the on-time of the lower bridge switch control signal, the controller may enable the lc-lc converter to have not only soft switching characteristics but also low switching loss and high conversion efficiency.
Drawings
Fig. 1 is a schematic diagram of a controller applied to the primary side of an inductor-capacitor resonant converter according to a first embodiment of the present invention.
Fig. 2 is a schematic diagram illustrating the operation of the lc tank, the primary winding, the first secondary winding, and the second secondary winding when the upper bridge switch is turned on.
Fig. 3 is a schematic diagram illustrating the operation of the lc tank, the primary winding, the first secondary winding, and the second secondary winding when the lower bridge switch is turned on.
Fig. 4 is a schematic diagram illustrating a dead time between the on-time of the upper bridge switch and the on-time of the lower bridge switch.
Fig. 5 is a schematic diagram illustrating a common mode voltage generating circuit.
Fig. 6 is a schematic diagram of a controller applied to the primary side of an inductor-capacitor resonant converter according to a second embodiment of the present invention.
Fig. 7 is a schematic diagram illustrating a common mode voltage generating circuit.
Fig. 8 is a flowchart of an operation method of a controller applied to the primary side of an inductor-capacitor resonant converter according to a third embodiment of the present invention.
Wherein the reference numerals are as follows:
100 inductor-capacitor resonant converter
101 capacitance voltage division circuit
102 upper bridge switch
104 lower bridge switch
106 inductance-capacitance resonance box
108 primary winding
110 first secondary side winding
112 second secondary side winding
120 bridge rectifier
200. 300 controller
202. 302 common mode voltage generating circuit
2022. 2064 first comparator
2024. 2066 second comparator
2026 first switch
2028 second switch
2030 a first current source
2032 second current source
2034. 3026 first capacitor
2035 third switch
2036 second capacitor
2039 Voltage-to-Current converter
2040. 3032 resistance
204 compensation voltage generating circuit
2042 compensator
2044 slope compensator
2046 adder
206 control signal generating circuit
2062 differential amplifier
2068 dead time controller
2070 upper bridge switch control signal generator
2072 lower bridge switch control signal generator
3022 first Voltage-to-Current converter
3024 second voltage-to-current converter
3030 third Voltage-to-Current converter
Cr, C1 and C2 capacitors
DT dead time
FRS first reset signal
FVCOMP first compensation voltage
HG upper bridge switch control signal
IPRI1 and IPRI2 primary side currents
IO1 first output Current
IO2 second output Current
IU charging current
ID discharge current
I1 first Current
I2 second Current
Lr inductor
LG lower bridge switch control signal
PRI Primary side
SH sampling signal
SEC Secondary side
SRS second reset signal
On time of TON1, TON2
VAC input voltage
VIN DC voltage
VOUT output voltage
VCr voltage step
Upper limit voltage of VTH
VTL lower limit voltage
VCrSEN sense Voltage
VCM common mode Voltage
VCOMP compensation voltage
VRAMP ramp voltage
First voltage of V1
Second voltage of V2
VBIAS DC bias
800 steps 806
Detailed Description
Referring to fig. 1, fig. 1 is a schematic diagram of a controller 200 applied to a primary side PRI of an inductor-inductor-capacitor (LLC) resonant converter 100 according to a first embodiment of the present invention. As shown in fig. 1, the controller 200 comprises a common mode voltage generating circuit 202, a compensation voltage generating circuit 204 and a control signal generating circuit 206, wherein the common mode voltage generating circuit 202 is coupled to a capacitor voltage dividing circuit 101 (composed of capacitors C1 and C2) of the primary-side PRI of the lc-lc converter 100, the compensation voltage generating circuit 204 is coupled to the secondary-side SEC of the lc-lc converter 100, and the control signal generating circuit 206 is coupled to the common mode voltage generating circuit 202, the compensation voltage generating circuit 204 and the primary-side PRI of the lc converter 100. In addition, the level of the ground of the primary side PRI of the lc-lc converter 100 and the level of the ground of the secondary side SEC of the lc-lc converter 100 may be the same or different.
Referring to fig. 2 and 3, fig. 2 illustrates that when an upper bridge switch 102 of the primary side PRI of the lc-lc converter 100 is turned on, a schematic diagram of the operation of an lc tank 106, a primary winding 108, a first secondary winding 110, and a second secondary winding 112, and figure 3 illustrates that when the lower bridge switch 104 of the primary side PRI of the lc-lc converter 100 is turned on, a schematic diagram of the operation of the lc tank 106, the primary winding 108, the first secondary winding 110, and the second secondary winding 112, wherein the upper bridge switch 102, the lower bridge switch 104, the lc-resonant tank 106, the primary winding 108, the first secondary winding 110, and the second secondary winding 112 are included in the lc-lc converter 100, in order to simplify fig. 1, an exciting inductor in the primary winding 108 is not shown in fig. 1. As shown in fig. 2, when the upper bridge switch 102 is turned on (the lower bridge switch 104 is turned off), the primary side current IPRI1 charges the capacitor Cr in the lc tank 106 through the upper bridge switch 102, the inductor Lr in the lc tank 106, and the primary winding 108. At this time, because the voltage polarity of the first secondary winding 110 is different from the voltage polarity of the second secondary winding 112 (as shown in fig. 1, the voltage polarity of the first secondary winding 110 is known to be different from the voltage polarity of the second secondary winding 112 through the black point position of the first secondary winding 110 and the black point position of the second secondary winding 112), only the first output current IO1 flows through the first secondary winding 110, that is, the output voltage VOUT of the secondary SEC of the lc-lc converter 100 can be generated by a dc voltage VIN, the inductor Lr, the primary winding 108, and the first secondary winding 110. In addition, the dc voltage VIN is generated by rectifying an input voltage VAC (alternating voltage) through the bridge rectifier 120. In addition, as shown in fig. 3, when the lower switch 104 is turned on (the upper switch 102 is turned off), the capacitor Cr starts to discharge, and a primary side current IPRI2 flows through the primary winding 108, the inductor Lr, and the lower switch 104. At this time, since the voltage polarity of the first secondary winding 110 is different from the voltage polarity of the second secondary winding 112, only the second output current IO2 flows through the second secondary winding 112. That is, the output voltage VOUT can be generated by the charge stored in the capacitor Cr, the inductor Lr, the primary winding 108, and the second secondary winding 112. Therefore, the voltage VCr across the capacitor Cr can be established according to the operations of fig. 2 and 3, wherein the voltage VCr is related to the dc voltage VIN and the voltage VCr is a sine wave. As shown in fig. 1, since the voltage VCr is a sine wave, a sense voltage VCrSEN generated by the capacitor voltage divider 101 according to the voltage VCr is also a sine wave and is also related to the dc voltage VIN.
In addition, as shown in fig. 4, the on-time TON1 of the upper bridge switch control signal HG is equal to the on-time TON2 of the lower bridge switch control signal LG, the upper bridge switch 102 and the lower bridge switch 104 are not turned on simultaneously, and a dead time (dead time) DT is provided between the on-time TON1 of the upper bridge switch control signal HG and the on-time TON2 of the lower bridge switch control signal LG, where HG represents an upper bridge switch control signal on the gate of the upper bridge switch 102 and LG represents a lower bridge switch control signal on the gate of the lower bridge switch 104.
Referring to fig. 5, fig. 5 is a schematic diagram illustrating the common mode voltage generating circuit 202. As shown in fig. 5, the common mode voltage generating circuit 202 includes a first comparator 2022, a second comparator 2024, a first switch 2026, a second switch 2028, a first current source 2030, a second current source 2032, a first capacitor 2034, a third switch 2035, a second capacitor 2036, a dc bias voltage VBIAS, a voltage-to-current converter 2039 and a resistor 2040, wherein the coupling relationship among the first comparator 2022, the second comparator 2024, the first switch 2026, the second switch 2028, the first current source 2030, the second current source 2032, the first capacitor 2034, the third switch 2035, the second capacitor 2036, the dc bias voltage VBIAS and the resistor 2040 can be referred to fig. 5, and will not be described herein again. As shown in fig. 5, the first comparator 2022 can control the first switch 2026 to charge the first capacitor 2034 with a charging current IU provided by the first current source 2030 according to the sensing voltage VCrSEN and a common mode voltage VCM, and the second comparator 2024 can control the second switch 2028 to discharge the first capacitor 2034 with a discharging current ID provided by the second current source 2032 according to the sensing voltage VCrSEN and the common mode voltage VCM, wherein when the charge of the first capacitor 2034 increased by the charging current IU is equal to the charge of the first capacitor 2034 decreased by the discharging current ID, a sampling signal SH controlling the third switch 2035 to be turned on enables a first voltage V1 on the first capacitor 2034 to be sampled and transferred to the second capacitor 2036. The voltage-to-current converter 2039 generates a first current I1 according to the first voltage V1, and the first current I1 and the resistor 2040 determine the common-mode voltage VCM. In addition, since the common mode voltage generating circuit 202 is well known to those skilled in the art, the present invention is not limited to the common mode voltage generating circuit 202 shown in fig. 5, that is, a circuit capable of generating the common mode voltage VCM according to the sensing voltage VCrSEN is within the scope of the present invention.
In addition, as shown in fig. 1, the compensation voltage generating circuit 204 includes a compensator 2042, a ramp compensator (ramp compensator)2044 and a summer 2046, wherein the compensator 2042 is coupled to the secondary side SEC of the lc-lc converter 100 for generating a first compensation voltage fvcmp associated with the output voltage VOUT according to the output voltage VOUT. In addition, the compensator 2042 has an isolation element for isolating the primary PRI of the lc converter 100 from the secondary SEC of the lc converter 100. In an embodiment of the invention, the isolation element is a photo coupler (photo coupler). However, the invention is not limited to the isolation element being an optocoupler, that is, in another embodiment of the invention, the isolation element may be other elements for isolating the primary-side PRI of the lc converter 100 from the secondary-side SEC of the lc converter 100. As shown in fig. 1, the adder 2046 is coupled to the compensator 2042, the slope compensator 2044 and the control signal generating circuit 206, and is configured to add the first compensation voltage fvcmp and a slope voltage VRAMP generated by the slope compensator to generate a compensation voltage VCOMP to the control signal generating circuit 206. Since the first compensation voltage fvcmp is related to the output voltage VOUT and the adder 2046 adds the first compensation voltage fvcmp and the ramp voltage VRAMP to generate the compensation voltage VCOMP, the compensation voltage VCOMP is also related to the output voltage VOUT. In addition, since the adder 2046 can add the first compensation voltage fvcmp and the ramp voltage VRAMP to generate the compensation voltage VCOMP, the compensation voltage VCOMP changes with the ramp voltage VRAMP. Therefore, the control signal generating circuit 206 can control the on-time TON1 of the upper bridge switch control signal HG and the on-time TON2 of the lower bridge switch control signal LG by the compensation voltage VCOMP varying with the ramp voltage VRAMP, that is, the control signal generating circuit 206 can control a minimum operating frequency of the lc-lc converter 100 by the compensation voltage VCOMP varying with the ramp voltage VRAMP.
In addition, as shown in fig. 1, the control signal generating circuit 206 includes a differential amplifier (differential amplifier)2062, a first comparator 2064, a second comparator 2066, a dead time (dead time) controller 2068, an upper bridge switch control signal generator 2070 and a lower bridge switch control signal generator 2072, wherein the coupling relationship among the differential amplifier 2062, the first comparator 2064, the second comparator 2066, the dead time controller 2068, the upper bridge switch control signal generator 2070 and the lower bridge switch control signal generator 2072 can refer to fig. 1, and will not be described herein again. As shown in fig. 1, the differential amplifier 2062 is coupled to the adder 2046 and the common-mode voltage generating circuit 202, and is used for generating an upper limit voltage VTH and a lower limit voltage VTL according to the compensation voltage VCOMP, the common-mode voltage VCM, and the formula (1):
VTH＝(VCOMP-VCM)×A+VCM
VTL＝VCM-(VCOMP-VCM)×A (1)
the first comparator 2064 is coupled to the differential amplifier 2062 and the capacitance voltage-dividing circuit 101, and is configured to generate a first reset signal (reset signal) FRS according to the upper limit voltage VTH and the sensing voltage VCrSEN; the second comparator 2066 is coupled to the differential amplifier 2062 and the capacitance voltage-dividing circuit 101, and is used for generating a second reset signal SRS according to the lower limit voltage VTL and the sensing voltage VCrSEN; the dead time controller 2068 is for generating a dead time DT; the upper bridge switch control signal generator 2070 is coupled to the first comparator 2064 and the dead time controller 2068, and is configured to generate an upper bridge switch control signal HG according to the first reset signal FRS and the dead time DT; and a lower bridge switch control signal generator 2072 coupled to the second comparator 2066 and the dead time controller 2068 for generating a lower bridge switch control signal LG according to the second reset signal SRS and the dead time DT, wherein the upper bridge switch control signal generator 2070 and the lower bridge switch control signal generator 2072 are SR flip flops (SR flip flops). As shown in fig. 1, since the first reset signal FRS and the dead time DT are respectively input to the R terminal and the S terminal of the upper bridge switch control signal generator 2070, the first reset signal FRS may turn off the upper bridge switch control signal HG and the dead time DT may turn on the upper bridge switch control signal HG, that is, the first reset signal FRS and the dead time DT may control the on time TON1 of the upper bridge switch control signal HG; since the second reset signal SRS and the dead time DT are input to the R terminal and the S terminal of the down-bridge switch control signal generator 2072, respectively, the second reset signal SRS may turn off the down-bridge switch control signal LG and the dead time DT may turn on the down-bridge switch control signal LG, that is, the second reset signal SRS and the dead time DT may control the on-time TON2 of the down-bridge switch control signal LG.
Therefore, the control signal generating circuit 206 may control the turning on and off of the upper bridge switch 102 and the lower bridge switch 104 of the primary side PRI of the lc-lc converter 100 by using the upper bridge switch control signal HG and the lower bridge switch control signal LG, respectively. The control method of the controller 200 controlling the lc-lc converter 100 is a current mode (current mode) control method. Because the controller 200 controls the lc-lc converter 100 by using the current mode control method, and the on-time TON1 of the upper bridge switch control signal HG is equal to the on-time TON2 of the lower bridge switch control signal LG, the controller 200 can make the lc-lc converter 100 not only have the soft switching characteristic, but also have the advantages of low switching loss and high conversion efficiency.
Referring to fig. 6, fig. 6 is a schematic diagram of a controller 300 applied to the primary side PRI of the inductor-capacitor resonant converter 100 according to a second embodiment of the present invention. As shown in fig. 6, the controller 300 is different from the controller 200 in that the controller 300 includes a control signal generating circuit 306 different from the control signal generating circuit 206. As shown in fig. 4, since the on-time TON1 of the upper bridge switch control signal HG is equal to the on-time TON2 of the lower bridge switch control signal LG, and the upper bridge switch control signal HG and the lower bridge switch control signal LG are not enabled at the same time, the control signal generating circuit 306 can generate the common mode voltage VCM according to the upper bridge switch control signal HG and the lower bridge switch control signal LG.
Referring to fig. 7, fig. 7 is a schematic diagram illustrating the common mode voltage generating circuit 302. As shown in fig. 7, the common mode voltage generating circuit 302 includes a first voltage-to-current converter 3022, a second voltage-to-current converter 3024, a first capacitor 3026, a dc bias voltage VBIAS, a third voltage-to-current converter 3030 and a resistor 3032, wherein the coupling relationship among the first voltage-to-current converter 3022, the second voltage-to-current converter 3024, the first capacitor 3026, the dc bias voltage VBIAS, the third voltage-to-current converter 3030 and the resistor 3032 can be referred to fig. 7, and will not be described herein again. As shown in fig. 7, the first voltage-to-current converter 3022 may generate the charging current IU according to the upper bridge switch control signal HG, and the second voltage-to-current converter 3024 may generate the discharging current ID according to the lower bridge switch control signal LG. In addition, as shown in fig. 4, since the upper bridge switch control signal HG and the lower bridge switch control signal LG are not simultaneously enabled, the charging current IU and the discharging current ID do not simultaneously charge and discharge the first capacitor 3026. In addition, since the on-time TON1 of the upper bridge switch control signal HG and the on-time TON2 of the lower bridge switch control signal LG are equal, the voltage on the first capacitor 3026 can be maintained at a second voltage V2. The third voltage-to-current converter 3030 then generates a second current I2 according to the second voltage V2, and the second current I2 and the resistor 3032 determine the common-mode voltage VCM. In addition, the remaining operation principle of the controller 300 is the same as that of the controller 200, and thus, the description thereof is omitted.
Referring to fig. 1, 4-8, fig. 8 is a flowchart illustrating an operation method of a controller applied to a primary side of an inductor-capacitor resonant converter according to a third embodiment of the present invention. The operation method of fig. 8 is illustrated by using the inductor-capacitor resonant converter 100 and the controller 200 of fig. 1, and the detailed steps are as follows:
step 800: starting;
step 802: the compensation voltage generation circuit 204 generates a compensation voltage VCOMP to the control signal generation circuit 206 according to the output voltage VOUT of the lc-lc resonant converter 100;
step 804: the common-mode voltage generating circuit 202 generates a common-mode voltage VCM to the control signal generating circuit 206;
step 806: the control signal generating circuit 206 generates the upper bridge switch control signal HG and the lower bridge switch control signal LG to control the upper bridge switch 102 and the lower bridge switch 104 of the primary side PRI of the lc-lc 100, respectively, according to the common mode voltage VCM and the sensing voltage VCrSEN related to the compensation voltage VCOMP and the input voltage VIN of the lc-lc 100, and jumps back to steps 802 and 804.
In step 802, as shown in fig. 1, the compensator 2042 in the compensation voltage generating circuit 204 can generate a first compensation voltage fvcmp related to the output voltage VOUT according to the output voltage VOUT. In addition, the compensator 2042 has the isolation element for isolating the primary side PRI of the lc converter 100 from the secondary side SEC of the lc converter 100. As shown in fig. 1, the adder 2046 in the compensation voltage generating circuit 204 can be used to add the first compensation voltage fvcmp and the ramp voltage VRAMP to generate the compensation voltage VCOMP to the control signal generating circuit 206. Since the first compensation voltage fvcmp is related to the output voltage VOUT and the adder 2046 adds the first compensation voltage fvcmp and the ramp voltage VRAMP to generate the compensation voltage VCOMP, the compensation voltage VCOMP is also related to the output voltage VOUT. In addition, since the adder 2046 can add the first compensation voltage fvcmp and the ramp voltage VRAMP to generate the compensation voltage VCOMP, the compensation voltage VCOMP changes with the ramp voltage VRAMP. Therefore, the control signal generating circuit 206 may control the on-time TON1 of the upper bridge switch control signal HG and the on-time TON2 of the lower bridge switch control signal LG by the compensation voltage VCOMP varying with the ramp voltage VRAMP, that is, the control signal generating circuit 206 may control the lowest operating frequency of the lc converter 100 by the compensation voltage VCOMP varying with the ramp voltage VRAMP.
In step 804, as shown in fig. 5, the first comparator 2022 may control the first switch 2026 to charge the first capacitor 2034 with the charging current IU provided by the first current source 2030 according to the sensing voltage VCrSEN and the common mode voltage VCM, and the second comparator 2024 may control the second switch 2028 to discharge the first capacitor 2034 with the discharging current ID provided by the second current source 2032 according to the sensing voltage VCrSEN and the common mode voltage VCM, wherein when the charge of the first capacitor 2034 increased by the charging current IU is equal to the charge of the first capacitor 2034 decreased by the discharging current ID, the sampling signal SH controlling the third switch 2035 to be turned on enables the first voltage V1 on the first capacitor 2034 to be sampled and transferred to the second capacitor 2036. The voltage-to-current converter 2039 may then generate a first current I1 according to the first voltage V1, and the first current I1 and the resistor 2040 may determine the common-mode voltage VCM.
In addition, in another embodiment of the present invention, as shown in fig. 6 and 7, the control signal generating circuit 306 may generate the common mode voltage VCM according to the upper bridge switch control signal HG and the lower bridge switch control signal LG. As shown in fig. 7, the first voltage-to-current converter 3022 may generate the charging current IU according to the upper bridge switch control signal HG, and the second voltage-to-current converter 3024 may generate the discharging current ID according to the lower bridge switch control signal LG. In addition, as shown in fig. 4, since the upper bridge switch control signal HG and the lower bridge switch control signal LG are not simultaneously enabled, the charging current IU and the discharging current ID do not simultaneously charge and discharge the first capacitor 3026. In addition, since the on-time TON1 of the upper bridge switch control signal HG and the on-time TON2 of the lower bridge switch control signal LG are equal, the voltage on the first capacitor 3026 can be maintained at the second voltage V2. The third voltage-to-current converter 3030 may then generate the second current I2 according to the second voltage V2, and the second current I2 and the resistor 3032 may determine the common-mode voltage VCM.
In step 806, as shown in fig. 1, the differential amplifier 2062 may generate the upper limit voltage VTH and the lower limit voltage VTL according to the compensation voltage VCOMP, the common mode voltage VCM, and equation (1); the first comparator 2064 may generate the first reset signal FRS according to the upper limit voltage VTH and the sensing voltage VCrSEN; the second comparator 2066 may generate the second reset signal SRS from the lower limit voltage VTL and the sensing voltage VCrSEN; dead time controller 2068 may be used to generate a dead time DT; the upper bridge switch control signal generator 2070 may generate an upper bridge switch control signal HG according to the first reset signal FRS and the dead time DT; and the lower bridge switch control signal generator 2072 may generate the lower bridge switch control signal LG according to the second reset signal SRS and the dead time DT. As shown in fig. 1, since the first reset signal FRS and the dead time DT are respectively input to the R terminal and the S terminal of the upper bridge switch control signal generator 2070, the first reset signal FRS may turn off the upper bridge switch control signal HG and the dead time DT may turn on the upper bridge switch control signal HG, that is, the first reset signal FRS and the dead time DT may control the on time TON1 of the upper bridge switch control signal HG; since the second reset signal SRS and the dead time DT are input to the R terminal and the S terminal of the down-bridge switch control signal generator 2072, respectively, the second reset signal SRS may turn off the down-bridge switch control signal LG and the dead time DT may turn on the down-bridge switch control signal LG, that is, the second reset signal SRS and the dead time DT may control the on-time TON2 of the down-bridge switch control signal LG.
Therefore, the control signal generating circuit 206 may control the turning on and off of the upper bridge switch 102 and the lower bridge switch 104 of the primary side PRI of the lc-lc converter 100 by using the upper bridge switch control signal HG and the lower bridge switch control signal LG, respectively. The control manner in which the controller 200 controls the inductor-capacitor resonant converter 100 is the control manner of the current mode.
In summary, the controller applied to the inductor-capacitor resonant converter and the operating method thereof provided by the present invention are to generate the common mode voltage by the common mode voltage generating circuit, generate the compensation voltage according to the output voltage by the compensation voltage generating circuit, and generate the upper bridge switch control signal and the lower bridge switch control signal according to the compensation voltage, the sensing voltage, and the common mode voltage by the control signal generating circuit, wherein the upper bridge switch control signal and the lower bridge switch control signal respectively control the upper bridge switch and the lower bridge switch. Therefore, compared with the prior art, because the controller controls the inductor-capacitor resonant converter by using the current mode control manner, and the on-time of the upper bridge switch control signal is equal to the on-time of the lower bridge switch control signal, the controller can enable the inductor-capacitor resonant converter not only to have the soft switching characteristic, but also to have the advantages of low switching loss, high conversion efficiency and the like.
The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention, and various modifications and changes may be made by those skilled in the art. Any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included in the protection scope of the present invention.
Claims (18)
1. A controller for a primary side of an inductor-capacitor resonant converter, comprising:
a common mode voltage generating circuit for generating a common mode voltage; and
a control signal generating circuit for generating an upper bridge switch control signal and a lower bridge switch control signal according to a compensation voltage related to an output voltage of the lc-lc converter, a sensing voltage related to an input voltage of the lc-lc converter, and the common mode voltage, wherein the upper bridge switch control signal and the lower bridge switch control signal respectively control an upper bridge switch and a lower bridge switch on a primary side of the lc-lc converter.
2. The controller of claim 1, further comprising:
and the compensation voltage generating circuit is coupled to the secondary side of the inductance-capacitance resonance converter and the control signal generating circuit and used for generating the compensation voltage to the control signal generating circuit according to the output voltage.
3. The controller of claim 2, wherein the compensation voltage generation circuit comprises:
a compensator, coupled to the secondary side of the lc-lc converter, for generating a first compensation voltage related to the output voltage according to the output voltage, wherein the compensator has an isolation element for isolating the primary side of the lc-lc converter from the secondary side of the lc-lc converter;
a slope compensator for generating a slope voltage; and
an adder, coupled to the compensator, the ramp compensator and the control signal generating circuit, for adding the first compensation voltage and the ramp voltage to generate the compensation voltage.
4. A controller as claimed in claim 3, wherein: the ramp voltage is used to control a minimum operating frequency of the inductor-capacitor resonant converter.
5. The controller of claim 1, wherein said control signal generating circuit comprises:
a differential amplifier coupled to the compensation voltage generating circuit and the common mode voltage generating circuit,
the common-mode voltage compensation circuit is used for generating an upper limit voltage and a lower limit voltage according to the compensation voltage and the common-mode voltage;
a first comparator, coupled to the differential amplifier, for generating a first reset signal according to the upper limit voltage and the sensing voltage;
a second comparator, coupled to the differential amplifier, for generating a second reset signal according to the lower limit voltage and the sensing voltage;
a dead time controller for generating a dead time;
an upper bridge switch control signal generator, coupled to the first comparator and the dead time controller, for generating the upper bridge switch control signal according to the first reset signal and the dead time; and
a lower bridge switch control signal generator, coupled to the second comparator and the dead time controller, for generating the lower bridge switch control signal according to the second reset signal and the dead time.
6. The controller of claim 1, wherein: the common mode voltage generating circuit generates the common mode voltage according to the sensing voltage.
7. The controller of claim 1, wherein: the common mode voltage generating circuit generates the common mode voltage according to the upper bridge switch control signal and the lower bridge switch control signal.
8. The controller of claim 1, wherein: the controller controls the inductor-capacitor resonant converter through a current mode.
9. The controller of claim 1, wherein: the upper bridge switch and the lower bridge switch are not turned on simultaneously.
10. The controller of claim 1, wherein: and a dead time is arranged between the starting time of the upper bridge switch and the starting time of the lower bridge switch, and the starting time of the upper bridge switch is equal to the starting time of the lower bridge switch.
11. A method of operating a controller for a primary side of an inductor-capacitor resonant converter, wherein the controller comprises a common-mode voltage generating circuit, a compensation voltage generating circuit, and a control signal generating circuit, the method comprising:
the compensation voltage generating circuit generates a compensation voltage according to the output voltage of the inductor-capacitor resonant converter,
generating a compensation voltage to the control signal generating circuit;
the common mode voltage generating circuit generates a common mode voltage to the control signal generating circuit; and
the control signal generating circuit generates an upper bridge switch control signal and a lower bridge switch control signal according to the compensation voltage, a sensing voltage related to an input voltage of the inductor-capacitor resonant converter and the common mode voltage, wherein the upper bridge switch control signal and the lower bridge switch control signal respectively control an upper bridge switch and a lower bridge switch on a primary side of the inductor-capacitor resonant converter.
12. The method of operation of claim 11, wherein: the compensation voltage generation circuit generating the compensation voltage includes:
a compensator in the compensation voltage generating circuit generates a first compensation voltage related to the output voltage according to the output voltage;
a slope compensator in the compensation voltage generating circuit generates a slope voltage; and
an adder in the compensation voltage generating circuit adds the first compensation voltage and the ramp voltage to generate the compensation voltage.
13. The method of operation of claim 12, wherein: the ramp voltage is used to control a minimum operating frequency of the inductor-capacitor resonant converter.
14. The method of operation of claim 11 wherein: the common mode voltage generating circuit generates the common mode voltage according to the sensing voltage.
15. The method of operation of claim 11, wherein: the common mode voltage generating circuit generates the common mode voltage according to the upper bridge switch control signal and the lower bridge switch control signal.
16. The method of operation of claim 11 wherein: the controller controls the inductor-capacitor resonant converter through a current mode.
17. The method of operation of claim 11, wherein: the upper bridge switch and the lower bridge switch are not turned on simultaneously.
18. The method of operation of claim 11 wherein: and a dead time is arranged between the starting time of the upper bridge switch and the starting time of the lower bridge switch, and the starting time of the upper bridge switch is equal to the starting time of the lower bridge switch.
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Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN117155137A (en) * | 2023-11-01 | 2023-12-01 | 艾科微电子(深圳)有限公司 | Power supply controller, asymmetric half-bridge power supply and control method |
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2021
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Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN117155137A (en) * | 2023-11-01 | 2023-12-01 | 艾科微电子(深圳)有限公司 | Power supply controller, asymmetric half-bridge power supply and control method |
| CN117155137B (en) * | 2023-11-01 | 2024-02-06 | 艾科微电子(深圳)有限公司 | Power supply controller, asymmetric half-bridge power supply and control method |
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