CN114584027A - Control method and control system of permanent magnet synchronous motor based on speed feedback variable frequency tracking - Google Patents

Control method and control system of permanent magnet synchronous motor based on speed feedback variable frequency tracking Download PDF

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CN114584027A
CN114584027A CN202210388278.5A CN202210388278A CN114584027A CN 114584027 A CN114584027 A CN 114584027A CN 202210388278 A CN202210388278 A CN 202210388278A CN 114584027 A CN114584027 A CN 114584027A
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current
obtaining
estimated
permanent magnet
magnet synchronous
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高晗璎
董垚
王文学
赵康旭
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Harbin University of Science and Technology
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Harbin University of Science and Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/13—Observer control, e.g. using Luenberger observers or Kalman filters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/72—Electric energy management in electromobility

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Abstract

A control method and a control system of a permanent magnet synchronous motor based on speed feedback frequency conversion tracking relate to the field of permanent magnet synchronous motor control. The invention aims at the problem that the estimation precision is low in the control of the permanent magnet synchronous motor without the position sensor in the prior art. The invention is based on the phase voltage u of the motorαβPhase current iαβAnd the feedback angular velocity estimate
Figure DDA0003594598450000011
Obtaining an estimated sinusoid
Figure DDA0003594598450000012
And cosine curve
Figure DDA0003594598450000013
From estimated sinusoids
Figure DDA0003594598450000014
And cosine curve
Figure DDA0003594598450000015
Obtaining an estimate of angular velocity of the rotor
Figure DDA0003594598450000016
And position angle estimate
Figure DDA0003594598450000017
Based on the angular velocity estimate
Figure DDA0003594598450000018
Position angle estimation
Figure DDA0003594598450000019
Obtaining a driving signal; the invention reduces the current error, improves the estimation precision and realizes the stable operation of the permanent magnet synchronous motor.

Description

Control method and control system of permanent magnet synchronous motor based on speed feedback variable frequency tracking
Technical Field
The invention relates to the field of permanent magnet synchronous motor control, in particular to a control method, namely a control system, of a permanent magnet synchronous motor based on speed feedback frequency conversion tracking.
Background
The PMSM has the advantages of high power output, low torque ripple, high efficiency, strong fault-tolerant capability and the like, and the driving system based on the PMSM has wide application prospect in the application fields of electric automobiles, ships, aerospace and the like which require high power and high reliability. PMSM vector control systems typically require the installation of position sensors to detect rotor position and speed information. The use of position sensors increases cost and volume while also reducing the reliability and safety of the control system. Therefore, in order to improve the reliability of the system, the position-sensorless detection technology is a feasible technical scheme and has important research significance. PMSM sensorless detection techniques generally fall into two categories: one is to estimate the position and the rotating speed of the rotor by using the salient pole effect of the motor rotor, is suitable for the high-frequency injection method of low speed and zero speed, has estimation precision independent of the speed, is insensitive to parameter change, and has a certain salient pole requirement on PMSM. In addition, the amplitude of the high-frequency interference signal must be correctly selected, otherwise, electromagnetic noise is generated, and meanwhile, the operation range and the dynamic characteristic of the high-frequency interference signal are also limited by bus voltage; the other type is a method based on a motor mathematical model and suitable for a medium-high speed region, such as an extended Kalman filtering method, a model reference method, a sliding mode observer method and the like, and estimation of the position and the rotating speed of a rotor of the method depends on the back electromotive force of the motor, so that the estimation precision is influenced, and the stability of the permanent magnet synchronous motor is influenced.
Disclosure of Invention
In order to solve the problems, the invention provides a control method, namely a control system, of a permanent magnet synchronous motor based on speed feedback frequency conversion tracking, which reduces current errors, improves estimation precision and realizes stable operation of the permanent magnet synchronous motor.
The invention provides a control method of a permanent magnet synchronous motor based on speed feedback frequency conversion tracking, which comprises the following steps:
s1, according to the phase voltage u of the motorαβPhase current iαβAnd the feedback angular velocity estimate
Figure BDA0003594598430000011
Obtaining an estimated sinusoid sin
Figure BDA0003594598430000012
And cosine curve cos
Figure BDA0003594598430000013
S2 finding the sine curve sin
Figure BDA0003594598430000014
And cosine curve cos
Figure BDA0003594598430000015
Obtaining an estimate of angular velocity of the rotor
Figure BDA00035945984300000111
And position angle estimate
Figure BDA0003594598430000016
S3, estimating value according to the angular speed
Figure BDA0003594598430000017
Position angle estimation
Figure BDA0003594598430000018
A drive signal is obtained.
Further, the sinusoid sin
Figure BDA0003594598430000019
And cosine curve cos
Figure BDA00035945984300000110
The acquisition method comprises the following steps:
s11, obtaining an angle estimation error parameter f;
s12, obtaining the current error estimation value
Figure BDA0003594598430000021
S13, estimating the error f of the angle estimation and the fed back angular velocity estimation value
Figure BDA0003594598430000022
Sum current error estimate
Figure BDA0003594598430000023
Inputting the flux linkage into a flux linkage observer to obtain a flux linkage estimated value
Figure BDA0003594598430000024
And
Figure BDA0003594598430000025
the flux linkage observer is as follows:
Figure BDA0003594598430000026
wherein the content of the first and second substances,
Figure BDA0003594598430000027
and kθF is an angle error parameter generated by the actual angle and the estimated angle;
s14 obtaining an estimated sine curve sin according to a current observer with frequency tracking and the flux linkage observer
Figure BDA0003594598430000028
And cosine curve cos
Figure BDA0003594598430000029
Further, the current observer with frequency tracking is:
Figure BDA00035945984300000210
where F () is the FVT function.
Further, the
Figure BDA00035945984300000211
And
Figure BDA00035945984300000212
the obtaining method comprises the following steps:
s121, estimating the value according to the stator current
Figure BDA00035945984300000213
Obtaining the current variable delta i, and obtaining the angular frequency omega by PI regulation of the current variable delta ifFurther obtain the stator current frequency ff；
S122, estimating the current error
Figure BDA00035945984300000214
Inputting an FVT function, wherein the FVT function is as follows:
Figure BDA00035945984300000215
wherein the content of the first and second substances,
Figure BDA00035945984300000216
further, the method for obtaining the angle error parameter f includes:
Figure BDA00035945984300000217
further, step S2 includes:
s21, establishing an extended state observer to obtain the rotor position angle estimated value
Figure BDA00035945984300000218
The state equation is as follows:
Figure BDA0003594598430000031
wherein the content of the first and second substances,
Figure BDA0003594598430000032
is the angular velocity estimation value of the motor rotor, J is the moment of inertia, P is the pole pair number, TLIs the load torque, Q is the total disturbance of the system;
s22, estimating the rotor position angle
Figure BDA0003594598430000033
As feedback to update rotor position angle error
Figure BDA0003594598430000034
The invention provides a control system of a permanent magnet synchronous motor based on speed feedback frequency conversion tracking, which comprises:
a flux linkage observer for observing the phase voltage u of the motorαβPhase current iαβAnd the feedback angular velocity estimate
Figure BDA0003594598430000035
Obtaining an estimated sinusoid sin
Figure BDA0003594598430000036
And cosine curve cos
Figure BDA0003594598430000037
An extended state observer for estimating the sine curve sin
Figure BDA0003594598430000038
And cosine curve cos
Figure BDA0003594598430000039
Obtaining an estimate of angular velocity of the rotor
Figure BDA00035945984300000310
And position angle estimate
Figure BDA00035945984300000311
A current loop module for estimating an angular velocity based on the current loop
Figure BDA00035945984300000312
Position angle estimation
Figure BDA00035945984300000313
A drive signal is obtained.
Further, the permanent magnet synchronous motor is a double three-phase permanent magnet synchronous motor.
As described above, the control method, i.e. the control system, of the permanent magnet synchronous motor based on speed feedback frequency conversion tracking according to the present invention has the following effects:
1. the control method is applied to the field of motor control in a multidimensional space, and realizes stable operation of the double three-phase permanent magnet synchronous motor based on the control without the position sensor.
2. The invention can obtain sine and cosine signals with speed and angle information through the novel flux linkage observer, introduces an estimation angle error parameter into the flux linkage observer, improves estimation precision through estimation speed feedback and reduces current error by adopting FVT tracking current frequency.
3. The novel three-order extended observer is used for processing sine and cosine signals obtained by the novel flux linkage observer, so that the position and the rotating speed of the rotor are estimated.
4. The invention is suitable for occasions with higher requirements on the reliability and the dynamic performance of the motor, such as aerospace, electric automobiles and the like.
Drawings
FIG. 1 is a block diagram of stator current frequency detection in accordance with an embodiment of the present invention;
FIG. 2 is a functional block diagram of the FVT function according to a specific embodiment of the present invention;
FIG. 3 is a schematic diagram of third order ESO based rotor position information estimation in accordance with an embodiment of the present invention;
FIG. 4 is a three-level ESO diagram of an embodiment of the present invention;
FIG. 5 is a block diagram of a position sensorless control system with velocity feedback according to an embodiment of the present invention;
FIG. 6 is a system flow diagram of an embodiment of the present invention;
FIG. 7 is a flow chart of a main routine of an embodiment of the present invention;
FIG. 8 is a flowchart of an interrupt routine in accordance with an embodiment of the present invention;
FIG. 9 is a DSP power supply circuit diagram of an embodiment of the invention;
FIG. 10 is a voltage sampling circuit diagram of an embodiment of the present invention;
FIG. 11 is an AC current sampling circuit diagram of an embodiment of the present invention;
FIG. 12 is a DC bias circuit diagram of an embodiment of the present invention;
FIG. 13 is a circuit diagram of an over-current protection circuit according to an embodiment of the present invention;
FIG. 14 is a driving circuit diagram of 2SD315AI according to an embodiment of the present invention;
FIG. 15 is a sine-cosine curve sin observed by a 1000r/min magnetic chain observer in accordance with an embodiment of the present invention
Figure BDA0003594598430000041
cos
Figure BDA0003594598430000042
FIG. 16 is a 1000r/min speed waveform of an embodiment of the present invention;
FIG. 17 is a 1000r/min angular waveform of an embodiment of the present invention;
FIG. 18 is a 1000r/min electromagnetic torque waveform of an exemplary embodiment of the present invention;
FIG. 19 is a graphical illustration of a 800r/min to 1000r/min dynamically varying speed waveform in accordance with an embodiment of the present invention;
FIG. 20 is a graph of a 800r/min to 1000r/min dynamic angle waveform according to an embodiment of the present invention;
FIG. 21 is a waveform of a 50r/min to 100r/min dynamically varying rotational speed according to an embodiment of the present invention;
FIG. 22 is a waveform of a 50r/min to 100r/min dynamic change angle of an embodiment of the present invention;
FIG. 23 is a sine and cosine curve observed by a20 r/min flux linkage observer in accordance with an embodiment of the present invention;
FIG. 24 is a20 r/min rpm waveform of an embodiment of the present invention;
FIG. 25 is a20 r/min angular waveform of an embodiment of the present invention;
FIG. 26 is a waveform of a20 r/min electromagnetic torque according to an embodiment of the present invention.
Detailed Description
The embodiments of the present invention are described below with reference to specific embodiments, and other advantages and effects of the present invention will be easily understood by those skilled in the art from the disclosure of the present specification. The invention is capable of other and different embodiments and of being practiced or of being carried out in various ways, and its several details are capable of modification in various respects, all without departing from the spirit and scope of the present invention. It is to be noted that the features in the following embodiments and examples may be combined with each other without conflict.
It should be noted that the drawings provided in the following embodiments are only for illustrating the basic idea of the present invention, and the drawings only show the components related to the present invention rather than the number, shape and size of the components in actual implementation, and the type, quantity and proportion of the components in actual implementation may be changed freely, and the layout of the components may be more complicated.
The invention provides a control method of a permanent magnet synchronous motor based on speed feedback frequency conversion tracking, wherein the permanent magnet synchronous motor is a double three-phase permanent magnet synchronous motor and comprises the following steps:
s1, according to the phase voltage u of the motorαβPhase current iαβAnd feedback angular velocity estimation
Figure BDA0003594598430000051
To obtain an estimated sinusoid sin
Figure BDA0003594598430000052
And cosine curve cos
Figure BDA0003594598430000053
The method specifically comprises the following steps:
s11, obtaining an angle estimation error parameter f;
the method for acquiring the angle error parameter f comprises the following steps:
Figure BDA0003594598430000054
wherein sin p theta and cos p theta are sine and cosine curves of actual angles, sin2 pθ+cos2p theta is 1, p is the motor pole pair number sin
Figure BDA0003594598430000055
cos
Figure BDA0003594598430000056
For estimated sine-cosine curves,. psifIs the rotor permanent magnet flux.
S12, obtaining the current error estimation value
Figure BDA0003594598430000057
The conventional current observer is:
Figure BDA0003594598430000058
in the above formula, R is stator resistance, LSIs the stator inductance and omega is the mechanical angular velocity of the motor.
In order to reduce the current error, the present embodiment uses the FVT function as the current error gain to track the current frequency, and obtains the current observer with frequency tracking as shown in equation (3):
Figure BDA0003594598430000059
where F () is the FVT function.
The above-mentioned
Figure BDA0003594598430000061
And
Figure BDA0003594598430000062
the obtaining method comprises the following steps:
s121, estimating the value according to the stator current
Figure BDA0003594598430000063
Obtaining current variable delta i, and obtaining angular frequency omega through PI regulationfFurther obtain the stator current frequency ff；
In order to track the changed frequency, the stator current frequency is detected by using a stator current frequency detection method based on a phase-locked loop shown in fig. 1, which specifically includes:
based on stator current estimates
Figure BDA0003594598430000064
Obtaining the current change amount Δ i:
Figure BDA0003594598430000065
in the above formula, the first and second carbon atoms are,
Figure BDA0003594598430000066
the estimated angle obtained for a PLL (phase locked loop) system.
The current variable delta i is regulated by PI to obtain angular frequency omegaf：
Figure BDA0003594598430000067
Further obtaining the stator current frequency ffComprises the following steps:
Figure BDA0003594598430000068
s122, estimating the current error
Figure BDA0003594598430000069
Inputting an FVT function, wherein the FVT function is as follows:
Figure BDA00035945984300000610
wherein the content of the first and second substances,
Figure BDA00035945984300000611
Kp，Krproportional gain and peak gain at the resonance frequency, ω, respectivelycTo cut off the frequency, the resonance gain and bandwidth can be influenced, ffTo track the stator current frequency. The principle of parameter adjustment isAdjustment of KrTo eliminate steady state error and adjust omegacTo suppress frequency fluctuations. In the experiment, each parameter needs to be adjusted online. To facilitate calculation and to combine the parameters of the motor and observer, ωcThe frequency range of (1) is 2000-4000, KpAnd KrThe parameters may be adjusted based on the real-time response and frequency characteristics of the observer, and appropriate parameters determined based on expected experimental results.
To accommodate frequency variations and computer digital implementations, the transfer function in the time domain must be converted to the Z domain:
Figure BDA00035945984300000612
in the above formula, ωeIs the electrical angular velocity;
and further converted into:
Figure BDA00035945984300000613
defining a state variable:
Figure BDA0003594598430000071
deriving the derivative of equation (10) to obtain the following state space equation:
Figure BDA0003594598430000072
the FVT function is obtained by equation (11):
Figure BDA0003594598430000073
based on the above analysis, the structure of the FVT function is shown in fig. 2.
S13, estimating the error f of the angle estimation and the fed back angular velocity estimation value
Figure BDA0003594598430000074
Sum current error estimate
Figure BDA0003594598430000075
Inputting the flux linkage into a flux linkage observer to obtain a flux linkage estimated value
Figure BDA0003594598430000076
And
Figure BDA0003594598430000077
the flux linkage equation of the dual three-phase PMSM is as follows:
Figure BDA0003594598430000078
and according to the voltage equation:
Figure BDA0003594598430000079
combining equation (13) and equation (14) yields an estimated sine-cosine curve:
Figure BDA00035945984300000710
estimating an error parameter f and estimating a velocity feedback value from the angle
Figure BDA00035945984300000711
The flux linkage observer is established as follows:
Figure BDA00035945984300000712
wherein the content of the first and second substances,
Figure BDA0003594598430000081
and kθAre all positive gain parametersF is an angle error parameter generated by the actual angle and the estimated angle;
s14 obtaining an estimated sine curve sin according to the following formula
Figure BDA0003594598430000082
And cosine curve cos
Figure BDA0003594598430000083
Figure BDA0003594598430000084
S2 finding the sine curve sin
Figure BDA0003594598430000085
And cosine curve cos
Figure BDA0003594598430000086
Obtaining an estimate of angular velocity of the rotor
Figure BDA0003594598430000087
And position angle estimate
Figure BDA0003594598430000088
The method specifically comprises the following steps:
s21, establishing an extended state observer to obtain the rotor position angle estimated value
Figure BDA0003594598430000089
As shown in fig. 4, the present embodiment employs a third order Extended State Observer (ESO), whose state equation is:
Figure BDA00035945984300000810
wherein the content of the first and second substances,
Figure BDA00035945984300000811
is a motor rotorSpeed, J is moment of inertia, P is the number of pole pairs, TLIs the load torque, Q is the total disturbance of the system;
s22, estimating the rotor position angle
Figure BDA00035945984300000812
As feedback to update rotor position angle error
Figure BDA00035945984300000813
Is provided with
Figure BDA00035945984300000814
Wherein L isdAnd LqD-axis and q-axis inductances when
Figure BDA00035945984300000815
When the temperature of the water is higher than the set temperature,
Figure BDA00035945984300000816
considered valid, as shown in fig. 3, the rotor position angle error is:
Figure BDA00035945984300000817
s3, estimating value according to the angular velocity
Figure BDA00035945984300000818
Position angle estimation
Figure BDA00035945984300000819
A drive signal is obtained.
As shown in fig. 5, based on the angular velocity estimation
Figure BDA00035945984300000820
Obtaining a rotation speed estimated value n, and calculating the rotation speed estimated value n and the position angle estimated value
Figure BDA00035945984300000821
Sending the obtained data into a current loop, and obtaining an estimated value n of the rotating speed and a given value n of the rotating speed*Making difference, and obtaining current through PI regulation
Figure BDA00035945984300000822
Electric current of
Figure BDA00035945984300000823
And phase voltages are obtained through PI regulation respectively, a driving signal of a driving circuit is obtained through an SVOWM modulation algorithm, SVPWM modulation is realized by controlling the on-off of a power device in the inverter, and then PMSM operation is controlled.
The invention provides a control system of a permanent magnet synchronous motor based on speed feedback frequency conversion tracking, which comprises a current inner ring and a speed outer ring as shown in figure 5, wherein the speed outer ring comprises:
a flux linkage observer for observing the phase voltage u of the motorαβPhase current iαβAnd the feedback angular velocity estimate
Figure BDA0003594598430000091
Obtaining an estimated sinusoid sin
Figure BDA0003594598430000092
And cosine curve cos
Figure BDA0003594598430000093
An extended state observer for estimating the sine curve sin
Figure BDA0003594598430000094
And cosine curve cos
Figure BDA0003594598430000095
Obtaining an estimate of angular velocity of the rotor
Figure BDA0003594598430000096
And position angle estimate
Figure BDA0003594598430000097
A current inner loop for estimating the angular velocity based on the angular velocity
Figure BDA0003594598430000098
Position angle estimation
Figure BDA0003594598430000099
A drive signal is obtained.
The modules in the current inner loop and the speed outer loop may be in the form of hardware circuits, software programs, or a combination of hardware circuits and software programs, in this embodiment,
a structural schematic diagram of the control system of this embodiment is shown in fig. 6, and a hardware circuit of the control system mainly includes a control circuit, a voltage and current sampling circuit, an overcurrent protection circuit, an IGBT drive circuit, a PMSM, and the like. The power side topological structure of the PMSM control system is in an AC-DC-AC topological form, namely, 220V power frequency alternating current is firstly processed by a rectifier bridge to obtain direct current bus voltage, and then the direct current bus voltage output by the rectifier bridge is filtered and stabilized by a non-polar filter capacitor and then is sent to a voltage type inverter. Processing the acquired phase voltage and phase current of the motor through a voltage and current sampling circuit, transmitting the processed phase voltage and phase current to a DSP control circuit, performing digital signal processing on voltage and current analog signals acquired by the sampling circuit through an ADC (analog to digital converter) conversion unit of the DSP, estimating and acquiring motor rotor position information on a DSP chip through a flux linkage observer, realizing double closed loops of rotating speed and current, and outputting the current closed loop to an SVPWM (space vector pulse width modulation) algorithm to acquire a driving signal of a driving circuit; SVPWM modulation is realized by controlling the on-off of a power device in the inverter, and then PMSM operation is controlled.
The main program of the control circuit mainly completes the contents of system initialization, interruption and the like, and the interruption program comprises AD sampling, fault diagnosis, calculation of a flux linkage observer and an ESO, a speed loop, a current loop and the like. In the main program flowchart of the system shown in fig. 7, initialization of the system is performed after a shut-off is closed at the time of starting operation of the system, and initial setting of each unit used in the program is completed. And after the initialization is finished, starting an interrupt, starting a timer and waiting for the interrupt.
The flow chart of the interrupt subroutine is shown in fig. 8, and is used for completing sampling of phase voltage and current, estimating rotor position and speed, speed loop PI regulation, current loop PI regulation and coordinate transformation, and outputting a control signal to the power module through an SVPWM algorithm so as to control the motor to operate.
The control circuit in the embodiment adopts a DSP architecture, a DSP chip selects TMS320F28335 of TI company, during the working process, the DSP chip is mainly responsible for the operation of the processed sampling signals, the extraction of instruction current, a current tracking control algorithm and a non-inductive control algorithm, the phase voltage and the phase current of the motor are processed by the voltage and current sampling circuit and are transmitted to the DSP control circuit, the voltage and current analog signals obtained by the sampling circuit are processed by the ADC conversion unit of the DSP through digital signal processing, then the position information of the motor rotor is obtained on the DSP chip through estimation of a flux linkage observer, double closed loops of rotating speed and current are realized, and the received modulation signal data is subjected to the operation of comparing the modulation wave with the carrier wave to obtain a PWM signal with a dead zone, and the PWM signal is amplified by a driving circuit and then drives a power switching tube in each phase inverter to work.
As shown in fig. 9, a DSP power supply circuit is provided, which uses a TPS767D301 chip to supply power to a DSP, outputs two stable dc voltages, supplies 1.9V to a DSP core, and supplies 3.3V dc to an I/O port.
As shown in fig. 10, a voltage sampling circuit is shown, and phase voltage and dc bus voltage are collected by an isolation operational amplifier, and the principle is that a voltage dividing resistor is used to obtain a small voltage at a high voltage side, the voltage is isolated by the isolation operational amplifier in a differential manner, the voltage is output at an output side of the isolation operational amplifier in a differential manner, and then a differential signal is converted into a single end through an operational amplifier and is transmitted to a DSP.
As shown in fig. 11, the ac current sampling circuit is used for collecting ac current of a motor by a current hall sensor, the model of the current hall sensor is HA2020 manufactured by YHDC company, the maximum sampling current value is 100A, the power supply is ± 15V, and the transformation ratio is 2000: 1.
As shown in fig. 12, a dc bias circuit is provided, the voltage amplitude of the sampled current sampling signal is limited to 0-3V by the bias circuit through the sampled potential signal, the bias voltage of 1.65V is generated by dividing the voltage by resistors R37 and R39, the resistor R39 and the capacitor C10 form a first-order RC filter circuit, and the schottky diode D2 forms a clamp circuit of the sampling voltage, so as to prevent the chip from being damaged due to too large voltage entering the DSP.
Fig. 13 shows an overcurrent protection circuit, which mainly functions to prevent the phase current of the motor from exceeding the rated value of the IGBT switch tube, causing the IGBT switch tube to be burned out, the circuit is composed of a comparator, the sampled current signal is compared with a limited value through a bias circuit, and if the voltage value of the biased current signal is higher than 4.5V or lower than 0.6V, the control system blocks the PWM output. The limit value is selected in relation to the motor rating and the gain of the sampling circuit.
The driving circuit is used for amplifying the low-level and low-power control signal output by the DSP, so that the power switching tube can be driven by the driving circuit. As shown in fig. 14, the driving circuit of this embodiment selects a driving module with a model number of 2SD315AI, which is introduced by the company condept, switzerland, and has two operating modes, i.e., a direct mode and a half-bridge mode, in which 8 pins MOD of the driver is shorted to VDD, and operates in the direct mode, at this time, channels a and B do not have a relationship, the two channels operate independently, and RC1 and RC2 are shorted to GND, and at this time, the state output SO1/SO2 also operates independently. The 8-pin MOD of the driver is in short circuit with the GND, the driver works in a half-bridge mode, dead time is generated between two channels, the dead time is adjusted by an RC (resistor-capacitor) network between pins 5 and 7, at the moment, INB is connected with high level enable, and INA is a total input end of two signals.
To further illustrate the invention, a simulation experiment is carried out on the basis of the control system, 537V of direct current bus voltage is given in a simulation model, 10kHz is selected as PWM switching frequency, 1000r/min is given as rotating speed, 20 N.m of initial load torque is suddenly increased to 40 N.m at 0.2s, and FIG. 6 is a sine and cosine curve sin observed by a flux linkage observer with speed feedback
Figure BDA0003594598430000101
cos
Figure BDA0003594598430000102
The amplitude of the curve in the graph is positive and negative 1, the phase difference is 90 degrees, compared with counter electromotive force, the influence of flux linkage and rotating speed is removed, and meanwhile speed feedback is introduced, so that the estimation is more accurate. FIGS. 16 and 17 show the rotational speed and angular waveform at 1000r/min, respectively. It can be seen from fig. 16 that the rotation speed rises faster in the starting stage, and returns to the steady state after a certain overshoot when reaching the given rotation speed, and meanwhile, the estimated rotation speed can always follow the actual rotation speed, and the closed loop effect is good in the steady state. It can also be seen from fig. 17 that the actual angle is highly consistent with the estimated angle, which can always follow the actual angle and has almost no phase delay. Fig. 18 is an electromagnetic torque waveform at 1000r/min, and it can be seen from the graph that the starting torque is large, the torque is restored to the given value after the rotating speed is stable, the load suddenly changes to 40N · m at 0.2s, the output electromagnetic torque can quickly reach the steady state and follow the load change, and the dynamic response is good.
Then, the dynamic variation simulation of the rotating speed is carried out, the given rotating speed is stepped from 800r/min to 1000r/min at medium and high speeds, the rotating speed is stepped from 50r/min to 100r/min at low speed, and the rotating speed and the angle waveform at 800r/min to 1000r/min at medium and high speeds are respectively shown in the graph 19 and the graph 20. It can be seen from fig. 19 that the rotation speed rises faster in the starting stage, and returns to the given and steady state after a certain overshoot is reached when the given rotation speed is reached, and meanwhile, the estimated rotation speed can always follow the actual rotation speed, and the closed loop effect is good in the steady state. The rotating speed given step is up to 1000r/min at 0.2s, the motor can quickly respond, the rotating speed can change along with the given value, meanwhile, the estimated rotating speed is always consistent with the actual rotating speed, and the good dynamic effect of the flux linkage observer is reflected. It can also be seen from fig. 20 that the actual angle remains synchronized with the estimated angle, which can always follow the actual angle and has almost no phase deviation. FIGS. 21 and 22 are dynamic waveforms of rotation speed and angle at low speed of 50r/min to 100r/min, respectively. It can be seen from fig. 21 that the rotational speed rises faster in the starting stage, reaches a steady state within a period of time and has less overshoot, and meanwhile, the estimated rotational speed has no deviation from the actual rotational speed, and the closed loop effect is good in the low-speed steady state. The rotating speed is given to change in steps at 0.2s, the motor can quickly respond and change along with the given value, meanwhile, the estimated rotating speed always follows the actual rotating speed, and the good dynamic effect of the flux linkage observer at low speed is reflected. It can also be seen from fig. 22 that at low speeds the actual angle still remains highly consistent with the estimated angle, which can always follow the actual angle and has almost no phase delay.
And (3) simulating static loading of the motor at low speed, starting the motor with load, setting the rotating speed to be 20r/min, setting the load torque to be 20 N.m, and setting a sine-cosine curve observed by a flux linkage observer with speed feedback at 20r/min in a graph 23. The amplitude of the curve in the graph is positive and negative 1, the phase difference of 90 degrees removes the influence of flux linkage and rotating speed in a low-speed state compared with counter electromotive force, and meanwhile, speed feedback is introduced, so that estimation is more accurate and the sine degree is higher. FIGS. 15 and 16 are rotational speed and angular waveforms at 20r/min at low speed steady state, respectively. It can be seen from fig. 24 that the rotational speed steadily and slowly rises in the starting stage, reaches a steady state within a period of time and hardly overshoots, and meanwhile, the estimated rotational speed can always follow the actual rotational speed, basically without error, and the closed loop effect is good in the steady state. The angular waveform of fig. 25 can also verify this: the estimated rotating speed always follows the actual rotating speed, and the steady state effect is good at low rotating speed. FIG. 26 is a waveform of electromagnetic torque at 20r/min, and output electromagnetic torque can follow a given value and always maintain a steady state, and also illustrates good low-speed carrying capacity.
The foregoing embodiments are merely illustrative of the principles and utilities of the present invention and are not intended to limit the invention. Any person skilled in the art can modify or change the above-mentioned embodiments without departing from the spirit and scope of the present invention. Accordingly, it is intended that all equivalent modifications or changes which may be made by those skilled in the art without departing from the spirit and scope of the present invention as defined in the appended claims.

Claims (8)

1. The control method of the permanent magnet synchronous motor based on speed feedback frequency conversion tracking is characterized by comprising the following steps:
s1, according to the phase voltage u of the motorαβPhase current iαβAnd the feedback angular velocity estimate
Figure FDA0003594598420000011
Obtaining an estimated sinusoid
Figure FDA0003594598420000012
And cosine curve
Figure FDA0003594598420000013
S2, according to the estimated sine curve
Figure FDA0003594598420000014
And cosine curve
Figure FDA0003594598420000015
Obtaining an estimate of angular velocity of the rotor
Figure FDA0003594598420000016
And position angle estimate
Figure FDA0003594598420000017
S3, estimating value according to the angular speed
Figure FDA0003594598420000018
Position angle estimation
Figure FDA0003594598420000019
A drive signal is obtained.
2. The method of claim 1, wherein the sinusoidal curve is a sinusoidal curve
Figure FDA00035945984200000110
And cosine curve
Figure FDA00035945984200000111
The acquisition method comprises the following steps:
s11, obtaining an angle estimation error parameter f;
s12, obtaining the current error estimated value according to the current observer with frequency tracking
Figure FDA00035945984200000112
S13, estimating the error f of the angle estimation and the fed back angular velocity estimation value
Figure FDA00035945984200000113
Sum current error estimate
Figure FDA00035945984200000114
Inputting the flux linkage into a flux linkage observer to obtain a flux linkage estimated value
Figure FDA00035945984200000115
And
Figure FDA00035945984200000116
the flux linkage observer is as follows:
Figure FDA00035945984200000117
wherein, the first and the second end of the pipe are connected with each other,
Figure FDA00035945984200000124
and kθF is an angle error parameter generated by the actual angle and the estimated angle;
s14 obtaining an estimated sinusoid according to a current observer with frequency tracking and the flux linkage observer
Figure FDA00035945984200000118
And cosine curve
Figure FDA00035945984200000119
3. The control method of the permanent magnet synchronous motor based on the speed feedback frequency conversion tracking according to claim 2, wherein the current observer with the frequency tracking is as follows:
Figure FDA00035945984200000120
where F () is the FVT function.
4. The method as claimed in claim 3, wherein the method comprises the step of controlling the PMSM based on the speed feedback and frequency conversion tracking
Figure FDA00035945984200000121
And
Figure FDA00035945984200000122
the obtaining method comprises the following steps:
s121, estimating the value according to the stator current
Figure FDA00035945984200000123
Obtaining the current variable delta i, and obtaining the angular frequency omega by PI regulation of the current variable delta ifFurther obtain the stator current frequency ff；
S122, estimating the current error
Figure FDA0003594598420000021
Inputting an FVT function, wherein the FVT function is as follows:
Figure FDA0003594598420000022
wherein the content of the first and second substances,
Figure FDA0003594598420000023
5. the method for controlling the permanent magnet synchronous motor based on the speed feedback frequency conversion tracking according to claim 2, wherein the method for obtaining the angle error parameter f comprises the following steps:
Figure FDA0003594598420000024
6. the method for controlling the permanent magnet synchronous motor based on the speed feedback variable frequency tracking according to claim 1, wherein the step S2 comprises:
s21, establishing an extended state observer to obtain the rotor position angle estimated value
Figure FDA0003594598420000025
The state equation is as follows:
Figure FDA0003594598420000026
wherein the content of the first and second substances,
Figure FDA0003594598420000027
is the angular velocity estimation value of the motor rotor, J is the moment of inertia, P is the pole pair number, TLIs the load torque, Q is the total disturbance of the system;
s22, estimating the rotor position angle
Figure FDA0003594598420000028
As feedback to update rotor position angle error
Figure FDA0003594598420000029
7. Permanent magnet synchronous machine's control system based on speed feedback frequency conversion tracking, its characterized in that includes:
a flux linkage observer for observing the phase voltage u of the motorαβPhase current iαβAnd the feedback angular velocity estimate
Figure FDA00035945984200000210
Obtaining an estimated sinusoid
Figure FDA00035945984200000211
And cosine curve
Figure FDA00035945984200000212
Extended state observer based on estimated sinusoids
Figure FDA00035945984200000213
And cosine curve
Figure FDA00035945984200000214
Obtaining an estimate of angular velocity of the rotor
Figure FDA00035945984200000215
And position angle estimate
Figure FDA00035945984200000216
A current loop module for estimating an angular velocity based on the current loop
Figure FDA00035945984200000217
Position angle estimation
Figure FDA00035945984200000218
A drive signal is obtained.
8. The system of claim 7, wherein the PMSM is a dual three-phase PMSM.
CN202210388278.5A 2022-04-13 2022-04-13 Control method and control system of permanent magnet synchronous motor based on speed feedback variable frequency tracking Pending CN114584027A (en)

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