CN114468393B - Constant power control circuit and method, tobacco stem and electronic cigarette - Google Patents

Constant power control circuit and method, tobacco stem and electronic cigarette Download PDF

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Publication number
CN114468393B
CN114468393B CN202210006587.1A CN202210006587A CN114468393B CN 114468393 B CN114468393 B CN 114468393B CN 202210006587 A CN202210006587 A CN 202210006587A CN 114468393 B CN114468393 B CN 114468393B
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power
load
module
signal
voltage
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CN114468393A (en
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宋利军
贺玉婷
宋朋亮
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Xi'an Wenxian Semiconductor Technology Co ltd
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Xi'an Wenxian Semiconductor Technology Co ltd
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    • A—HUMAN NECESSITIES
    • A24—TOBACCO; CIGARS; CIGARETTES; SIMULATED SMOKING DEVICES; SMOKERS' REQUISITES
    • A24F—SMOKERS' REQUISITES; MATCH BOXES; SIMULATED SMOKING DEVICES
    • A24F40/00—Electrically operated smoking devices; Component parts thereof; Manufacture thereof; Maintenance or testing thereof; Charging means specially adapted therefor
    • A24F40/50—Control or monitoring
    • A—HUMAN NECESSITIES
    • A24—TOBACCO; CIGARS; CIGARETTES; SIMULATED SMOKING DEVICES; SMOKERS' REQUISITES
    • A24F—SMOKERS' REQUISITES; MATCH BOXES; SIMULATED SMOKING DEVICES
    • A24F40/00—Electrically operated smoking devices; Component parts thereof; Manufacture thereof; Maintenance or testing thereof; Charging means specially adapted therefor
    • A24F40/40—Constructional details, e.g. connection of cartridges and battery parts
    • A—HUMAN NECESSITIES
    • A24—TOBACCO; CIGARS; CIGARETTES; SIMULATED SMOKING DEVICES; SMOKERS' REQUISITES
    • A24F—SMOKERS' REQUISITES; MATCH BOXES; SIMULATED SMOKING DEVICES
    • A24F40/00—Electrically operated smoking devices; Component parts thereof; Manufacture thereof; Maintenance or testing thereof; Charging means specially adapted therefor
    • A24F40/50—Control or monitoring
    • A24F40/53—Monitoring, e.g. fault detection

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Abstract

The invention discloses a constant power control circuit and method, a tobacco stem and an electronic cigarette, wherein the constant power control circuit comprises: the power detection module and the duty ratio adjustment module; the power detection module multiplies the detected load voltage signal representing the voltage on the load circuit and the second detected voltage signal representing the current flowing through the load to obtain a detected power signal representing the load power, the detected power signal is output to the duty ratio adjustment module, and the duty ratio adjustment module generates the on time and the on period of the switch of the load circuit according to the detected power signal to control the switch of the load circuit to be turned on and off so as to output constant power.

Description

Constant power control circuit and method, tobacco stem and electronic cigarette
Technical Field
The invention relates to the field of electronic circuits, in particular to a circuit of an electronic cigarette.
Background
An electronic cigarette is an electronic product imitating a cigarette, and comprises a cigarette rod and a cigarette bullet, which are usually separated, and the cigarette rod and the cigarette bullet are assembled together for smoking after being purchased by a consumer. The control circuit, the battery and the like are usually arranged in the tobacco stem, the atomizer is usually arranged in the tobacco cartridge, the atomizer comprises a tobacco pipe for storing tobacco tar and a heating wire for heating the tobacco tar pipe to generate smoke, and the smoke quantity is regulated by controlling the power of the heating wire.
The smoke size and taste of the electronic cigarette are strongly related to the output voltage, current and power of the control circuit, the smoke amount generally changes along with the change of the output voltage, current and power, and particularly the voltage of a battery of the electronic cigarette changes along with the use of a user, so that the smoke size and taste change, and the experience of the user is deteriorated.
Disclosure of Invention
The invention aims to solve the technical problem of providing a constant power control circuit and a constant power control method, which are used for controlling the power output from a battery to a heating wire to be constant power, so that the smoke amount is kept stable, the taste is improved, and the waste is avoided. The technical scheme is as follows:
in one aspect, the present invention provides a constant power control circuit comprising: the power detection module and the duty ratio adjustment module;
the power detection module multiplies the detected load voltage signal representing the voltage on the load circuit and the second detected voltage signal representing the current flowing through the load to obtain a detected power signal representing the load power, the detected power signal is output to the duty ratio adjustment module, and the duty ratio adjustment module generates the on time and the switching period of the switch of the load circuit according to the detected power signal to control the switch of the load circuit to be turned on and off so as to output constant power.
Preferably, the power detection module comprises a current sampling module, a voltage sampling module and a multiplier;
the input end of the voltage sampling module is electrically connected with the load circuit and used for detecting the load voltage to obtain a load voltage signal, the input end of the current sampling module is electrically connected with the load circuit and used for detecting the load current to obtain a second detection voltage signal, the output end of the voltage sampling module is electrically connected with one input end of the multiplier, the output end of the current sampling module is electrically connected with the other input end of the multiplier, and the output end of the multiplier is electrically connected with the input end of the duty ratio regulating module and used for outputting a detection power signal representing the load power to the duty ratio regulating module.
Preferably, the duty cycle adjustment module comprises an error amplifier, a PWM comparator and a driving module;
one input end of the error amplifier is electrically connected with the output end of the power detection module, and the other input end of the error amplifier receives a preset reference power signal and is used for comparing the detection power signal with the reference power signal to obtain an amplified power difference value;
one input end of the PWM comparator is electrically connected with the output end of the error amplifier, the other input end of the PWM comparator receives the sawtooth wave signal, the output end of the PWM comparator is electrically connected with the input end of the driving module, the output end of the driving module is electrically connected with the control end of the load circuit switch, and when the time points of the starting time and the switching period are reached, the PWM comparator outputs high and low levels to the driving module according to the starting time signal and the switching period signal, and the driving module controls the switching-off and the switching-on of the load circuit switch.
Preferably, the duty cycle adjustment module further comprises an RS flip-flop;
the output end of the PWM comparator is electrically connected with the R input end of the RS trigger, and the S input end of the RS trigger receives a clock pulse signal and is used for generating a switching period of a load circuit switch;
the Q output end of the RS trigger is electrically connected with the input end of the driving module, the output end of the driving module is electrically connected with the control end of the load circuit switch, and when the time points of the opening time and the switching period are reached, the RS trigger outputs high and low levels to the driving module according to the opening time signal and the switching period signal, and the driving module controls the opening and the closing of the load circuit switch.
Preferably, the relationship between the on-time and the switching period is:
the opening time is smaller than the switching period;
the on time is counted from the starting time of the switching period;
in the opening time, the load circuit switch is in a conducting state;
at a switching cycle time other than the on time, the load circuit switch is in an off state.
In another aspect, the present invention provides a constant power control method, including:
s1: detecting load voltage, converting the load voltage into an electric signal, and then obtaining a noise-removed load voltage signal through a low-pass filter;
s2: detecting load current, converting the load current into an electric signal, and then obtaining a second detection voltage signal which is used for removing noise and represents the current flowing through the load through a low-pass filter;
S3: the load voltage signal and the second detection voltage signal are subjected to a multiplier to obtain a detection power signal which can represent load power;
s4: comparing the detected power signal with a preset reference power voltage to obtain a power difference value;
s5: comparing the power difference with the sawtooth wave to generate the on time of the load circuit switch;
s6: generating a switching period of a load circuit switch, wherein the switching period is a fixed value;
s7: and controlling the duty ratio of the switch of the load circuit according to the opening time and the switching period to control the disconnection and the connection of the load circuit.
Further, step S1 further includes:
s11: different voltage sampling coefficients are obtained when the load voltage is detected so as to obtain different constant powers.
8. The constant power control method according to claim 6, wherein step S2 further comprises:
s21: the current sampling proportion of the detected load current is obtained to obtain the constant power with different settings.
9. The constant power control method according to claim 6, wherein step S4 further comprises:
s41: and amplifying the power difference value by an error amplifier to obtain the coefficient of the error amplifier to adjust the power difference value precision.
10. The constant power control method according to claim 6, wherein step S4 further comprises:
s42 is to obtain different constant powers by adjusting a preset reference power voltage.
In yet another aspect, the present invention provides a tobacco rod comprising the constant power control circuit described above.
In still another aspect, the present invention provides an electronic cigarette, including the tobacco rod described above.
The beneficial effects of the invention are as follows: by utilizing the scheme of the invention, when the electronic cigarette is in a cigarette lighting and heating state, the voltage and the current of the load are detected in real time, a detection power signal representing real-time power is obtained through multiplication operation, and a periodic duty ratio signal is generated according to the detection power signal to control the on and off of a circuit switch of an external load, so that the constant load power of the electronic cigarette is realized.
Drawings
FIG. 1 is a schematic circuit diagram of a first embodiment of an analog constant power control according to the present invention;
FIG. 2 is a schematic diagram of a load voltage and current detection module according to the present invention;
FIG. 3 is a schematic diagram of a reference power setting module according to the present invention;
FIG. 4 is a schematic diagram of a switch duty adjusting circuit according to the present invention;
FIG. 5 is a circuit schematic of a digital constant power control second embodiment of the present invention;
FIG. 6 is a schematic diagram of an ADC module according to the present invention;
FIG. 7 is a schematic diagram of a duty cycle digital processing module according to the present invention for calculating the waveform of the switching period T;
FIG. 8 is a schematic circuit diagram of a third embodiment of an analog constant power control according to the present invention;
FIG. 9A is a schematic circuit diagram of a fourth embodiment of an analog constant power control according to the present invention;
FIG. 9B is a schematic diagram of a load cell and current sensing circuit according to the present invention;
FIG. 9C is a schematic diagram of a divider circuit according to the present invention;
FIG. 9D is a schematic diagram of an integrator circuit according to the present invention;
FIG. 9E is a schematic diagram of a sawtooth generator according to the present invention;
FIG. 9F is a schematic diagram of a PWM comparator according to the present invention;
FIG. 9G is a schematic diagram of an RS flip-flop according to the present invention;
FIG. 10 is a schematic circuit diagram of a fifth embodiment of an analog constant power control according to the present invention.
Detailed Description
In order that the invention may be readily understood, a more particular description thereof will be rendered by reference to specific embodiments that are illustrated in the appended drawings. Preferred embodiments of the present invention are shown in the drawings. This invention may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided so that this disclosure will be thorough and complete.
Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this invention belongs. The terminology used in the description of the invention herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the invention.
Aiming at the problem to be solved by the invention, five constant power control circuits and methods capable of solving the technical problem are provided, wherein the first embodiment, the third embodiment, the fourth embodiment and the fifth embodiment are all analog constant power schemes, the second embodiment is a digital constant power scheme, and reference and combination can be performed among different schemes aiming at different scenes so as to achieve better effects.
Example 1
As shown in fig. 1-4, the present embodiment discloses an analog constant power control circuit including: a load voltage current detection module 11, a reference power setting module 12, a duty ratio adjustment module 13; the load voltage and current detection module 11 inputs the detected load voltage signal and the second detected voltage signal corresponding to the load current signal to the duty ratio adjustment module 13, the reference power setting module 12 inputs the reference power signal to the duty ratio adjustment module 13, and the duty ratio adjustment module 13 generates on-time and on-period of the load circuit switch according to the load voltage signal, the second detected voltage signal and the reference power signal, and controls the load circuit switch to be turned on and off so that the output power and the reference power have a proportional relationship, wherein the ratio is, for example, 1:1, 1:2, 1:3, 1:4, 2:1, 3:1, 4:1, and the like, preferably 1:1, and the output power is equal to the reference power at this time.
Preferably, for the load voltage current detection module 11, please refer to fig. 2, which is a schematic circuit diagram of a preferred load voltage current detection module 11 according to the present invention.
The load voltage and current detection module 11 comprises a sampling MOS tube M2, an intermediate MOS tube M0, a detection current operational amplifier A1, a first voltage dividing resistor R1, a second voltage dividing resistor R2 and a third conversion resistor R3; the load circuit switch comprises a Power MOS tube M1.
Specifically, the source electrode of the Power MOS tube M1 is electrically connected to a Power supply, where the Power supply may be a battery voltage, or may be a voltage obtained by converting the battery voltage, the gate electrode of the Power MOS tube M1 is electrically connected to the output end of the duty ratio adjusting module 13, the drain electrode of the Power MOS tube M1 is electrically connected to the load, and is also electrically connected to the first voltage dividing resistor R1, one end of the second voltage dividing resistor R2 is connected in series with the first voltage dividing resistor R1, and the other end of the second voltage dividing resistor R2 is grounded, and one end of the second voltage dividing resistor R2 connected to the first voltage dividing resistor R1 is also electrically connected to the duty ratio adjusting module for outputting a load voltage signal; the drain electrode of the Power MOS tube M1 is also used for externally connecting a load, the other end of the load is grounded, the load comprises an atomizer of an electronic cigarette and the like, and the Power on and Power off of the load can be realized through the Power MOS tube M1.
Specifically, the source electrode of the sampling MOS tube is electrically connected with a Power supply, the grid electrode of the sampling MOS tube is electrically connected with the grid electrode of the Power MOS tube M1, the drain electrode of the sampling MOS tube is electrically connected with the inverting input end of the detection current operational amplifier A1, the inverting input end is also electrically connected with the source electrode of the middle MOS tube M0, the grid electrode of the middle MOS tube M0 is electrically connected with the output end of the detection current operational amplifier, the drain electrode of the middle MOS tube M0 is electrically connected with the third switching resistor, the other end of the third switching resistor R3 is grounded, and the non-inverting input end of the detection current operational amplifier A1 is electrically connected with the drain electrode of the Power MOS tube M1; one end of the third converting resistor R3 connected with the middle MOS transistor M0 is further electrically connected with the duty cycle adjusting module 13, so as to output a second detection voltage signal.
Specifically, the first voltage dividing resistor R1 and the second voltage dividing resistor R2 are used for sampling the voltage division of the load voltage and outputting the load voltage signal V vsen The third converting resistor R3 is used for converting the sampling current into voltage and outputting a second detection voltage signal V isen 。
The load voltage is recorded as V load The load current is denoted as I load The load power is described in further detail belowPressure signal V vsen Second detection voltage signal V isen With load voltage V load Load current I load Is a relationship of (3).
Wherein, KI:1 is the ratio of the width-to-length ratio of the channel of the Power MOS tube M1 to the width-to-length ratio of the channel of the sampling MOS tube M2, then the load voltage and current detection module 11 circuit can obtain a real-time sampling load voltage signal V vsen And a second detection voltage signal V converted by sampling the load current in real time isen The following is shown:
thus, the load voltage signal V is obtained vsen And a second detection voltage signal V isen So that the load power can be calculated later.
Further, the sampling of the load voltage is described in detail as follows:
as shown in fig. 2, AT is a terminal to which load is externally connected.
The circuit within the dashed box on the right side of the AT is for detecting the load voltage and current.
The source electrode of the Power MOS tube M1 is connected with a Power supply VDD, the grid electrode of the Power MOS tube M1 is connected with a circuit for outputting a control PWM signal, and the drain electrode of the Power MOS tube M1 is respectively connected with a load and a voltage dividing circuit to form a parallel connection relation between the load circuit and the voltage dividing circuit. Thus, it is possible to obtain:
the above completes the sampling of the load voltage.
Next, the current sampling is described as follows:
circuit design principle: the resistances of the first voltage dividing resistor R1 and the second voltage dividing resistor R2 are extremely large, so that the current of the Power MOS transistor M1 can be regarded as equal to the load current; the Power MOS tube M1 and the sampling MOS tube M2 work in a linear region, and voltages of a source electrode, a grid electrode and a drain electrode of the Power MOS tube M1 and the sampling MOS tube M2 are respectively equal, so that currents of the Power MOS tube M1 and the sampling MOS tube are in a proportional relation.
Specifically, the resistances of R1 and R2 are much greater than the load, so that the current flowing through the Power MOS tube M1 is mainly that flowing through the load in the parallel circuit, and the current flowing through the other circuit, namely R1 and R2, is that flowing relative to the load current I lad And can be neglected. The following relationship can thus be obtained:
the current flowing through M1 is denoted as I M1 The current flowing through M2 is denoted as I M2 ，
Has I M1 ＝I load
For KI:1, which is the aspect ratio of the channels of the two mos tubes M1 and M2.
Through the parallel connection of two MOS pipes to two MOS pipes all work in the linear region, and the voltage of their three terminal is the same respectively, can ensure like this that the electric current that flows through M1 forms a invariable proportional relation with the electric current that flows through M2: i M1 ＝KI*I M2 ，
Specifically, M1 is a high-power switching tube, and M2 is a common MOS tube. This is because M1 also provides the function of supplying current to the load and controlling the switch to achieve a constant power, which requires that the current through M1 be much greater than the current through M2, i.e., KI in the above relationship is also a large value.
In addition, since the high-power switching tube works in the saturation region, the loss is large, so that the M1 is required to be set to work in the linear region so as to reduce the loss of the MOS tube.
Therefore, in order to match the currents of M2 and M1, M2 must also be operated in the linear region, and it must be ensured that both M1 and M2 have the same voltage across their three terminals. The source electrode of M2 is connected to VDD as M1, the grid electrode of M2 is connected to M1, so that the source electrode voltage of M1 is equal to the source electrode voltage of M2, the grid electrode voltage is equal, and only the drain electrode is connected to different resistors, so that the detection current operational amplifier A1 is introduced, and the effect of A1 is to ensure that the drain electrode voltage of M2 is equal to the drain electrode voltage of M1.
Binding I M1 ＝I load And I M1 ＝KI*I M2 ，
At the same time, there are:
thereby V can be obtained isen And I load Is a direct proportional relationship to:
the above completes the sampling of the load current.
Preferably, please refer to fig. 3, which is a schematic diagram of a reference power setting module 12 according to a preferred embodiment of the present invention.
The reference power setting module 12 includes a power setting operational amplifier A3, a third MOS transistor M3, a fourth MOS transistor M4, and a fourth switching resistor R4;
the non-inverting input end of the power setting operational amplifier A3 is a second reference voltage V2 generated in the constant power control circuit, the source electrode of the third MOS tube M3 is electrically connected with a power supply, the grid electrode of the third MOS tube M3 is electrically connected with the output end of the power setting operational amplifier A3, the drain electrode of the third MOS tube M3 is electrically connected with the inverting input end of the power setting operational amplifier A3 and is also electrically connected with an external power setting resistor Rset, and the other end of the external power setting resistor Rset is grounded;
the source electrode of the fourth MOS tube M4 is electrically connected with a power supply, the grid electrode of the fourth MOS tube M4 is electrically connected with the grid electrode of the third MOS tube M3, the drain electrode of the fourth MOS tube M4 is electrically connected with the fourth switching resistor R4, and the other end of the fourth switching resistor R4 is grounded;
the third MOS tube M3 and the fourth MOS tube M4 work in a saturation region, and the source voltage and the grid voltage are equal, so that the currents of the third MOS tube M3 and the fourth MOS tube M4 are in a proportional relation;
The external power setting resistor Rset is used for setting reference power and different resistancesThe external power setting resistor Rset with the value corresponds to different reference powers, the fourth conversion resistor R4 is used for converting the current flowing through the external power setting resistor Rset into voltage and outputting a second reference voltage signal V psen 。
Wherein V is 1 ，V 2 Is a first reference voltage and a second reference voltage generated inside a constant power control circuit, R set An external power setting resistor Rset for setting reference power, and a fourth conversion resistor with a resistance value R 4 Resistance value R of the third switching resistor 3 And when layout is performed, the match is considered, and the components with the same model and specification are selected.
Set V ref1 ＝V 1 The second reference voltage signal V can be obtained by the circuit of the reference power setting module 12 psen With external power setting resistor R set The relationship of (2) is as follows:
further, the reference power setting module 12 circuit is described in detail as follows:
v1 and V2 are the power supply VDD converted by the internal voltage conversion module and are fixed values.
A2 and A3 are two identical operational amplifiers, the output voltage being equal to the input voltage.
Because the detection current does not need to be supplied to an external load, the detection current can be set to a small value, and therefore, the two MOS tubes in the circuit can adopt common MOS tubes which are marked as M3 and M4 and all work in a saturation region, and because the MOS tubes in the saturation region only ensure Vgs, namely the voltage difference between the grid electrode and the source electrode of the MOS tube is equal, the currents flowing through the two MOS tubes can be ensured to form a certain proportional relation:
I M3 ＝K P *I M4
P set The terminal of the external power setting resistor Rset has the following relation:
thus, according to the three formulas:
the sampling can be realized by adopting a common MOS tube, setting the MOS tube to be in a saturation region and ensuring that the voltage difference between a grid electrode and a source electrode is equal. In this way, an operational amplifier is saved compared to the circuit of the voltage current detection module, since there is no need to adjust the drain voltages to equalize them.
Preferably, referring to fig. 1, the duty cycle adjustment module 13 includes a detection power multiplier 1301, an on-time adjustment module 1303, a switching period adjustment module 1304, and a driving module 1305;
the input end of the detection power multiplier 1301 is electrically connected with the output end of the load voltage and current detection module 11, the output end of the detection power multiplier 1301 is electrically connected with the input end of the on-time adjustment module 1303, and the detection power multiplier 1301 is used for applying the load voltage signal V vsen And a second detection voltage signal V isen Converting into detection power;
the output end of the on-time adjusting module 1303 is electrically connected with the input end of the driving module 1305, and is used for calculating the on-time of the load circuit switch according to the detected power, and outputting an on-time signal to the driving module 1305, and the driving module 1305 controls the load circuit switch to be turned off;
The input end of the switching period adjusting module 1304 is electrically connected with the output end of the reference power setting module 12, the output end of the switching period adjusting module 1304 is electrically connected with the input end of the driving module 1305, and is used for calculating the switching period of the load circuit switch according to the reference power, outputting a switching period signal to the driving module 1305, and controlling the switch of the load circuit to be turned on by the driving module 1305;
the switching period adjusting module 1304 is further electrically connected with the on-time adjusting module 1303, and the switching period adjusting module 1304 outputs a control signal according to the switching period signal to control the switching period adjusting module 1304 and the on-time adjusting module 1303 to reset at the end of each period;
the output end of the driving module 1305 is electrically connected to the gate of the Power MOS transistor M1 of the load voltage and current detecting module 11 (i.e. the control end of the load circuit switch), so as to control the turn-off and turn-on of the load circuit switch.
Preferably, referring to fig. 3, the reference power setting module 12 further includes a reference voltage operational amplifier A2, wherein the non-inverting input terminal of the reference voltage operational amplifier A2 is a first reference voltage generated inside the constant power control circuit, and the output terminal of the reference voltage operational amplifier A2 is electrically connected to the inverting input terminal for converting the first reference voltage into a reference voltage and outputting a first reference voltage signal V ref1 ；
Referring to fig. 1 and 4, the duty cycle adjustment module 13 further includes a reference power multiplier 1302; an input end of the reference power multiplier 1302 is electrically connected with an output end of the reference power setting module 12, and an output end of the reference power multiplier 1302 is electrically connected with an input end of the switching period adjusting module 1304;
specifically, the reference power multiplier 1302 has the same specification as the detection power multiplier 1301, and is configured to convert the first reference voltage signal and the second reference voltage signal into adjusted reference power, so as to offset the temperature and the process angle deviation of the delay multiplier amplification factor caused by the detection power multiplier 1301.
As can be seen from the above, the duty cycle adjustment module 13 mainly performs the following functions:
converting the output Multi1 of the detection Power multiplier 1301 into current, charging the capacitor C, and generating Ton, namely the on time of the Power tube;
converting the output Multi2 of the reference Power multiplier 1302 into a current, charging the capacitor C, generating T, i.e., the switching period of the Power tube;
the duty cycle is the ratio of on time to switching period.
The driving module 1305 controls the Power MOS switching period to be T and the on time to be Ton according to the on time and the switching period obtained above.
Preferably, please refer to fig. 4, which is a schematic diagram of a preferred duty cycle adjustment circuit according to the present invention.
The on-time adjusting module 1303 includes a detection power operational amplifier A4, a fifth MOS transistor M5, a sixth MOS transistor M6, a fifth switching resistor R5, a first switching capacitor C1, a first switch K1, a second switch K2, a first comparator Comp1, and a common RS trigger;
the non-inverting input end of the detection power operational amplifier A4 is electrically connected with the output end of the detection power multiplier 1301, the source electrode of the fifth MOS tube M5 is electrically connected with the power supply, the grid electrode of the fifth MOS tube M5 is electrically connected with the output end of the detection power operational amplifier A4, the drain electrode of the fifth MOS tube M5 is electrically connected with the inverting input end of the detection power operational amplifier A4 and is also electrically connected with the fifth converting resistor R5, and the other end of the fifth converting resistor R5 is grounded;
the source electrode of the sixth MOS tube M6 is electrically connected with a power supply, the grid electrode of the sixth MOS tube M6 is electrically connected with the grid electrode of the fifth MOS tube M5, the drain electrode of the sixth MOS tube M6 is electrically connected with the first conversion capacitor C1 through the first switch K1, the other end of the first conversion capacitor C1 is grounded, and the second switch K2 is connected with the first conversion capacitor C1 in parallel;
the non-inverting input end of the first comparator Comp1 is electrically connected with the first conversion capacitor C1, the inverting input end of the first comparator Comp1 is an internal third reference voltage, and the output end of the first comparator Comp1 is electrically connected with the common RS trigger;
The on-time adjusting module 1303 charges the first conversion capacitor C1 by converting the output of the detected power multiplier 1301 into a current, generates a high-low level flip-flop in combination with the first comparator Comp1, and inputs a high level or a low level to the common RS flip-flop, thereby generating an on-time of the load circuit switch.
Preferably, the switching period adjusting module 1304 includes a reference power operational amplifier A5, a seventh MOS transistor M7, an eighth MOS transistor M8, a sixth switching resistor R6, a second switching capacitor C2, a third switch K3, a fourth switch K4, and a second comparator Comp2;
the non-inverting input end of the reference power operational amplifier A5 is electrically connected with the output end of the reference power setting module 12, the source electrode of the seventh MOS tube M7 is electrically connected with the power supply, the grid electrode of the seventh MOS tube M7 is electrically connected with the output end of the reference power operational amplifier A5, the drain electrode of the seventh MOS tube M7 is electrically connected with the inverting input end of the reference power operational amplifier A5 and is also electrically connected with the sixth switching resistor R6, and the other end of the sixth switching resistor R6 is grounded;
the source electrode of the eighth MOS tube M8 is electrically connected with a power supply, the grid electrode of the eighth MOS tube M8 is electrically connected with the grid electrode of the seventh MOS tube M7, the drain electrode of the eighth MOS tube M8 is electrically connected with the second conversion capacitor C2 through the third switch K3, the other end of the second conversion capacitor C2 is grounded, and the fourth switch K4 is connected with the second conversion capacitor C2 in parallel;
The non-inverting input end of the second comparator Comp2 is electrically connected with the second conversion capacitor C2, the inverting input end of the second comparator Comp2 is an internal third reference voltage, and the output end of the second comparator Comp2 is electrically connected with the shared RS trigger;
the standard power operational amplifier A5 and the detection power operational amplifier A4 have the same specification, the second comparator Comp2 and the first comparator Comp1 have the same specification, the sixth conversion resistor R6 and the seventh conversion resistor have the same specification, and the second conversion capacitor C2 and the first conversion capacitor C1 have the same specification;
the on-time adjusting module 1303 charges the second conversion capacitor C2 by converting the output of the reference power multiplier 1302 into a current, and generates a high-low level flip-flop in combination with the second comparator Comp2, and inputs a high level or a low level to the common RS flip-flop, thereby generating a switching period of the load circuit switch;
the output end of the second comparator Comp2 is further electrically connected to the first switch K1, the second switch K2, the third switch K3 and the fourth switch K4, so as to control the disconnection and connection of the four switches according to the generated high-low level inversion signals, thereby controlling the charge and discharge of the second conversion capacitor C2 and the first conversion capacitor C1. The first switch K1 and the third switch K3 are turned on and off simultaneously, the second switch K2 and the fourth switch K4 are turned on and off simultaneously, and the first switch K1 and the third switch K3 are not turned on simultaneously with the second switch K2 and the fourth switch K4.
Preferably, the relationship between the on-time and the switching period is:
the opening time is smaller than the switching period;
the on time is counted from the starting time of the switching period;
in the opening time, the load circuit switch is in a conducting state;
at a switching cycle time other than the on time, the load circuit switch is in an off state.
The acquisition of the duty cycle related parameters is described in detail below:
as shown in fig. 4, the multiplier 1 and the multiplier 2 are two identical multipliers, and the amplification factor is a. Two identical multipliers are arranged and can be used for counteracting the deviation of the amplification factor of a single multiplier along with the temperature and the process angle, so that the accuracy of the whole circuit system is improved.
A4 and A5 are two identical operational amplifiers, comp1 and comp2 are two identical comparators. R5 and R6 are equal and layout, and C1 and C2 are equal and layout.
The duty cycle adjusting module 13 circuit can obtain the low level time Ton and the period T of the Driver signal as follows:
wherein by setting R 5 ＝R 6 ，C 1 ＝C 2 The duty can be obtained as:
and according to the four voltage signals V in the load voltage current detection module 11 circuit and the reference power setting module 12 circuit vsen ，V isen ，V psen ，V ref1 The duty ratio duty can be obtained as:
Wherein, by setting R 3 ＝R 4 The power can be obtained as:
the circuit can set the parameters as constant values during initial design, and the circuit is specifically as follows:
V 1 ，V 2 is an internal reference voltage, is a fixed value;
R 1 ，R 2 is a matching resistor and is designed to be fixed in proportion, and therefore,is also a constant value; k (K) p The current mirror ratio is also a constant value;
R set the resistor is an external resistor, and is a fixed value once set;
therefore, it is known from the above formula that the output power P is a constant value.
And the resistor R can be set by electrically connecting external power with different resistance values set Different reference powers and thus different output powers P can be obtained.
Further, how the duty cycle adjustment circuit switch control section gets Ton and the period T is described in detail below:
regarding the MOS transistors M5, M6, M7, M8:
the duty ratio duty adjusting circuit belongs to an internal circuit, and the current of the duty ratio duty adjusting circuit does not need to be supplied to an external load, so the duty ratio duty adjusting circuit can be set to a small value, and therefore 4 MOS tubes in the duty ratio duty adjusting circuit can be common MOS tubes, marked as M5, M6, M7 and M8 and all work in a saturation region, and the current flowing through the two MOS tubes can be guaranteed to be in a certain proportional relation as long as the MOS tubes in the saturation region guarantee Vgs, namely the voltage difference of the grid electrode and the source electrode of the MOS tubes is equal:
I M5 ＝K P *I M6
I M 7＝K P*I M 8
In the present duty cycle adjusting circuit, however, K P =1, therefore there is:
I M5 ＝I M6
I M7 ＝I M8
regarding the switches K1, K2, K3, K4:
the four switches are controlled to be turned on and off by the output of the comparator comp 2;
wherein K1 and K3 are conducted simultaneously to charge C1 and C2;
k2 and K4 are simultaneously conducted to discharge C1 and C2, wherein the discharge is instantaneous discharge, and the time is negligible compared with the charging time.
Thus, q=c=u=i×t
Where C is unchanged, I is a constant value, and as time t increases, U increases accordingly.
The charging duration of C1 is smaller than C2, so that the comparator comp1 connected with C1 is inverted every time.
When C2 is full, comp2 is reversed, and meanwhile, the input level of comp2 is also subjected to switch control, namely when the input level is reversed to a high level, K1 and K3 are controlled to be simultaneously turned off, K2 and K4 are simultaneously turned on, so that fully charged C1 and C2 are simultaneously discharged. Since the discharge is instantaneous, the time is negligible with respect to the charge duration. Therefore, comp2 is immediately inverted when discharging is finished, the low-level control K1 and K3 are released to be simultaneously turned on, and the low-level control K2 and K4 are simultaneously turned off, so that C1 and C2 start to be charged simultaneously, and the next period starts.
By the above cyclic reciprocation, the charging period of C1 can be set to Ton, and the charging period of C2 can be set to T.
These two times are input to the drive by a common RS flip-flop, which in this embodiment is composed of nor gates and is inverted by an inverter in the drive, so as to be input to M1 in fig. 2, and control the on-off of M1.
How Ton, T is related to the sampled voltage current and the set reference power is obtained as follows:
since q=c=u=i×t:
I M5 ＝I M6 and I M7 ＝I M8 ，
Then there are:
thus, there are:
similarly, the following can be obtained:
thus, there are:
in another aspect, the present invention provides a method of constant power control, comprising:
s1: detecting load voltage and load current, converting into load voltage signal V vsen And a second detection voltage signal V isn Then obtaining a detection power signal representing the real-time load power through a detection power multiplier 1301;
s2: setting a reference power, converting it into a voltage signal to obtain a second reference voltage signal V representing the reference power psen ；
S3: converting the detected power signal into current, charging the capacitor, and generating the on-time of the load circuit switch;
s4: converting a second reference voltage signal representing the reference power into a current, charging the capacitor, and generating a switching period of the load circuit switch;
s5: and controlling the duty ratio of the switch of the load circuit according to the opening time and the switching period to realize the disconnection and the connection of the load circuit.
Further, step S2 further includes:
s21: setting a reference voltage, converting into a voltage signal V ref1 Thereafter, the reference power signal representing the reference power is passed through a reference power multiplier 1302 together with the second reference voltage signal.
Further, step S2 further includes:
s22: the step of setting the reference power includes replacing an external power setting resistor RsetR set Different external power setting resistors RsetR set Representing different reference powers.
Preferably, the load voltage signal V vsen And real-time load voltage V load The relation of (2) is:
wherein R1 and R2 are voltage dividing resistors.
Preferably, the second detection voltage signal V isen And real-time load current I load The relation of (2) is:
where R3 is a conversion resistor for converting load current into a voltage signal and KI is a current mirror ratio.
Preferably, the second reference voltage signal V psen With an external power setting resistor RsetR set The relation of (2) is:
where V2 is a reference voltage inside the circuit, R 4 Is a switching resistor, K p Is the current mirror ratio.
Preferably, the on time is:
wherein R is 5 For the conversion resistance, C 1 Is a capacitor, V 3 For the internal reference voltage, a is the amplification factor.
Preferably, the switching period is:
wherein R is 6 For the conversion resistance, C 2 Is a capacitor, V 3 For the internal reference voltage, a is the amplification factor.
Further, the duty cycle is:
according to four voltage signals V vsen ，V isen ，V psen ，V ref1 The duty can be obtained as:
wherein, by setting R 3 ＝R 4 The power can be obtained as:
the circuit may be set as follows in the initial design:
V 1 ，V 2 is an internal reference voltage, is a fixed value;
R 1 ，R 2 is a matching resistor and is designed to be fixed in proportion, and therefore,is also a constant value; k (K) p The current mirror ratio is also a constant value;
R set the resistor is an external resistor, and is a fixed value once set;
therefore, the output power P is ensured to be a constant value, and constant power output is realized.
The acquisition of parameters in the above method is described in detail in the previous circuit introduction, and is not described here again.
In still another aspect, the present invention provides a tobacco stem, including the constant power control circuit of any one of the above and an external power setting resistor, wherein the external power setting resistor is electrically connected to the reference power setting module.
In still another aspect, the invention provides an electronic cigarette comprising the tobacco rod.
It can be seen that the scheme of this embodiment can realize the following functions: after the circuits are connected and electrified, according to the four voltage signals V in the circuits of the load voltage and current detection module 11 and the reference power setting module 12 vsen ，V isen ，V psen ，V ref1 The Power MOS switch is input into the duty ratio adjusting module 13, and then the duty ratio adjusting module 13 generates the communication time Ton and the period T of the control Driver, and finally the Power MOS switch is controlled by the drive Driver module to ensure that the output Power P is a fixed value.
And the resistor R can be set by adjusting external power set Different reference powers are set, and an external resistor is simply replaced, so that the requirements of equipment of different types on power can be met, and the production and assembly costs are saved.
All the circuits are realized by adopting analog circuits, multiple digital-to-analog conversion or analog-to-digital conversion is not needed, and components and parts are saved, so that the circuit is economical and practical.
Example two
As shown in fig. 5, the present embodiment discloses a digital constant power control circuit, including: the load voltage and current detection module 21, the reference power setting module 22, the analog-to-digital conversion module 23 and the duty ratio digital processing module 24;
the load voltage and current detection module 21 inputs the detected load voltage signal Vvsen and the second detected voltage signal Visen representing the load current signal to the analog-to-digital conversion module 23, the reference power setting module 22 inputs the second reference voltage signal representing the reference power and the first reference voltage signal to the analog-to-digital conversion module 23, the analog-to-digital conversion module 23 converts the load voltage signal, the second detected voltage signal, the second reference voltage signal and the first reference voltage signal into digital signals and inputs them to the duty ratio digital processing module 24, the duty ratio digital processing module 24 generates the on-time and the on-off period of the load circuit switch, and controls the load circuit switch to be turned on and off so that the output power is in a proportional relation with the reference power, wherein the ratio is, for example, 1:1, 1:2, 1:3, 1:4, 2: 1. 3:1, 4:1, etc., preferably 1:1, where the output power is equal to the reference power.
Referring to fig. 2-3, the control principle of the load voltage and current detection module 21 and the reference power setting module 22 is the same as that of the analog circuit in the first embodiment, and will not be described herein.
Preferably, referring to fig. 5-6, the analog-to-digital conversion module 23 includes a selector 2301, an analog-to-digital converter 2302;
the input end of the selector 2301 is electrically connected with the load voltage and current detection module 21 and the reference power setting module 22 respectively, the control end of the selector is electrically connected with the output end of the duty ratio digital processing module 24, the duty ratio digital processing module 24 controls the selector 2301 to output a load voltage signal, a second detection voltage signal, a second reference voltage signal and a first reference voltage signal in a time-sharing manner, and the output end of the selector 2301 is electrically connected with the input end of the analog-to-digital converter 2302 to output the load voltage signal, the second detection voltage signal, the second reference voltage signal and the first reference voltage signal to the analog-to-digital converter 2302 in a time-sharing manner;
the input end of the analog-to-digital converter 2302 is electrically connected to the output end of the duty digital processing module 24, the output end of the analog-to-digital converter 2302 is electrically connected to the input end of the duty digital processing module 24, and is used for converting the load voltage signal, the second detection voltage signal, the second reference voltage signal and the first reference voltage signal into digital signals and outputting the digital signals to the duty digital processing module 24, and the sampling quantization index of the digital signals is adjusted according to the sampling quantization index control instruction of the duty digital processing module 24.
Specifically, for the ADC 2302, please refer to fig. 6, which is a schematic diagram of a preferred ADC circuit according to the present invention.
The analog-to-digital converter ADC adopts a successive approximation method by sampling value V of an input signal sh Sequentially comparing the logic output with the reference voltage value generated by the DAC conversion network, and sequentially generating logic output from the highest bit to the lowest bit.
Wherein V is in Is the output of MUX, is the load voltage signal V vsen Second detection voltage signal V isen Second reference voltage signal V psen First reference voltage signal V ref1 One of them; v (V) ref Is the reference voltage of ADC, V vsen ，V isen ，V psen ，V ref1 The specific acquisition manner of (a) is the same as that of the first embodiment, and will not be described in detail herein.
Taking a 10-bit analog-to-digital converter ADC as an example, 10 bits represent the 10 th power of 2, i.e. the sampling quantization index is 1024V psen (D) Represents V psen Quantized values after ADC conversion, i.e. V ref Divided into 1024 parts, each having a voltage value of Vref/1024, the Vref value being fixed, V psen The fraction is the quantized value of Vpsen, which is expressed by Vpsen (D), and is specifically shown as follows:
v can be obtained by the same method vsen ，V isen ，V ref1 Is shown below:
preferably, the input end of the duty ratio digital processing module 24 is electrically connected with the output end of the analog-to-digital conversion module 23, and the output end of the duty ratio digital processing module 24 is also electrically connected with the selector 2301 and the analog-to-digital converter 2302;
The duty cycle digital processing module 24 employs a fixed on time T on The switching period adjusting module 2401 generates a switching period T according to the sampled load voltage signal, the sampled second detection voltage signal, the sampled second reference voltage signal and the sampled quantized value of the first reference voltage signal, and outputs an on-time signal and a switching period signal to control the on-off of the switch of the load circuit;
the duty cycle digital processing module 24 also generates a duty cycle signal based on the on time T on And a switching period T outputs a selection control instruction, controlling the selector 2301 to output one of the load voltage signal, the second detection voltage signal, the second reference voltage signal, and the first reference voltage signal to the analog-to-digital converter 2302;
preferably, the relationship between the on-time and the switching period is:
the opening time is smaller than the switching period;
the on time is counted from the starting time of the switching period;
in the opening time, the load circuit switch is in a conducting state;
at a switching cycle time other than the on time, the load circuit switch is in an off state.
Specifically, for the switching period adjustment module 2401, please refer to fig. 7, which is a schematic diagram of the waveform principle of the switching period T obtained by the switching period adjustment module 2401 according to a preferred embodiment of the present invention.
This control scheme uses a fixed Ton time, e.g., 512 clk, based on sampled V psen (D)/V vsen (D)/V isen (D)/V ref1 (D) Value, to adjust T time.
In order to multiplex the analog-to-digital converter ADC, a time-division sampling method is used.
As shown in fig. 7, when adc_ch=0, sample V psen (D) And only adopt each time of enablingThe sample is once, since the power is only equal to R set In relation, once R set Setting, the power is constant, and only one sampling is needed.
When adc_ch=1/2/3, sample V respectively vsen (D)/V isen (D)/V ref1 (D)。
Specifically, the following method for obtaining the required switching period T according to the circuit setting when the Ton time is set to 512 clk is fixed:
assuming that the required switching period T is N clk, there are:
meanwhile, the voltage and current values of the load can be obtained through back-pushing, and the voltage and current values are as follows:
in addition, because the power is constant, the following relationship can be constructed:
wherein K is 4 For a fixed value, the power P and an external resistor R set In an inverse proportional relationship. Then according to V psen Quantized value of (2)The following relationship can be obtained:
/>
in addition, according to the definition of power, there is the following relationship:
thereby obtaining the following steps:
wherein the following relationships are included:
then from the above formulae it is possible to obtain a composition containing V psen (D)/V vsen (D)/V isen (D)/V ref1 (D) N value of (2):
in the above, V 2 ，KI，V ref1 ，K p ，K 4 ，R 1 ，R 2 Are all constant values, and can be obtained according to calculation And the ratio is adjusted to the N value in proportion, and finally a fixed P value is obtained, so that constant power is realized.
In another aspect, the present invention provides a constant power control method, including:
s1: detecting load voltage V load And load current I load Converted into a load voltage signal V vsen And a second detection voltage signal V isen ；
S2: setting a reference power and a reference voltage, converting the reference power and the reference voltage into a voltage signal to obtain a second reference voltage signal V representing the reference load power psen And a first reference voltage signal V characterizing a reference voltage ref1 ；
S3: to load voltage signal V vsen And a second detection voltage signal V isen Second reference voltage signal V psen And a first reference voltage signal V ref1 Time-sharing sampling is converted into a digital signal;
s4: setting a fixed on time according to the load voltage signal V vsen And a second detection voltage signal V isen Second reference voltage signal V psen And a first reference voltage signal V ref1 The converted digital signal adjusts the switching period;
s5: and controlling the duty ratio of the switch of the load circuit according to the opening time and the switching period to realize the disconnection and the connection of the load circuit.
Further, the load voltage signal V vsen With load voltage V load The relation is:
wherein R1 and R2 are first voltage dividing resistor and second voltage dividing resistor;
when the quantization index of the sample is 1024, V vsen The quantized values of (2) are:
wherein V is ref Is the reference voltage at the time of analog-to-digital conversion.
Further, a second detection voltage signal V isen And I load The load current relationship is:
the third conversion resistor R3 is used for converting load current into voltage signals, and KI is a current mirror proportion;
when the quantization index of the sample is 1024, V isen The quantized values of (2) are:
wherein V is ref Is the reference voltage at the time of analog-to-digital conversion.
Further, the relationship between the second reference voltage signal and the external power setting resistor is:
wherein V2 is a second reference voltage inside the circuit, R 4 Is the fourth switching resistor, K p Is the current mirror ratio;
wherein, when the sampling quantization index is 1024, V psen The quantized values of (2) are:
output a first reference voltage signal V ref1 The quantized values of (2) are:wherein V is ref Is the reference voltage at the time of analog-to-digital conversion.
Further, the on time is set to 512 clk, the switching period is N clk, and the duty ratio is:
further, the load power is set toWherein K is 4 Is of a fixed value, R set Setting a resistor for external power, the switching period is as follows:
wherein V is 2 For an internal second reference voltage, R 1 ，R 2 Is a first voltage dividing resistor, a second voltage dividing resistor, K p For the current mirror ratio, R set Setting a resistor for external power; thus V 2 ，KI，V ref1 ，K p ，K 4 ，R 1 ，R 2 Are all constant values, and can be obtained according to calculationAnd the ratio is adjusted to the N value in proportion, and finally a fixed P value is obtained, so that constant power is realized.
The acquisition of parameters in the above method is described in detail in the previous circuit introduction, and is not described here again.
In still another aspect, the present invention provides a tobacco stem, including the constant power control circuit of any one of the above and an external power setting resistor, wherein the external power setting resistor is electrically connected to the reference power setting module.
In still another aspect, the invention provides an electronic cigarette comprising the tobacco rod.
It can be seen that the scheme of this embodiment can realize the following functions:
after the circuits are connected and electrified, according to four voltage signals V in the circuits of the load voltage and current detection module 21 and the reference power setting module 22 vsen ，V isen ，V psen ，V ref1 The output power P is input to the duty ratio digital processing module 24, and then the on-time Ton and the switching period T are generated by the circuit of the duty ratio digital processing module 24, so that the on-off of the load circuit is controlled in real time, and the output load power P is ensured to be a constant value.
And the resistor R can be set by adjusting external power set Different reference powers are set without modifying other circuit parameters and configurations. Therefore, the power requirements of equipment of different types can be met by simply replacing an external resistor, and the production and assembly costs are saved.
The analog-to-digital conversion module 23 and the duty ratio digital processing module 24 are implemented using digital control circuits, unlike the analog control circuit of the first embodiment. Including selector 2301MUX, analog-to-Digital converter 2302ADC, and Digital.
In the scheme, the selector is controlled to output the collected voltage, current and power related parameter values one by one in a time sharing way, and an analog-to-digital converter 2302 is used for carrying out analog-to-digital conversion on the parameter values. Thus, the number of components such as the analog-to-digital converter 2302 can be reduced, and the cost can be reduced.
The scheme adopts fixed Ton time, such as 512 clk, according to samplingV of (2) psen (D)/V vsen (D)/V isen (D)/V ref1 (D) The value is used for adjusting the T time, the more accurate adjustment time can be obtained through the scheme, in addition, the quantization bit number and the fixed Ton time of the analog-to-digital converter can be modified, and various methods for adjusting the T time can be obtained.
In the embodiment, only one selector and one analog-to-digital converter are needed, and a plurality of analog-to-digital converters are not needed, so that the cost for realizing the constant power of the electronic cigarette can be reduced.
Example III
As shown in fig. 8, the present embodiment discloses an analog constant power control circuit. Comprising the following steps: the load voltage and current detection module 31 detects the power multiplier 32 and the duty cycle adjustment module 33.
The load voltage and current detection module 31 inputs the detected load voltage signal and a second detection voltage signal corresponding to the load current signal to the detection power multiplier 32, multiplies the load voltage signal and the second detection voltage signal to generate a detection power signal, and outputs the detection power signal to the duty cycle adjustment module 33, and the duty cycle adjustment module 33 generates on-time and on-off period of the load circuit switch according to the detection power signal, and controls the load circuit switch to conduct and close, so as to output with constant power.
Preferably, the load voltage and current detection module 31 includes a voltage sampling module 3101 and a current sampling module 3102;
the implementation of the load voltage and current detection module 31 and the detection power multiplier 32 is similar to that of the first embodiment, and will not be repeated here.
The difference between the present embodiment and the first embodiment is mainly that the duty ratio adjusting module 33, referring to fig. 8, preferably, the duty ratio adjusting module 33 includes an on-time adjusting module, a switching period adjusting module, and a driving module 3305;
the input end of the on-time adjusting module is electrically connected with the output end of the detection power multiplier 32, the output end of the on-time adjusting module is electrically connected with the input end of the driving module 3305, the on-time adjusting module is also electrically connected with the switching period adjusting module, and is used for calculating the on-time of the load circuit switch according to the detection power signal and the control signal of the switching period adjusting module, outputting the on-time signal to the driving module 3305, and controlling the load circuit switch to be disconnected by the driving module 3305;
The output end of the switching period adjusting module is electrically connected with the input end of the driving module 3305 and is used for calculating the switching period of the load circuit switch and outputting a switching period signal to the driving module 3305, the driving module 3305 controls the load circuit switch to be turned on, the switching period adjusting module also outputs a control signal according to the switching period signal and controls the switching period adjusting module and the starting time adjusting module to reset the state when each period is finished;
the output end of the driving module 3305 is electrically connected with the Power MOS transistor gate of the load voltage and current detecting module 31 (the Power MOS transistor is also used as a load circuit switch here), so as to control the disconnection and connection of the load circuit switch.
Preferably, the on-time adjusting module includes a first voltage controlled current source i1, a first conversion capacitor C1, a first switch K1, a first comparator 3301, a first RS flip-flop 3303;
the control end of the first voltage-controlled current source i1 is electrically connected with the output end of the detection power multiplier 32, the output current of the first voltage-controlled current source i1 is in direct proportion to the output voltage of the detection power multiplier 32, the output end of the first voltage-controlled current source i1 is electrically connected with the first conversion capacitor C1, the other end of the first conversion capacitor C1 is grounded, the first voltage-controlled current source i1 is used for charging the first conversion capacitor C1, and the first switch K1 is connected with the first conversion capacitor C1 in parallel and is used for controlling charging and discharging of the first conversion capacitor C1;
The non-inverting input end of the first comparator 3301 is electrically connected with the first conversion capacitor C1, the inverting input end of the first comparator 3301 is connected with a first internal reference voltage REF, the output end of the first comparator 3301 is electrically connected with the R input end of the first RS trigger 3303, the Q output end of the first RS trigger 3303 is electrically connected with the driving module 3305, the Q/output end of the first RS trigger 3303 is electrically connected with the control end of the first switch K1, and the S input end of the first RS trigger 3303 is electrically connected with the switching period adjusting module;
the on-time adjusting module charges the first conversion capacitor C1 by converting the output of the detected power multiplier 32 into current through the first voltage-controlled current source i1, and generates high-low level inversion in combination with the first comparator 3301, and inputs high level or low level to the first RS trigger 3303, thereby generating on-time of the load circuit switch;
when the time reaches the opening time, the Q output end of the first RS trigger 3303 sends an electric signal to the driving module 3305, so that the disconnection of the load circuit switch is controlled;
when the time reaches the starting time, the Q/output end output signal of the first RS trigger 3303 controls the first switch K1 to be turned on, the first conversion capacitor C1 is discharged, the first comparator 3301 generates high-low level turnover again after the instantaneous discharge is finished, and the Q/output end output signal of the first RS trigger 3303 controls the first switch K1 to be turned off; or after the instant discharging is finished, the first switch K1 is kept on until the S input end of the first RS trigger 3303 receives a signal, and the Q/output end of the first RS trigger 3303 outputs a signal to control the first switch K1 to be turned off.
Preferably, the switching period adjusting module includes a reference power setting module (refer to fig. 3 of the first embodiment), a second voltage-controlled current source i2, a second conversion capacitor C2, a second switch K2, a second comparator 3302, and a second RS flip-flop 3304;
the control end of the second voltage-controlled current source i2 is electrically connected with the output end of the reference power setting module, the output current of the second voltage-controlled current source i2 is in direct proportion to the output voltage of the reference power setting module, the output end of the second voltage-controlled current source i2 is electrically connected with the second conversion capacitor C2, the other end of the second conversion capacitor C2 is grounded and used for charging the second conversion capacitor C2, and the second switch K2 is connected with the second conversion capacitor C2 in parallel and used for controlling the charging and discharging of the second conversion capacitor C2;
the non-inverting input end of the second comparator 3302 is electrically connected with the second conversion capacitor C2, the inverting input end of the second comparator 3302 is connected with the first internal reference voltage REF, the output end of the second comparator 3302 is electrically connected with the S input end of the second RS trigger 3304, and the Q output end of the second RS trigger 3304 is electrically connected with the control end of the second switch K2;
the output end Q of the second RS trigger 3304 is also electrically connected with the S input end of the first RS trigger 3303 of the starting time adjusting module, and is also electrically connected with the R input end of the second RS trigger 3304 through a rising edge trigger pulse generator;
The switching period adjusting module charges the second conversion capacitor C2 through the second voltage-controlled current source i2, generates high-low level turnover by combining the second comparator 3302, and inputs high level or low level to the second RS trigger 3304 so as to generate a switching period of the load circuit switch;
when the time reaches the switching period, the Q output end of the second RS trigger 3304 sends out an electric signal to the S input end of the first RS trigger 3303, and the Q output end of the first RS trigger 3303 is controlled to send out an electric signal to the driving module 3305, so that the conduction of the load circuit switch is controlled to start the next period;
when the time reaches the switching period, the output signal of the Q output end of the second RS trigger 3304 controls the second switch K2 to be turned on, the second conversion capacitor C2 is discharged, the second comparator 3302 generates high-low level turnover again after the instantaneous discharge is finished, the second switch K2 is controlled to be turned off through the Q output signal of the second RS trigger 3304, or a signal output by a rising edge trigger pulse generator is waited to be input to the R input end of the second RS trigger 3304, and the output signal of the Q output end of the second RS trigger 3304 is triggered again to control the second switch K2 to be turned off;
when the time reaches the switching period, the Q output terminal of the second RS flip-flop 3304 outputs a signal to trigger the rising edge trigger pulse generator to generate a trigger pulse, which is input to the R input terminal of the second RS flip-flop 3304, and resets the state of the second RS flip-flop 3304. The reset method of the second RS flip-flop 3304 is not limited to this.
Preferably, the second voltage-controlled current source i2 has the same specification as the first voltage-controlled current source i1, the second comparator 3302 has the same specification as the second comparator 3302, and the second conversion capacitor C2 has the same specification as the first conversion capacitor C1;
preferably, the load voltage and current detection module comprises a voltage sampling module and a current sampling module;
the input end of the voltage sampling module is electrically connected with the load circuit for detecting the load voltage, the input end of the current sampling module is electrically connected with the load circuit for detecting the second detection voltage signal, the output end of the voltage sampling module is electrically connected with the input end of the detection power multiplier, the output end of the current sampling module is electrically connected with the input end of the detection power multiplier for inputting the detected load voltage signal and the second detection voltage signal into the detection power multiplier, and the load voltage and current detection module in the first embodiment can be referred to for specific.
Preferably, the relationship between the on-time and the switching period is:
the opening time is smaller than the switching period;
the on time is counted from the starting time of the switching period;
in the opening time, the load circuit switch is in a conducting state;
at a switching cycle time other than the on time, the load circuit switch is in an off state.
The following detailed description of the principles and parameters is provided:
in this embodiment, in the on-state, the load voltage signal is preferably a BAT voltage, the BAT voltage and the BAT current are sampled, and the product is then generated by generating i1 through a voltage-controlled current source, and charging C1 to generate a sawtooth Ramp, comparing the sawtooth Ramp with the first internal reference voltage REF through a PWM comparator, and then generating a duty ratio duty to control the Powermos transistor connected to the load circuit.
Further, according to the circuit arrangement, and since q=c=u=i×t, the following relationship can be obtained:
further, the parameters are described as follows:
wherein T is on Is on time, T is switching period, duty is Duty cycle, P is load power;
C1/C2 is a proportional quantity, R2/R1 is a proportional quantity, REF is an internal fixed reference voltage, REF1 is an input voltage of a voltage-controlled current source generating I2, BAT is a load voltage, I is a load current, K1 is a voltage division coefficient of a detected load voltage, K2 is a current conversion coefficient after the detected load current is converted into a voltage, and sampling accuracy depends on K1 and K2.
Thus, different constant powers can be set by modifying the values of K1 and K2.
Further, the following details the logic of the operation of the circuit:
The output Q output of the following RS flip-flop: when Ton time is up, the upper drive DR is turned off; the load current becomes 0, and at the moment, the Q/output end enables the output current of the first voltage-controlled current source i1 to be 0 no matter the C1 is switched on or off;
after reaching period T, the following flip-flop Q has the following three output lines:
1. discharging the capacitor C2;
2. triggering the rising edge trigger pulse generator to delay one pulse time to generate a reset pulse signal, outputting the reset pulse signal to the R input end of the trigger, enabling the Q output end of the trigger to be inverted again, inverting the C2 switch below, and discharging only one pulse time after discharging;
3. the signal is sent to the S input end of the first RS trigger, so that the driving module DR conducts a switch of the load circuit and starts to provide load current; at the same time, the first switch K1 is also opened.
When the first switch K1 is turned off, the start time of Ton is also the start time of T.
In another aspect, the present invention provides a constant power control method, including:
s1: detecting the load voltage BAT and the load current I, converting the load voltage BAT and the load current I into a load voltage signal and a second detection voltage signal representing the load current signal, and obtaining a detection power signal representing real-time load power through a detection power multiplier 32;
S2: converting the detected power signal into current i1 through a voltage-controlled current source, charging a capacitor C1, and comparing the detected power signal with an internal fixed reference voltage REF by combining a comparator so as to generate the starting time of a load circuit switch;
s3: generating a current i2 through a voltage-controlled current source, charging a capacitor C2, and comparing the capacitor C2 with an internal fixed reference voltage REF through a comparator, thereby generating a switching period of a load circuit switch;
s4: and controlling the duty ratio of the switch of the load circuit according to the opening time and the switching period to realize the disconnection and the connection of the load circuit.
Further, step S1 further includes:
s11: the voltage division coefficient K1 of the detection load voltage and the current conversion coefficient K2 after the detection load current is converted into voltage are modified, so that different reference powers are set.
Further, on time T on The calculation method comprises the following steps:
wherein C1 is a capacitor, R1 is a resistor of a voltage-controlled current source, REF is an internal fixed reference voltage, REF 1 is an input voltage of the voltage-controlled current source generating i2, BAT is a load voltage, K1 is a voltage division coefficient of a detected load voltage, and K2 is a current conversion coefficient after detecting that the load current is converted into a voltage.
Further, the switching period T calculating method includes:
Wherein C2 is a capacitor, R2 is a resistor of a voltage-controlled current source, REF is an internal fixed reference voltage, REF1 isGenerating the input voltage of the voltage controlled current source of i 2.
Further, the Duty ratio Duty calculation method is as follows:
the load power P is:
wherein C1/C2 is a proportional quantity, R2/R1 is a proportional quantity, REF is an internal fixed reference voltage, REF1 is an input voltage of a voltage-controlled current source generating I2, BAT is a load voltage, I is a load current, K1 is a voltage division coefficient of a detected load voltage, K2 is a current conversion coefficient after the detected load current is converted into a voltage, and sampling accuracy depends on K1 and K2.
The acquisition of parameters in the above method is described in detail in the previous circuit introduction, and is not described here again.
In still another aspect, the present invention further provides a tobacco stem, including a constant power control circuit and an external power setting resistor, where the external power setting resistor is electrically connected to the reference power setting module.
In still another aspect, the invention provides an electronic cigarette comprising the tobacco rod.
It can be seen that the scheme of this embodiment can realize the following functions: after the circuit is connected and electrified, the load Power is sampled and valued to a certain extent according to the load voltage detection module, the current sampling module 3102 and the multiplier, and is input into the duty ratio adjusting module 33, then the on-time Ton and the period T for controlling the Driver are generated by the circuit of the duty ratio adjusting module 33, and finally the Power MOS switch is controlled by the Driver module, so that the output Power P is ensured to be a fixed value.
And, different reference powers can be set by adjusting the sampling coefficients K1 and K2.
All the circuits are realized by adopting analog circuits, multiple digital-to-analog conversion or analog-to-digital conversion is not needed, and components and parts are saved, so that the circuit is economical and practical.
Example IV
As shown in fig. 9A-9G, in one aspect, the present invention provides a constant power control circuit comprising: a load power detection module 41, a duty cycle adjustment module 42, the duty cycle adjustment module 42 including a divider 4103 and an integrator 4104;
the load voltage detection module 4101 samples the load voltage BAT in the whole process, and the load voltage BAT may be a battery voltage or other voltages, and the load current detection module 4102 samples the load current CS in the whole process.
Preferably, the load power detection module 41 inputs the detected load voltage signal to the divider 4103, the output terminal REF of the divider 4103 is electrically connected to one input terminal of the integrator 4104, the load power detection module 41 outputs the detected second detected voltage signal representing the load current signal to the other input terminal of the integrator 4104, the output terminal of the integrator 4104 is electrically connected to the duty cycle adjustment module 42, and the duty cycle adjustment module 42 generates the on time and the switching period of the load circuit switch, and controls the load circuit switch to be turned on and off, so as to output at constant power.
Preferably, the load power detection module 41 includes a load voltage detection module 4101, a load current detection module 4102.
The input end of the load voltage detection module 4101 is electrically connected with the load circuit for detecting the load voltage, the input end of the load current detection module 4102 is electrically connected with the load circuit for obtaining a second detection voltage signal representing the load current, the output end of the load voltage detection module 4101 is electrically connected with the input end of the divider 4103, the output end of the divider 4103 is electrically connected with the non-inverting input end of the integrator 4104, the output end of the load current detection module 4102 is electrically connected with the inverting input end of the integrator 4104, the output end of the integrator 4104 is electrically connected with the input end of the duty ratio adjustment module 42 for dividing and integrating the detected load voltage signal and the second detection voltage signal corresponding to the load current signal, and outputting the divided and integrated signal to the duty ratio adjustment module 42.
Preferably, referring to fig. 9C, the divider 4103 comprises a first operational amplifier 41031, a second capacitor C2, a third resistor R3, and an internal multiplier 41032;
the non-inverting input terminal of the first operational amplifier 41031 is electrically connected to the second internal reference voltage V2, the inverting input terminal thereof is electrically connected to the output terminal of the internal multiplier 41032 through the third resistor R3, the inverting input terminal thereof is also electrically connected to one terminal of the second capacitor C2, the other terminal of the second capacitor C2 is electrically connected to the output terminal of the first operational amplifier 41031, and the output terminal of the first operational amplifier 41031 is also electrically connected to one of the input terminals of the internal multiplier 41032 and serves as the output terminal of the divider; the other input of the internal multiplier 41032 is connected to the load voltage signal;
Preferably, referring to fig. 9D, the integrator 4104 includes a second operational amplifier 41041, a fourth capacitor C4, a sixth resistor R6, and a clamp 41042;
the non-inverting input terminal of the second operational amplifier 41041 is electrically connected to the output terminal of the divider, the inverting input terminal of the second operational amplifier 41041 is connected to the second detection voltage signal through the sixth resistor R6, the inverting input terminal of the second operational amplifier 41041 is also electrically connected to one end of the fourth capacitor C4, the other end of the fourth capacitor C4 is electrically connected to the output terminal of the second operational amplifier 41041, the output terminal of the second operational amplifier 41041 is electrically connected to the clamp circuit 41042, and the output terminal of the clamp circuit 41042 is used as the output terminal of the integrator. The clamp circuit 41042 is used to set the output voltage of the second operational amplifier 41041 to a certain range.
Preferably, the duty cycle adjustment module 42 further comprises a sawtooth generator 4201, a PWM comparator 4202, and an RS flip-flop 4203;
an input terminal of the PWM comparator 4202 is electrically connected to a first output terminal of the sawtooth generator 4201, another input terminal of the PWM comparator 4202 is also electrically connected to an output terminal of the integrator 4104, an output terminal of the PWM comparator 4202 is electrically connected to an R input terminal of the RS flip-flop 4203, an on-time of the load circuit switch is generated, and an on-time signal is output to the R input terminal of the RS flip-flop 4203;
The S input terminal of the RS flip-flop 4203 is electrically connected to the second output terminal of the sawtooth generator 4201, and is configured to generate a switching cycle of the load circuit switch, and output a switching cycle signal to the S input terminal of the RS flip-flop 4203;
the Q output end of the RS trigger 4203 is electrically connected to the input end of the driving module 4204, the output end of the driving module 4204 is electrically connected to the Power MOS transistor gate of the load voltage and current detection module, when the time point of the on time and the switching period is reached, the RS trigger 4203 outputs high and low levels to the driving module 4204 according to the on time signal and the switching period signal, and the driving module 4204 controls the switching of the load circuit switch;
an input of the sawtooth generator 4201 is electrically connected to an output of the drive module 4204 for resetting the state of the sawtooth generator 4201 at the end of a cycle.
Preferably, referring to fig. 9E, the sawtooth wave generator 4201 includes a first pulse generator 42011, a fourth comparator S4, a third capacitor C3, a third comparator S3, and a fifth resistor R5;
the input end of the first pulse generator 42011 is the input end of a sawtooth wave generator, the output end of the first pulse generator 42011 is electrically connected with the non-inverting input end of the fourth comparator S4, and the inverting input end of the fourth comparator S4 is grounded; the output end of the fourth comparator S4 is electrically connected with one end of the third capacitor C3, the other end of the third capacitor C3 is grounded, and the output end of the fourth comparator S4 is a first output end of the sawtooth wave generator;
The output end of the fourth comparator S4 is further electrically connected to the non-inverting input end of the third comparator S3, the inverting input end of the third comparator S3 is connected to a sixth internal reference voltage, the output end of the third comparator S3 is electrically connected to one end of the fifth resistor R5, the other end of the fifth resistor R5 is grounded, and the output end of the third comparator S3 is the second output end of the sawtooth wave generator 4201;
preferably, referring to fig. 9f, the pwm comparator 4202 includes a second comparator S2, a fourth resistor R4, and a second pulse generator 42021;
the second comparator S2 has its non-inverting input electrically connected to the first output of the sawtooth generator 4201 and its inverting input electrically connected to the output of the integrator 4104;
the output end of the second comparator S2 is grounded through a fourth resistor R4;
the output of the second comparator S2 is electrically connected to the second pulse generator 42021, and the output of the second pulse generator 42021 is the output of the PWM comparator 4202. When the second pulse generator 42021 receives a high signal, the inverter U5 delays the output of the pulse, so that the and gate outputs a high signal, and when a pulse time elapses, the inverter U5 outputs a low signal to the and gate U4, and the and gate U4 outputs a low signal. The principle of the first pulse generator 42011 is similar and will not be described again.
Preferably, the relationship between the on-time and the on-period is:
the opening time is less than the opening period;
the start time is counted from the start time of the start period;
in the opening time, the load circuit switch is in a conducting state;
at the switching cycle time other than the on time, the load circuit switch is in an off state.
Preferably, the S input of the RS flip-flop 4203 is electrically connected to the periodic pulse signal generator for generating a switching period of the load circuit switch, and outputting a switching period signal to the S input of the RS flip-flop 4203.
As can be seen from fig. 9C, the detected battery voltage BAT is obtained by a divider 4103, wherein the output value of the multiplier is divided by the amplification factor of the multiplier to obtain a constant value k, and then the voltage signal REF and the second detected voltage signal CS are converted into current signals and integrated by an integrator 4104 into a fourth capacitor C4 to obtain an EAO voltage signal.
The EAO voltage signal is compared with a sawtooth waveform Ramp generated by the sawtooth waveform generator 4201 to generate an on-time Ton in duty cycle for controlling the turn-on and turn-off of the power mos transistor of the load circuit 4205. The other output of sawtooth generator 4201 outputs a CLK signal representing the period of T.
Different constant power values can be set by modifying k and the current sampling ratio.
Specifically, fig. 9B shows a load battery and a current detection circuit suitable for the present embodiment. Where output BATT is the sampled load voltage signal, CS is the voltage signal that characterizes the sampled load current dependence, and GATE is the signal that drives the output that controls the switching of the load circuit 4205.
Specifically, fig. 9D shows an integrator 4104 circuit suitable for use in the present embodiment. The in-phase input is the calculated voltage signal REF, the inverse input is the second detection voltage signal CS, and the EAO voltage signal can be obtained by the integrator 4104. Further, the integrating circuit is also connected with a clamping circuit 41042, so that the output voltage is controlled in a range of 0.4V-1.9V. This ensures that the output voltage is more efficient when compared to the output of the subsequent sawtooth generator 4201.
When the power mos tube is turned on, the CS voltage, the voltage signal REF, and the resistor R6 are calculated to obtain a current, and the current charges the capacitor C4, so as to satisfy the following equation:
Ton*(CS-REF)/R6＝C4*(REF-EAO)；
when the power mos tube turns off, the voltage at CS is 0, the capacitor C4 discharges, and in order to reach steady state, the following equation is satisfied:
Toff*REF/R6＝C4*(REF-EAO)；
Bringing the period t=ton+toff, ref=k/BAT, cs=k1×iload into the above two equations yields:
p=iload×bat×ton/t=k/K1, P being the output constant power value.
Specifically, fig. 9E shows a specific circuit of a sawtooth wave generator 4201 suitable for use in the present embodiment. The sawtooth generator 4201 has two output terminals, one is a Ramp output terminal, i.e., a first output terminal, and the other is a CLK output terminal, i.e., a second output terminal. Wherein Ramp is used for input to PWM comparator 4202, which compares with the output of the previous integrator to generate Ton time and corresponding signal; the other CLK is applied to the S input of the RS flip-flop 4203 to generate a T time and corresponding signal, the period T of this embodiment being determined by the sawtooth generator 4201. Further, the sawtooth generator 4201 has an input, the input signal is GATE, which is a signal driving the output control load circuit switch 4205. After the GATE inputs the high level, the not GATE U2 delays a time sequence pulse, that is, both input ends of the and GATE are at the high level at the time of the time sequence pulse, so that the and GATE U3 outputs the high level, thereby controlling the switch S4 to be turned on to realize the instant discharging for the C3, after the time sequence pulse, the not GATE U2 outputs the low level, thereby the and GATE U3 outputs the low level, controlling the switch S4 to be turned off, thereby charging the capacitor C3, and the saw-tooth wave starts to climb from 0. Here, the not gate U2 and the and gate U3 together constitute a first pulse generator 42011.
Here, I1=c3=v6, so that a value of the period T can be obtained, where I1 is an output current of the current source. Thereafter, the control switch S3 is turned on, and CLK outputs a high level to the RS flip-flop 4203.
Specifically, fig. 9F shows a PWM comparator 4202 circuit suitable for use in the present embodiment. Inputs are a Ramp signal output by the sawtooth wave generator 4201 and an EAO signal representing a real-time power value output by the integrating circuit. The two signals are compared by the PWM comparator 4202, and when they are equal, a high level pulse is generated, which indicates that Ton time in the period has arrived, and the signal is input to an input terminal of the RS flip-flop 4203 and finally output to GATE, and controls the load circuit 4205 to switch off. Here, the NOT gate U5 and the AND gate U4 together form a second pulse generator
Specifically, fig. 9G shows an RS flip-flop 4203 circuit suitable for use in the present embodiment. Wherein the R input receives a signal representative of Ton and the S input receives a signal representative of T. And output to GATE for controlling the switching on and off of the switches of the load circuit 4205.
In another aspect, the present invention provides a constant power control method, including:
s1: detecting a load voltage, converting the load voltage into a load voltage signal, and obtaining a reference voltage signal inversely proportional to the load voltage signal through a divider 4103;
S2: detecting a load current, converting into a second detection voltage signal representing the load current signal;
s3, the reference voltage signal and the second detection voltage signal are subjected to an integrator 4104 to obtain an integrated signal;
s4: comparing the integrated power signal with the sawtooth wave to generate the on time of the load circuit switch;
s5: generating a switching period of a load circuit switch, wherein the switching period is a fixed value;
s6: and controlling the duty ratio of the switch of the load circuit according to the opening time and the switching period to realize the disconnection and the connection of the load circuit.
Further, step S1 further includes:
s11: the coefficients of the divider 4103 are modified to set different constant powers.
Further, step S2 further includes:
s21: the current sampling ratio of the detected load current is modified to set a different constant power.
In yet another aspect, the present invention provides a tobacco rod comprising the constant power control circuit described above.
In still another aspect, the present invention provides an electronic cigarette, including the tobacco rod described above.
It can be seen that the scheme of this embodiment can realize the following functions: after the circuit is connected and electrified, the load Power is sampled and valued to a certain extent according to the load voltage detection module 4101, the load current detection module 4102, the divider 4103 and the integrator 4104, and is input into the duty ratio adjustment module 42, then the on-time Ton and the period T of the Driver of the driving module 4204 are generated by the circuit of the duty ratio adjustment module 4233, and finally the on-off of the Power MOS switch of the load circuit 4205 is controlled by the Driver of the driving module 4204, so that the output Power P is ensured to be a fixed value.
Also, different reference powers may be set by adjusting the current sampling coefficient and the coefficient k of the divider 4103.
All the circuits are realized by adopting analog circuits, multiple digital-to-analog conversion or analog-to-digital conversion is not needed, and components and parts are saved, so that the circuit is economical and practical.
Example five
As shown in fig. 10, the present embodiment discloses an analog constant power control circuit.
In one aspect, the present invention provides a constant power control circuit comprising: the power detection module 51, the duty cycle adjustment module 5252.
Wherein the power detection module 51 includes: a current sampling module 5101, a voltage sampling module 5102, and a multiplier 5105.
The duty cycle adjustment module 5252 includes an error amplifier 5201, a PWM comparator 5202, and a drive module 5204.
The power detection module 51 multiplies the detected load voltage signal representing the voltage on the load circuit and the second detected voltage signal representing the current flowing through the load to generate a detected power signal capable of representing the load power, and outputs the detected power signal to the duty ratio adjustment module 52, and the duty ratio adjustment module 52 generates the on time and the on period of the switch of the load circuit 5205 according to the detected power signal, and controls the switch of the load circuit 5205 to be turned on and off, so as to ensure constant power output.
Preferably, the power detection module 51 includes a current sampling module 5101, a voltage sampling module 5102, a low pass filter, a multiplier 5105;
the input end of the voltage sampling module 5102 is electrically connected with the load circuit 5205 and is used for detecting load voltage to obtain a load voltage signal, the input end of the current sampling module 5101 is electrically connected with the load circuit 5205 and is used for detecting load current to obtain a second detection voltage signal, the output end of the voltage sampling module 5102 is electrically connected with the input end of the first low-pass filter 5103, the output end of the first low-pass filter 5103 is electrically connected with the input end of the multiplier 5105, the output end of the current sampling module 5101 is electrically connected with the input end of the second low-pass filter 5104, the output end of the second low-pass filter 5104 is electrically connected with the input end of the multiplier 5105, the output end of the multiplier 5105 is electrically connected with the input end of the duty ratio adjusting module 52 and is used for carrying out low-pass filtering on the detected load voltage signal and the second detection voltage signal to remove noise, then the detection power signal capable of representing load power is generated after multiplication operation, and the detection power signal is output to the duty ratio adjusting module 52.
Preferably, the duty cycle adjustment module 52 includes an error amplifier 5201, a PWM comparator 5202, a drive module 5204;
The negative input end of the error amplifier 5201 is electrically connected with the output end of the power detection module 51, and the positive input end of the error amplifier 5201 receives a preset reference power voltage signal and is used for comparing the detected power signal with the reference power signal to obtain a power difference value;
the input end of the PWM comparator 5202 is electrically connected with the output end of the error amplifier 5201, the other input end of the PWM comparator 5202 receives a sawtooth wave signal, the output end of the PWM comparator 5202 is electrically connected with the input end of the driving module 5204, the output end of the driving module 5204 is electrically connected with the control end of the load circuit switch, when the time point of the on time and the switching period is reached, the PWM comparator 5202 outputs high and low levels to the driving module 5204 according to the on time signal and the switching period signal, and the driving module 5204 controls the disconnection and the connection of the load circuit switch.
Preferably, the duty cycle adjustment module further comprises an RS flip-flop 5203;
the output end of the PWM comparator 5202 is electrically connected with the R input end of the RS trigger 5203, and the S input end of the RS trigger 5203 receives a clock pulse signal and is used for generating a switching period of a load circuit switch;
the Q output end of the RS trigger 5203 is electrically connected with the input end of the driver 5204 module, the output end of the driver 5204 module is electrically connected with the Power MOS tube grid electrode of the load circuit 5205, and when the time point of the opening time and the switching period is reached, the RS trigger 5203 outputs high and low levels to the driver 5204 module according to the opening time signal and the switching period signal, and the driver 5204 module controls the opening and the closing of the switch of the load circuit 5205;
Preferably, the relationship between the on-time and the switching period is:
the opening time is smaller than the switching period;
the on time is counted from the starting time of the switching period;
during the on time, the load circuit 5205 switches to an on state;
at a switching cycle time other than the on time, the load circuit 5205 switches to an off state.
Preferably, the positive electrode of the input terminal of the error amplifier 5201 is set to the reference power VREF to be achieved, and the negative electrode of the input terminal is the output terminal of the multiplier 5105.
The output of the error amplifier 5201 is: v (V) Output of ＝K*(V + -V - )
Error V Output of Compared with the sawtooth wave RAMP, the duty ratio duty is generated by the PWM comparator 5202, so that the on and off of the switching MOS transistor of the load circuit 5205 are finally controlled. Where CLK is a period T that is greater than Ton time controlled by the output of PWM comparator 5202.
Accordingly, the different constant power in the present embodiment can be set by setting the value of the positive input terminal VREF of the error amplifier 5201.
In another aspect, the present invention provides a constant power control method, including:
s1: detecting load voltage, converting the load voltage into an electric signal, and then obtaining a noise-removed load voltage signal through a low-pass filter;
S2: detecting load current, converting the load current into an electric signal, and then obtaining a second detection voltage signal with noise removed through a low-pass filter;
s3: the load voltage signal and the second detection voltage signal are passed through a multiplier 5105 to obtain a detection power signal which can represent load power;
s4: comparing the detected power signal with a preset reference power voltage to obtain a power difference value;
s5: comparing the power difference with the sawtooth wave to generate the opening time of the switch of the load circuit 5205;
s6: generating a switching period of switching of the load circuit 5205, wherein the switching period is a fixed value;
s7: the duty cycle of the switch of the load circuit 5205 is controlled according to the on-time and the switching period to control the turn-off and turn-on of the load circuit 5205.
Further, step S1 further includes:
s11: and obtaining a voltage sampling coefficient when the load voltage is detected so as to obtain different constant powers.
Further, step S2 further includes:
s21: the current sampling ratio of the detected load current is obtained to obtain different constant powers.
Further, step S4 further includes:
s41: and amplifying the power difference value by an error amplifier to obtain the coefficient of the error amplifier to adjust the power difference value precision.
Further, step S4 further includes:
s42: different constant powers are obtained by adjusting a preset reference power voltage.
The acquisition of parameters in the above method is described in detail in the previous circuit introduction, and is not described here again.
In yet another aspect, the present invention provides a tobacco rod comprising the constant power control circuit described above.
In still another aspect, the present invention provides an electronic cigarette, including the tobacco rod described above.
It can be seen that the scheme of this embodiment can realize the following functions: after the circuit is connected and electrified, load power is sampled to a certain value according to the current sampling module 5101, the voltage sampling module 5102, the low-pass filter and the multiplier 5105, the load power is input into the duty ratio adjusting module 5252, the on-time Ton and the period T for controlling the Driver are generated through the circuit of the duty ratio adjusting module 5252, and finally the Driver 5204Driver module controls the load circuit switch to ensure that the output power P is a fixed value.
And different constant power can be obtained through different voltage and current sampling coefficients.
All the circuits are realized by adopting analog circuits, multiple digital-to-analog conversion or analog-to-digital conversion is not needed, and components and parts are saved, so that the circuit is economical and practical.

Claims (10)

1. A constant power control circuit, the constant power control circuit comprising: the power detection module and the duty ratio adjustment module;
the power detection module multiplies the detected load voltage signal representing the voltage on the load circuit and the second detected voltage signal representing the current flowing through the load to obtain a detected power signal representing the load power, the detected power signal is output to the duty ratio adjustment module, and the duty ratio adjustment module generates the on time and the on period of the switch of the load circuit according to the detected power signal, controls the switch of the load circuit to be turned on and off, and outputs the detected power signal with constant power;
the duty ratio adjusting module comprises an error amplifier, a PWM comparator and a driving module;
one input end of the error amplifier is electrically connected with the output end of the power detection module, the other input end of the error amplifier receives a preset reference power signal and is used for comparing the detection power signal with the reference power signal to obtain an amplified power difference value, and the coefficient of the error amplifier is used for adjusting the precision of the power difference value;
One input end of the PWM comparator is electrically connected with the output end of the error amplifier, the other input end of the PWM comparator receives a sawtooth wave signal, the output end of the PWM comparator is electrically connected with the input end of the driving module, the output end of the driving module is electrically connected with the control end of the load circuit switch, the PWM comparator compares the power difference value with the sawtooth wave signal to generate a duty ratio, and when the time points of the starting time and the switching period are reached, the PWM comparator outputs high and low levels to the driving module according to the starting time signal and the switching period signal, and the driving module controls the switching-off and the switching-on of the load circuit switch;
wherein the constant power control circuit further comprises: a reference power setting module; the reference power setting module is connected with the other input end of the error amplifier and is used for outputting the reference power signal;
the reference power setting module is also used for being connected with an external power setting resistor, the external power setting resistor is used for setting the reference power signal, and the external power setting resistors with different resistance values correspond to different reference powers.
2. The constant power control circuit according to claim 1, wherein the reference power setting module includes a power setting operational amplifier, a third MOS transistor, a fourth MOS transistor, and a fourth switching resistor;
the non-inverting input end of the power setting operational amplifier is used for accessing a second reference voltage, the source electrode of the third MOS tube is electrically connected with a power supply, the grid electrode of the third MOS tube is electrically connected with the output end of the power setting operational amplifier, the drain electrode of the third MOS tube is respectively electrically connected with the inverting input end of the power setting operational amplifier and the external power setting resistor, the other end of the external power setting resistor is grounded, the source electrode of the fourth MOS tube is electrically connected with the power supply, the grid electrode of the fourth MOS tube is electrically connected with the grid electrode of the third MOS tube, the drain electrode of the fourth MOS tube is electrically connected with the fourth conversion resistor, and the other end of the fourth conversion resistor is grounded; the fourth conversion resistor is used for converting current flowing through the external power setting resistor into voltage and outputting a reference voltage signal corresponding to the reference power signal.
3. The constant power control circuit according to claim 1, wherein the power detection module comprises a current sampling module, a voltage sampling module, a multiplier;
The input end of the voltage sampling module is electrically connected with the load circuit and used for detecting load voltage to obtain a load voltage signal, the input end of the current sampling module is electrically connected with the load circuit and used for detecting load current to obtain a second detection voltage signal, the output end of the voltage sampling module is electrically connected with one input end of the multiplier, the output end of the current sampling module is electrically connected with the other input end of the multiplier, and the output end of the multiplier is electrically connected with the input end of the duty ratio regulating module and used for outputting a detection power signal representing load power to the duty ratio regulating module.
4. The constant power control circuit according to claim 1, wherein the duty cycle adjustment module further comprises an RS flip-flop;
the output end of the PWM comparator is electrically connected with the R input end of the RS trigger, and the S input end of the RS trigger receives a clock pulse signal and is used for generating the switching period of the load circuit switch;
the Q output end of the RS trigger is electrically connected with the input end of the driving module, the output end of the driving module is electrically connected with the control end of the load circuit switch, and when the starting time and the time point of the switching period are reached, the RS trigger outputs high and low levels to the driving module according to the starting time signal and the switching period signal, and the driving module controls the switching-off and the switching-on of the load circuit switch.
5. The constant power control circuit of claim 1, wherein the relationship between the on-time and the switching period is:
the on time is less than the switching period;
the on-time is counted from the starting time of the switching period;
in the opening time, the load circuit switch is in a conducting state;
at the switching cycle time other than the on time, the load circuit switch is in an off state.
6. A constant power control method applied to the constant power control circuit according to any one of claims 1 to 5, characterized by comprising:
s1: detecting load voltage, converting the load voltage into an electric signal, and then obtaining a noise-removed load voltage signal through a low-pass filter;
s2: detecting load current, converting the load current into an electric signal, and then obtaining a second detection voltage signal which is used for removing noise and represents the current flowing through the load through a low-pass filter;
s3: the load voltage signal and the second detection voltage signal are subjected to multiplier to obtain a detection power signal representing load power;
s4: comparing the detected power signal with a preset reference power signal to obtain a power difference value, amplifying the power difference value through an error amplifier to obtain a coefficient of the error amplifier to adjust the power difference value precision, wherein the reference power signal is output by a reference power setting module connected with the error amplifier, the reference power setting module is also connected with an external power setting resistor, the external power setting resistor is used for setting the reference power signal, and the external power setting resistors with different resistance values correspond to different reference powers;
S5: comparing the power difference with the sawtooth wave to generate the opening time of a load circuit switch;
s6: generating a switching period of a load circuit switch, wherein the switching period is a fixed value;
s7: and controlling the duty ratio of a switch of the load circuit according to the opening time and the switching period to control the opening and the closing of the load circuit.
7. The constant power control method according to claim 6, wherein the step S1 further includes:
s11: different voltage sampling coefficients when the load voltage is detected are obtained so as to obtain different constant powers;
step S2 further includes:
s21: and obtaining the current sampling proportion of the detected load current so as to obtain different constant powers.
8. The constant power control method according to claim 6, wherein step S4 further includes:
s42: different constant powers are obtained by adjusting a preset reference power voltage.
9. A tobacco rod comprising a constant power control circuit as claimed in any one of claims 1 to 5.
10. An electronic cigarette comprising the tobacco rod of claim 9.
CN202210006587.1A 2022-01-04 2022-01-04 Constant power control circuit and method, tobacco stem and electronic cigarette Active CN114468393B (en)

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