CN114465858A - Electronic device, signal processing method, and computer-readable storage medium - Google Patents

Abstract

An electronic device, a signal processing method, and a non-transitory computer-readable storage medium are provided. The electronic device includes processing circuitry configured to: performing phase adjustment on the received complex signal according to the compensation phase estimated by using the pilot signal; obtaining a received bit sequence based on the phase-adjusted complex signal through a low resolution analog-to-digital conversion process; and demodulating the received bit sequence based on a demodulation neural network obtained by training with the pilot signal to obtain modulation symbols of the received complex signal. According to at least one aspect of the embodiments of the present disclosure, it is possible to improve the conversion accuracy of the low resolution analog-to-digital conversion unit at the receiving end using the compensation phase obtained based on the pilot signal, and to implement compensation for the mixed distortion through the demodulation neural network obtained based on the pilot signal training, so that accurate demodulation decision can be achieved also for a high transmission data rate signal such as a terahertz signal.

Description

Electronic device, signal processing method, and computer-readable storage medium

Technical Field

The present application relates to the field of wireless communication technology, and more particularly, to an electronic device, a signal processing method, and a non-transitory computer-readable storage medium suitable for demodulating a high transmission data rate signal such as a terahertz signal.

Background

Terahertz communication has a high transmission data rate, and therefore a receiver is ideally expected to have a high-rate analog-to-digital conversion device. In practical application, due to the cost and technical limitations, only a single-bit receiver and other devices can be used to complete acquisition by using a low-precision analog-to-digital conversion device, which causes that a receiving end is difficult to accurately judge a received signal, thereby affecting the accuracy of demodulation judgment. In addition, the terahertz frequency band is located between the microwave frequency band and the optical frequency band, the communication device of the frequency band is difficult to manufacture, and hardware mismatch effects such as power amplifier nonlinear effect, in-phase (I branch) and quadrature (Q branch) imbalance and carrier phase noise exist. Such hardware mismatch effects will cause mixed distortion of the received signal, resulting in degradation of communication performance, thereby further affecting the accuracy of demodulation decision.

Therefore, in the related art, accurate demodulation of a high transmission data rate signal such as a terahertz signal cannot be achieved.

Disclosure of Invention

The following presents a simplified summary of the disclosure in order to provide a basic understanding of some aspects of the disclosure. However, it should be understood that this summary is not an exhaustive overview of the disclosure. It is not intended to identify key or critical elements of the disclosure or to delineate the scope of the disclosure. Its sole purpose is to present some concepts of the disclosure in a simplified form as a prelude to the more detailed description that is presented later.

In view of the above-mentioned problems, the present disclosure proposes an electronic device, a signal processing method, and a non-transitory computer-readable storage medium suitable for processing a high transmission data rate signal such as a terahertz signal, which are capable of improving the conversion accuracy of a low resolution analog-to-digital conversion unit (e.g., a single bit analog-to-digital conversion unit) and compensating for mixed distortion using a phase estimation and demodulation neural network obtained based on a pilot signal at a receiving end, thereby improving the accuracy of demodulation decision.

According to an aspect of the disclosure, there is provided an electronic device comprising processing circuitry configured to: performing phase adjustment on the received complex signal according to the compensation phase estimated by using the pilot signal; obtaining a received bit sequence based on the phase-adjusted complex signal through a low resolution analog-to-digital conversion process; and demodulating the received bit sequence based on a demodulation neural network obtained by training with the pilot signal to obtain modulation symbols of the received complex signal.

According to still another aspect of the present disclosure, there is provided a signal processing method including: performing phase adjustment on the received complex signal according to the compensation phase estimated by using the pilot signal; obtaining a received bit sequence based on the phase-adjusted complex signal through a low resolution analog-to-digital conversion process; and demodulating the received bit sequence based on a demodulation neural network obtained by training with the pilot signal to obtain modulation symbols of the received complex signal.

According to yet another aspect of the present disclosure, there is also provided a non-transitory computer-readable storage medium storing executable instructions that, when executed by a processor, cause the processor to perform the respective functions of the above-described signal processing method or electronic device.

According to other aspects of the present disclosure, there are also provided computer program codes and computer program products for implementing the above-described signal processing method according to the present disclosure.

According to at least one aspect of the embodiments of the present disclosure, it is possible to improve the conversion accuracy of the low resolution analog-to-digital conversion unit at the receiving end using the compensation phase obtained based on the pilot signal, and to implement compensation for the mixed distortion through the demodulation neural network obtained based on the pilot signal training, so that accurate demodulation decision can be achieved also for a high transmission data rate signal such as a terahertz signal.

Additional aspects of the disclosed embodiments are set forth in the description section that follows, wherein the detailed description is presented to fully disclose the preferred embodiments of the disclosed embodiments without imposing limitations thereon.

Drawings

The drawings described herein are for illustrative purposes only of selected embodiments and not all possible implementations, and are not intended to limit the scope of the present disclosure. In the drawings:

fig. 1 is a diagram illustrating constellation distortion of a received signal caused by mixed distortion in the prior art;

fig. 2 is a schematic block diagram illustrating one configuration example of an electronic device according to an embodiment of the present disclosure;

FIG. 3 is a schematic block diagram illustrating an example circuit implementation of an electronic device in accordance with an embodiment of the present disclosure;

FIG. 4 schematically illustrates an example of a demodulation neural network that an electronic device can employ in accordance with an embodiment of the present disclosure;

FIG. 5 is a schematic diagram illustrating a data structure of a complex signal received by an electronic device according to an embodiment of the disclosure;

FIG. 6 is a schematic diagram illustrating an example of an information interaction process according to an embodiment of the present disclosure;

FIG. 7 is an illustration of an example simulation result used to illustrate performance degradation caused by phase noise;

fig. 8 is a diagram of example simulation results for illustrating the performance of demodulation processing by an electronic device according to an embodiment of the present disclosure;

fig. 9 is a flowchart illustrating a process example of a signal processing method according to an embodiment of the present disclosure;

fig. 10 is a block diagram illustrating a first example of a schematic configuration of an eNB to which the techniques of this disclosure may be applied;

fig. 11 is a block diagram illustrating a second example of a schematic configuration of an eNB to which the techniques of this disclosure may be applied;

fig. 12 is a block diagram showing an example of a schematic configuration of a smartphone to which the technique of the present disclosure may be applied;

fig. 13 is a block diagram showing an example of a schematic configuration of a car navigation device to which the technique of the present disclosure can be applied.

While the disclosure is susceptible to various modifications and alternative forms, specific embodiments thereof have been shown by way of example in the drawings and are herein described in detail. It should be understood, however, that the description herein of specific embodiments is not intended to limit the disclosure to the particular forms disclosed, but on the contrary, the intention is to cover all modifications, equivalents, and alternatives falling within the spirit and scope of the disclosure. It is noted that throughout the several views, corresponding reference numerals indicate corresponding parts.

Detailed Description

Examples of the present disclosure will now be described more fully with reference to the accompanying drawings. The following description is merely exemplary in nature and is not intended to limit the present disclosure, application, or uses.

Example embodiments are provided so that this disclosure will be thorough, and will fully convey the scope to those skilled in the art. Numerous specific details are set forth such as examples of specific components, devices, and methods to provide a thorough understanding of embodiments of the present disclosure. It will be apparent to those skilled in the art that specific details need not be employed, that example embodiments may be embodied in many different forms and that neither should be construed to limit the scope of the disclosure. In certain example embodiments, well-known processes, well-known structures, and well-known technologies are not described in detail.

The description will be made in the following order:

1. description of the problem

2. Configuration example of electronic device

2.1 configuration example

2.2 example Circuit implementation

2.3 determination of the Compensation phase

2.4 training of the demodulation neural network

3. Examples of information interaction Processes

4. Simulation example

5. Method embodiment

6. Application example

<1. description of the problems >

Terahertz communication has a high transmission data rate, and therefore a receiver is ideally expected to have a high-rate analog-to-digital conversion device. In practical application, due to cost and technical limitations, acquisition can only be completed by using a low-precision analog-to-digital conversion device by using equipment such as a single-bit receiver and the like. Currently, single-bit receivers for low-frequency band signals generally use likelihood information decision to complete the demodulation of the signal. However, for terahertz communication, due to the existence of terahertz hardware mismatch, accurate expression of likelihood information of a received signal is not easy to obtain, so that accurate judgment of a terahertz single-bit receiver cannot be completed by using a traditional method. This makes it difficult for the receiving end to make an accurate decision on the received signal, thereby affecting the accuracy of demodulation decision.

In addition, the terahertz frequency band is located between the microwave frequency band and the optical frequency band, the communication device of the frequency band is difficult to manufacture, and hardware mismatch effects such as power amplifier nonlinear effect, in-phase (I branch) and quadrature (Q branch) imbalance and carrier phase noise exist. Such hardware mismatch effects will cause mixed distortion of the received signal, resulting in degradation of communication performance, thereby further affecting the accuracy of demodulation decision.

More specifically, for terahertz communication employing Quadrature Phase Shift Keying (QPSK) modulation, it is ideal that a bi-bit group (also referred to herein as "modulation symbol") is formed at a transmitter for every two bits of a baseband signal (binary bit sequence), and I-path carrier signals cos (2 π f) are respectively transmitted as the bi-bit group_ct) and Q-path carrier signal sin (2 π f)_ct) (wherein, f_cCarrier frequency, t represents time) and then added to obtain a transmission symbol or transmission symbol (transmission symbol) of QPSK modulation. As an example, for four different bi-bit groups or modulation symbols 11,10,00,01, transmission symbols of 1+ j,1-j, -1-j, -1+ j, respectively, may be obtained after QPSK modulation. In the QPSK constellation, each transmission symbol corresponds to one constellation point, and an angle between a vector composed of each constellation point and an origin and a coordinate axis is 45 degrees, each constellation point and the coordinate axis having an equal distance therebetween.

However, in practical applications, IQ imbalance may exist in the transmitter. For example, can be given by ∈_T，φ_TThe imbalance factors of the IQ path amplitude and phase are respectively expressed, and the I path carrier signal becomes (1+ epsilon)_T)cos(2πf_ct-φ_T) The Q carrier signal becomes (1-epsilon)_T)sin(2πf_ct+φ_T). At the same time, further consider at the transmitterPhase noise theta_TThen the ideal transmitted signal (complex signal) s after QPSK modulation will become the following form:

wherein, mu_T＝cosφ_T-j∈_Tsinφ_T,v_T＝∈_Tcosφ_T-jsinφ_T，v_Ts^*Is the mirror interference term caused by IQ imbalance.

Here, as an example, the phase noise θ_TMay have the form of a block walk model, i.e. theta_TWithin the kth transport block is a fixed value theta_kAnd differ between adjacent transport blocks by a gaussian distributed random walk term Δ θ_k. Phase noise theta of k +1 th transmission block_k+1Can be expressed in the following form:

wherein the content of the first and second substances,

representing a mean of 0 and a variance of

The normal distribution of (c),

also known as the walking term variance.

QPSK modulated signal at transmitter

Will be amplified by a Power amplifier (Power amplifier). The non-linear characteristics of the power amplifier may result in a signal

Further distortion of (2). The non-linear characteristic of the power amplifier can be represented, for example, using a memoryless polynomial model, and the signal output by the power amplifier can be represented in the form:

wherein 2K-1 represents the polynomial order.

The signal s output by the power amplifier of the transmitter_PAWill be transmitted over the channel to the receiver side. Because the terahertz communication generally adopts antennas with extremely high directional gain at the transmitting and receiving ends, only one effective transmission path in a channel can be considered, and the effective transmission path can be represented by a flat fading channel model. For example, the signal received at the receiver side can be represented in the following form:

y＝hs_PA+ w … equation (4)

Where h is the channel fading factor,

is Additive White Gaussian Noise (AWGN),

represents a mean of 0 and a variance of

Circular Symmetric Complex Gaussian (CSCG) distribution.

For the received signal y, the ideal processing at the receiver is to combine it with the I-path carrier signal cos (2 pi f), respectively_ct) and Q-path carrier signal sin (2 π f)_ct) to obtain the real and imaginary parts of the complex signal (which may also be referred to as the I and Q signals). However, similarly to the transmitter, IQ imbalance and phase noise may also occur at the receiver. With e as_R，φ_RRespectively representing the amplitude imbalance factor and the phase imbalance factor of the IQ path, and the I path carrier signal at the receiverBecomes (1 +. epsilon)_R)cos(2πf_ct-φ_R) The Q carrier signal becomes (1-epsilon)_R)sin(2πf_ct+φ_R). While further taking into account the phase noise theta at the receiver_RThe final received signal can then be represented in the form:

wherein, mu_R＝cosφ_R+j∈_Rsinφ_R,v_R＝∈_Rcosφ_R-jsinφ_R。

The received signal in the form of equation (5) above may be expressed in the form of a complex signal as follows:

wherein the content of the first and second substances,

and

are respectively complex signals

Real and imaginary parts of (c).

For complex signals such as in the form of equation (6) involving various distortions as discussed above, obtained at the receiving end with prior art linear equalization strategies

And

the schematic diagram of the formed constellation points is shown in fig. 1, wherein the included angle θ between the vector formed by the constellation points and the origin and the coordinate axis is no longer 45 degrees in the ideal case, and the distance d between the constellation points and the origin is₁、d₂、d₃、d₄No longer equal, i.e. a distortion of the constellation occurs.

As can be seen from the above discussion about distortion, there are many reasons for causing distortion of the constellation diagram shown in fig. 1, and in order to solve this problem, a compensation (or predistortion) algorithm at the transmitting end and a compensation algorithm at the receiving end are generally needed to be jointly solved, where the predistortion algorithm at the transmitting end may handle power amplifier nonlinearity and IQ spur imbalance at the transmitting end, and the compensation algorithm at the receiving end may handle IQ spur imbalance and phase noise at the receiving end. However, the hardware of the terahertz frequency band is difficult to manufacture and high in cost, and in order to perform hardware mismatch effect research, a compensation link of a transmitting end often needs a high-precision high-rate sampling link, so that the design difficulty is high and the realization is difficult; the hybrid distortion of the terahertz channel cannot be effectively compensated by using the compensation technology of the receiving end only.

To this end, the present disclosure provides an electronic device, a signal processing method, and a non-transitory computer-readable storage medium suitable for processing a high transmission data rate signal such as a terahertz signal, which are capable of improving the conversion accuracy of a low resolution analog-to-digital conversion unit (e.g., a single bit analog-to-digital conversion unit) and compensating for mixed distortion using a phase estimation and demodulation neural network obtained based on a pilot signal at a receiving end, thereby improving the accuracy of demodulation decision.

The electronic device according to the present disclosure may be an electronic device on a user equipment side, or may be an electronic device on a network side, as long as it can function as a receiving end device and perform corresponding signal processing.

The electronic apparatus on the user apparatus side may be implemented as various user apparatuses, for example, a mobile terminal such as a smartphone, a tablet Personal Computer (PC), a notebook PC, a portable game terminal, a portable/cryptographic dog-type mobile router, and a digital camera, or a vehicle-mounted terminal such as a car navigation apparatus. The user equipment described above may also be implemented as a terminal (also referred to as a Machine Type Communication (MTC) terminal) that performs machine-to-machine (M2M) communication. Further, the user equipment may include a wireless communication module (such as an integrated circuit module including a single chip) or the like mounted on each of the above-described terminals.

The electronic device on the network side may be a base station device itself, for example, an eNB (evolved node B), a gNB, and any type of TRP (transmission and reception port). The TRP may have a transmitting and receiving function, and may receive information from or transmit information to, for example, a user equipment and a base station apparatus. In one example, the TRP may provide a service to the user equipment and be controlled by the base station equipment. That is, the base station apparatus can provide a service to the user equipment through the TRP. In some specific embodiments or examples below, a description is made with a base station apparatus as an example of an electronic apparatus on a network side, but the present disclosure is not limited thereto, and may be suitably applied to a case of an electronic apparatus having a similar function.

<2. configuration example of electronic apparatus >

[2.1 configuration example ]

Fig. 2 is a block diagram showing one configuration example of an electronic apparatus according to an embodiment of the present disclosure.

As shown in fig. 2, the electronic device 200 may include a phase adjustment unit 210, a low resolution analog-to-digital conversion unit 220, a demodulation unit 230, and an optional compensated phase estimation unit 240.

Here, each unit of the electronic device 200 may be included in the processing circuit. The electronic device 200 may include one processing circuit or may include a plurality of processing circuits. Further, the processing circuitry may include various discrete functional units to perform various different functions and/or operations. It should be noted that these functional units may be physical entities or logical entities, and that units called differently may be implemented by the same physical entity.

According to an embodiment of the present disclosure, the phase adjustment unit 210 of the electronic device 200 may perform phase adjustment on the received complex signal according to the compensation phase estimated using the pilot signal. The low resolution analog-to-digital conversion unit 220 may obtain a received bit sequence based on the phase-adjusted complex signal through a low resolution analog-to-digital conversion process. Demodulation unit 230 may demodulate the received bit sequence based on a demodulation neural network obtained by training with the pilot signal to obtain modulation symbols of the received complex signal.

As an example, the complex signal and the pilot signal may each include a QPSK modulated signal, and are, for example, high transmission data rate signals such as terahertz signals. Here, although the terminology of QPSK modulation is used, it is understood by those skilled in the art that in the context of the present disclosure, reference to QPSK modulation may encompass relative phase shift QPSK (OQPSK), differential DQPSK (DQPSK) modulation, and the like.

As described heretofore in the section of "description of problem", for a high transmission data rate signal such as a terahertz signal employing QPSK modulation, a complex signal, for example, in the form of equation (6) is received by the electronic apparatus 200 due to the influence caused by various distortions

Problems such as constellation distortion as shown in fig. 1 may arise. Accordingly, the electronic apparatus 200 may estimate the compensation phase θ according to the using the pilot signal through the phase adjustment unit 210^*Phase adjusting the received complex signal to obtain a phase-adjusted complex signal of the form:

with properly determined compensating phase theta^*The phase adjustment process described above may be adapted to rotate the constellation of the received signal appropriately, for example to maximize the minimum distance between the constellation point and different coordinate axes (including the x-axis and the y-axis), for example such that d in fig. 1₁、d₂、d₃And d₄The minimum distance in (1) is maximized. Such phase adjustments will increase the probability of a correct decision of the received signal. Details of how the compensation phase is determined will be described later in the section "determination of compensation phase".

As an example, a phaseThe bit adjustment unit 210 may be configured to control a local oscillator (not shown) to perform phase rotation on the received complex signal according to the compensated phase, so as to implement the above-mentioned phase adjustment. A local oscillator, for example comprised in the electronic device 200 or connected to the electronic device 200, may be used to generate the I-path carrier signal and the Q-path carrier signal at the electronic device 200 at the receiving end for obtaining the real part and the imaginary part of the complex signal. When the phase adjustment unit 210 performs phase adjustment by controlling the local oscillator, the phase of the carrier signal is actually directly adjusted, and the received complex signal is multiplied by the carrier signal, so that the complex signal after phase adjustment can be directly obtained

Real part of

And imaginary part

The phase-adjusted complex signal can be changed from equation (6) to the following form:

the complex signal after phase adjustment for the phase adjustment unit 210

The low resolution analog-to-digital conversion unit 220 may perform a low resolution analog-to-digital conversion process to obtain a received bit sequence. Here, the low resolution analog-to-digital conversion processing unit 220 may be configured to obtain a real part received bit sequence and an imaginary part received bit sequence based on a real part and an imaginary part of the phase-adjusted complex signal, respectively, through the low resolution analog-to-digital conversion process. For example, the low resolution analog-to-digital conversion processing unit 220 may obtain the real part of each transmission symbol of the received complex signal by oversampling the real part and the imaginary partA partial received bit sequence and an imaginary received bit sequence. In other words, the phase adjustment unit 210 can perform single-bit analog signal acquisition for the I/Q branch, each time the real part of the acquired signal

And imaginary part

Can be expressed in the following forms

Since the correct probability of a received signal decision in the presence of noise depends on the real part of the respective transmitted symbol acquired

And imaginary part

The comparison with 0, and thus on the minimum distance (e.g. d in fig. 1) of the respective constellation point to the different coordinate axes (including the x-axis and the y-axis)₁、d₂、d₃And d₄Minimum distance) and thus increasing the minimum distance after phase adjustment will increase the probability of a correct decision of the received signal (i.e., increase the real part of each transmitted symbol)

And imaginary part

The probability of correctness of the comparison result with 0).

In the above manner, for each transmitted symbol of the received complex signal, e.g. byThe low resolution adc unit 220 performs single bit oversampling to obtain a real bit sequence

And imaginary bit sequence

Where N is an oversampling multiple of the signal, which may be 10, for example. Such real and imaginary received bit sequences may also be collectively represented as received bit sequences

The received bit sequence of the current transmission symbol of the received complex signal obtained by the low resolution analog-to-digital conversion unit 220, for example

Demodulation unit 230 may demodulate the pilot signal based on a demodulation neural network obtained by training it with the pilot signal to obtain a modulation symbol of the transmission symbol. The demodulation neural network is obtained by training, for example, a pilot signal labeled with modulation symbols, and can characterize a mapping relationship between a received bit sequence of transmission symbols of the received complex signal and corresponding modulation symbols, further details of which will be described later in the section "training of demodulation neural network".

The above describes a configuration example of an electronic device according to an embodiment of the present disclosure. With the electronic device according to the embodiment of the present disclosure, the conversion accuracy of the low resolution analog-to-digital conversion unit can be improved at the receiving end using the compensation phase obtained based on the pilot signal, and the compensation of the mixed distortion is realized through the demodulation neural network obtained based on the pilot signal training, so that accurate demodulation decision can be realized also for a high transmission data rate signal such as a terahertz signal.

[2.2 example Circuit implementation ]

Next, an example circuit implementation of the electronic device 200 shown in fig. 2 will be described with reference to fig. 3. Fig. 3 is a schematic block diagram illustrating an example circuit implementation 300 of an electronic device according to an embodiment of the present disclosure, in which a phase adjustment unit 310, ADCs 320a and 320b (also collectively referred to as ADCs 320 when distinction is not required) as single bit flip-flops, a demodulation unit 330, and a compensated phase estimation unit 340 are shown as examples of the phase adjustment unit 210, the low resolution analog-to-digital conversion unit 220, the demodulation unit 230, and the compensated phase estimation unit 240, respectively, in fig. 2. Further, fig. 3 also shows an antenna 350, a local oscillator LO, a phase shifter 360, and multiplication circuits 370a and 370b (also collectively referred to as the multiplication circuit 370 when distinction is not necessary) which are basic configurations of an electronic device on the receiving side. These components (antenna 350, local oscillator LO, phase shifter 360, multiplication circuit 370) may be optional additional portions of example circuit implementation 300, or may be additional circuit portions connected to example circuit implementation 300, as the present disclosure is not limited thereto.

The respective units 310 to 340 of fig. 3 may be used to implement the functionality of the respective units 210 to 240 of the electronic device 200 described hereinbefore with reference to fig. 2. For example, for complex signals received via antenna 350

The phase adjustment unit 310 estimates the compensated phase θ from the pilot signal, for example, by the compensated phase estimation unit 340^*Controlling the local oscillator LO to rotate its phase corresponds to a carrier signal cos (2 π f)_ct) and sin (2 π f)_ct) and the multiplication circuits 370a and 370b may utilize the corresponding phase adjusted carrier signal cos (2 pi f)_ct+θ^*) And sin (2 π f)_ct+θ^*) And complex signal

Multiplying to directly obtain a phase-adjusted complex signal

OfPart (A)

And imaginary part

In this manner, in an example circuit implementation of an electronic device such as fig. 3 of an embodiment of the present disclosure, phase compensation of a received signal may be achieved through analog devices.

Here, the ADCs 320a and 320b, which are examples of the low resolution analog-to-digital conversion processing unit, may respectively correspond to the real part

And imaginary part

Performing single bit oversampling to obtain a received complex signal

Of each transmission symbol

And imaginary bit sequence

Where N is an oversampling multiple of the signal, which may be 10, for example. The low resolution analog-to-digital conversion processing unit performing single bit oversampling can be implemented by using an analog signal collector such as a high speed flip-flop or the like. For example, a DI HMC729LC3C high speed flip-flop may be employed as the ADC 320 in this example.

Received complex signals obtained by the ADCs 320a and 320b

Of each transmission symbol

And imaginary bit sequence

Is input to a demodulation unit 330 to obtain a received complex signal

For each transmission symbol n.

As an example, the demodulation unit 330 may perform demodulation processing by using a Deep feed forward neural network (DFNN), which fits a mapping relationship between the received bit sequence and the modulation symbol by using a fitting capability of the Deep feed forward neural network to fit the mapping relationship with arbitrary precision, thereby completing demodulation decision.

Fig. 4 schematically illustrates an example of a demodulation neural network 400 that may be employed. The network includes an input layer 410, an hidden layer 420, and an output layer 430. The input layer 410 has, for example, 2N input channels (N is an oversampling multiple and is, for example, 10) to input, for each transmission symbol of the received complex signal, a real part bit sequence of the transmission symbol

And imaginary bit sequence

Formed received bit sequence

The number of layers of the hidden layer 420 may be 3, for example, the number of neurons in each hidden layer may be (10,10,10), and the activation function of the hidden layer is a Tan-Sigmoid (Tansig) function, i.e., f (x) is 2/(1+ e)^-2x) -1. The activation function of the output layer 430 is a Linear rectification Unit (ReLU) function, i.e., f (x) max (x, 0).

The demodulation neural network may characterize the mapping between the received bit sequence and the corresponding modulation symbol for each transmission symbol of the received complex signal, for example, by the following equation:

wherein, DFNN (is) represents the input-output relation of the demodulation neural network, and n represents the received bit sequence of the current transmission symbol obtained by the demodulation neural network

The modulation symbol corresponding to the transmission symbol may be, for example, one of the dibit groups 11,10,00, 01.

As mentioned above, before the electronic device of the embodiment of the present disclosure receives the data signal and performs corresponding processing, it is necessary to complete the determination of the compensation phase and the training of the demodulation neural network by using the pilot signal. Accordingly, in the example circuit implementation of fig. 3, if the antenna 350 receives not a data signal but a pilot signal, the pilot signal will be processed similarly to the data signal by the phase adjustment unit 310, the ADCs 320a and 320b as examples of the low resolution analog-to-digital conversion unit, etc. to obtain a received bit sequence of the pilot signal, and a compensation phase is determined using the received bit sequence, for example, by the processing of the compensation phase estimation unit 340, etc., or the processing of the demodulation unit 440, etc. trains the demodulation neural network.

Next, the process of determining the compensation phase and training of the demodulation neural network that may be implemented by the electronic device of embodiments of the present disclosure will be described with continued reference to the example circuit implementation of fig. 3.

[2.3 determination of compensated phase ]

As an example, the electronic device of the embodiment of the present disclosure may estimate the compensation phase from the transmission symbol of the first pilot signal and the first received bit sequence obtained based on the received first pilot signal, for example, by processing of the compensation phase estimation unit 340 of fig. 3 or the like.

Here, it is assumed that the transmitting end transmits a first pilot signal having a complex signal form described below

p₁＝p_1I+jp_1Q… equation (11)

p_1IAnd p_1QThe real and imaginary parts of the first pilot signal, respectively, may be 1 or-1, respectively, depending on the currently transmitted symbol of the pilot signal.

The transmitting end may obtain the above first pilot signal, for example, by performing QPSK modulation on a bit sequence including a plurality of dibit groups ab, where each dibit group ab obtains one transmission symbol in the first pilot signal after modulation. Each transmission symbol in the first pilot signal may correspond to at least two adjacent constellation points in the QPSK constellation, and preferably may correspond to only two adjacent constellation points in the QPSK constellation. Distance between any two adjacent constellation points and coordinate axis (such as d shown in fig. 1) due to symmetry of the constellation points in QPSK constellation₁、d₂、d₃And d₄) I.e. the distances of all four constellation points from the coordinate axis can be characterized.

Accordingly, when the transmitting end transmits the first pilot signal, QPSK modulation may be performed using a bit sequence capable of obtaining transmission symbols corresponding to the selected two adjacent constellation points, thereby obtaining the first pilot signal including the respective two of 1+ j,1-j, -1-j, -1+ j. As an example, the individual transmission symbols in the first pilot signal may alternately correspond to two adjacent constellation points in a QPSK constellation. For example, if two dibit groups 11 and 10 are selected, the transmission symbol of the first pilot signal includes 1+ j and 1-j (corresponding to two adjacent constellation points on the right side in the QPSK constellation), the first pilot signal may be represented as [1+ j,1-j, …,1+ j,1-j ]. The approach of this example may simplify the generation of the first pilot signal compared to an approach that employs all four transmission symbols to constitute the first pilot signal.

The electronic device at the receiving end may obtain the first pilot signal p through the antenna 350 of fig. 3₁Received signal of

(hereinafter also referred to as received first pilot signal or simply first pilot signal as appropriate)

). For the received first pilot signal

The electronic device may perform phase adjustment, for example, by phase adjustment unit 310 of fig. 3, and may obtain a first received bit sequence based on the phase-adjusted, received first pilot signal by low resolution analog-to-digital conversion processing, for example, by ADC 320 of fig. 3.

For example, for a first pilot signal received via antenna 350

The phase adjustment unit 310 performs a phase rotation of the carrier signal, i.e. a phase rotation of the carrier signal, by the local oscillator LO with a compensation phase θ (t) to be determined which varies continuously over time, e.g. between 0 and π, and the multiplication circuits 370a and 370b perform a corresponding phase adjustment of the carrier signal cos (2 π f_ct + θ (t)) and sin (2 π f)_ct + θ (t)) and a first pilot signal

Multiplying to obtain a phase adjusted received first pilot signal

Real part of

And imaginary part

Here, the ADCs 320a and 320b may respectively align the phase-adjusted first pilot signals

Real part of

And imaginary part

Performing single-bit oversampling to obtain a real part bit sequence of each transmission symbol of the received first pilot signal

And imaginary bit sequence

Where N is an oversampling multiple of the signal, which may be 10, for example. The above real part bit sequence

And imaginary bit sequence

May be collectively referred to as a first received bit sequence

The electronic device may also compensate the processing of the phase estimation unit 340 based on the first received bit sequence (c:, by the processing of the electronic device

And

) Relative to the first pilot signal p₁Estimating a compensation phase from the error of the transmitted symbol; and the received first pilot signal may be phase adjusted based on the estimated compensation phase. For example, the processing circuit is further configured to determine a compensation phase that minimizes the error as a final compensation phase.

As an example, the compensated phase estimation unit 340 may estimate a real part bit sequence of each transmission symbol from the received first pilot signal

And imaginary bit sequence

And a first pilot signal p₁I.e. the first pilot signal p₁Real part p of_1IAnd imaginary part p_1Q) And comparing the two phases to determine the optimal compensation phase. More specifically, it is possible here to continuously vary (e.g., continuously increase or decrease) the compensation phase θ (t) in accordance with the phase adjustment obtained after the phase adjustment using the supplementary phase

And

with respect to p, each bit (also referred to as a sampling point) in_1IAnd p_1QAnd judging the optimal phase by the number of the judgment errors. After each phase change, the compensated phase estimation unit 340 re-decides the first received bit sequence of the current transmitted symbol of the first pilot signal

If the number of the erroneous sampling points is reduced, the compensation phase is favorable for improving the demodulation accuracy, and if the number of the erroneous sampling points is increased, the compensation phase is unfavorable for improving the demodulation accuracy. In this way, for example, the first received bit sequence may be made

The compensation phase with the smallest number of erroneous sampling points is determined as the final compensation phase. Therefore, although the distance d between each constellation point and the coordinate axis at the current compensation phase in the constellation diagram shown in fig. 1 cannot be directly calculated₁、d₂、d₃And d₄Can still determine that d can be achieved₁、d₂、d₃And d₄The minimum distance in (a) maximizes the optimal compensation phase for this effect.

[2.4 training of the demodulation neural network ]

As an example, the electronic device of the embodiment of the present disclosure may obtain the demodulation neural network through training using a second received bit sequence obtained based on the received second pilot signal, which is marked with a modulation symbol of the second pilot signal, through, for example, processing of the demodulation unit 330 of fig. 3.

Here, it is assumed that the transmitting end transmits a second pilot signal having a complex signal form as described below

p₂＝p_2I+jp_2Q… equation (12)

p_2IAnd p_2QThe real and imaginary parts, respectively, of the second pilot signal, which may be 1 or-1, respectively, depending on the currently transmitted symbol of the pilot signal. The transmitting end obtains the above second pilot signal, for example, by performing QPSK modulation on a bit sequence including a plurality of dibit groups ab, where each dibit group ab obtains one transmission symbol in the second pilot signal after modulation.

Preferably, the second pilot signal p₂All transmission symbols corresponding to different dibit groups (i.e., different modulation symbols) are included so that the pilot signal is used to train the demodulation neural network to obtain a mapping between the bit sequence of each transmission symbol and each modulation symbol of the received complex signal. In other words, the second pilot signal p₂The transmission symbols in (a) may correspond to four constellation points in a QPSK constellation.

Accordingly, when the transmitting end transmits the second pilot signal, QPSK modulation may be performed using a bit sequence capable of obtaining all four constellation points (e.g., using a bit sequence including four different dibit groups or modulation symbols 11,10,00, 01), so as to obtain the second pilot signal including 1+ j,1-j, -1-j, -1+ j. As an example, the second pilot signal p₂May randomly correspond to four constellation points in a QPSK constellation. For example, the second pilot signal may be represented as [1+ j, -1+ j,1-j, -1-j, -1-j,1+ j, …,1-j,1+ j]。

The electronic device at the receiving end can obtain the above-mentioned second effect through the antenna 350 of fig. 3Two pilot signals p₂Received signal of

(hereinafter also referred to as received second pilot signal or simply second pilot signal as appropriate)

). For the received second pilot signal

The electronic device may perform phase adjustment, for example, by phase adjustment unit 310 of fig. 3, and may obtain a second received bit sequence based on the phase-adjusted, received second pilot signal by low-resolution analog-to-digital conversion processing, for example, by ADC 320 of fig. 3.

For example, for a second pilot signal received via antenna 350

The phase adjustment unit 310 uses the compensated phase θ determined by the compensated phase estimation unit^*The multiplication circuits 370a and 370b use the respective phase-adjusted carrier signal cos (2 pi f) by phase-rotating it, i.e. by phase-rotating the carrier signal, by the local oscillator LO_ct+θ^*) And sin (2 π f)_ct+θ^*) And a second pilot signal

Multiplying to obtain a phase adjusted received second pilot signal

Real part of

And imaginary part

Here, ADC320a and 320b may respectively align the phase-adjusted second pilot signals

Real part of

And imaginary part

Performing single-bit oversampling to obtain a real part bit sequence of each transmission symbol of the received second pilot signal

And imaginary bit sequence

Where N is an oversampling multiple of the signal, which may be 10, for example. The above real part bit sequence

And imaginary bit sequence

May be collectively referred to as a second received bit sequence

A second received bit sequence of the current transmitted symbol of the second pilot signal, e.g., obtained by the ADC 320

Is input to demodulation section 330, and demodulation section 330 may output a second received bit sequence

Inputting into a demodulating neural network as shown in FIG. 4, and obtaining an output result of the demodulating neural network

The output result

Is a modulation symbol or a dibit group of the currently transmitted symbols of the second pilot signal obtained by the demodulation neural network, which may be, for example, one of dibit groups 11,10,00, 01. The demodulation unit 330 may demodulate the neural network based on the modulation symbols of the second pilot signal

The difference between the two parameters is used for constructing a loss function, and iterative training is carried out in any appropriate mode such as a gradient descent method, so that the optimal value of each parameter of the demodulation neural network is determined when the loss function obtains the minimum value or does not change any more. Based on the demodulation neural network constructed by the present disclosure, a person skilled in the art may implement training of the demodulation neural network in any suitable manner, and details are not described here.

<3. example of information interaction Process >

Next, an example of an information interaction process of the embodiment of the present disclosure will be described with reference to fig. 5 and 6.

Referring first to fig. 5, fig. 5 is a schematic diagram for explaining a data structure of a complex signal received by an electronic device according to an embodiment of the present disclosure. As shown in fig. 5, a complex signal received by an electronic device according to an embodiment of the present disclosure may include, for example, three parts, namely, a first pilot signal for estimating a compensation phase, a second pilot signal for training a demodulation neural network, and a data signal, in sequence. As an example, here, the first pilot signal may have the exemplary form of the first pilot signal described above in the "determination of the compensation phase" section, the second pilot signal may have the exemplary form of the second pilot signal described above in the "training of the demodulation neural network" section, and the complex signal as the data signal may be an arbitrary QPSK modulated signal. A complex signal including the above three parts in order may be transmitted from the transmitting end to the receiving end at each communication.

Referring next to fig. 6, fig. 6 is a schematic diagram illustrating an example of an information interaction process of an embodiment of the present disclosure, in which a sending end and a receiving end (here, the receiving end may be, for example, the electronic device 200 or the example circuit implementation 300 thereof described hereinbefore) and information interaction between the two are schematically illustrated. Here, fig. 6 shows an example of information exchange between a transmitting end and a receiving end in, for example, one-time communication in which three parts of a complex signal such as that shown in fig. 5 are transmitted. As shown in fig. 6, first, a transmitting end transmits a first pilot signal to a receiving end. The electronic device at the receiving end determines a compensation phase from the transmission symbols of the first pilot signal and a first received bit sequence obtained based on the received first pilot signal. Then, the transmitting end transmits a second pilot signal to the receiving end. The electronic device at the receiving end obtains a demodulation neural network through training using a second received bit sequence obtained based on the received second pilot signal, which is marked with a modulation symbol of the second pilot signal.

Then, the transmitting end transmits a complex signal as a data signal to the receiving end. The electronic device at the receiving end can perform phase adjustment on the received complex signal according to the compensation phase estimated by using the first pilot signal. Then, the electronic device at the receiving end can obtain a received bit sequence based on the phase-adjusted complex signal through a low-resolution analog-to-digital conversion process. Then, the electronic device at the receiving end may demodulate the received bit sequence based on the demodulation neural network obtained by training with the second pilot signal to obtain the modulation symbols of the received complex signal.

The example flow illustrated in fig. 6 may be implemented by the electronic device 200 or the example circuit implementation 300 thereof at the receiving end and the electronic device at the transmitting end in communication therewith described above with reference to fig. 2 to 4, and thus advantages and benefits described in the configuration examples of the electronic device above may be obtained and will not be further described herein.

Note that the transmitting end (or the electronic device of the transmitting end) referred to herein may be as long as it is capable of generating and transmitting the corresponding first pilot signal, second pilot signal, and data signal and is capable of communicating in cooperation with the electronic device 200 or its exemplary circuit implementation 300 of the receiving end. Therefore, various related art electronic devices having a transmitter function can be adopted to realize the electronic device of the transmitting end by appropriate configuration. For example, when the electronic device at the receiving end is a network side device such as a base station, the electronic device at the transmitting end may be a user equipment capable of communicating with the electronic device; when the electronic device at the receiving end is a user device, the electronic device at the transmitting end may be a network side device such as a base station capable of communicating with the electronic device, and details are not described here.

<4. simulation example >

Next, simulation results regarding demodulation processing performed by the electronic apparatus of the embodiment of the present disclosure will be described with reference to fig. 7 and 8.

In the examples of fig. 7 and 8, the received and processed complex signal is a terahertz QPSK signal, and the model of the mixed distortion described heretofore in the section "description of problem" is employed. More specifically, in the examples of fig. 7 and 8, the (amplitude and phase) IQ imbalance parameters at the transmitting end and the receiving end are ∈_T＝∈_R＝0.2，φ_T＝φ_R2 °, phase noise θ of transmitting end and receiving end_TAnd theta_RE.g. from respective wandering terms according to equation (2)

And

and (4) determining. Here, the phase noise θ is set_TAnd theta_RVariance of wandering term of

Due to phase noise theta_TAnd theta_RThe run term of (A) is in accordance with Gaussian distribution, so that the receiving and transmitting end influences the generated total phase noise theta_T+θ_RWhich is also a random variable with a mean value of 0, the distribution of which can be determined according to equation (2). In addition, for the signal model of the power amplifier output in equation (3), a memoryless polynomial of three terms (K — 3) is usedModel, in which the polynomial coefficient b_2k-1(k is 1, 2, 3) is b₁＝1.0108+j0.0858，b₃＝0.0879-j0.1583，b₅-1.0992-j 0.8991. For the channel model in equation (4), the variance of Additive White Gaussian Noise (AWGN) with the channel fading factor h 1 is set

Set appropriately in the simulation according to the required signal-to-noise ratio.

Referring first to fig. 7, fig. 7 is an explanatory diagram for explaining an example simulation result of performance degradation caused by phase noise. Fig. 7 shows the influence on demodulation performance caused by total phase noise generated by the influence of the transceiving ends in the case where compensation phase estimation or phase adjustment is not performed (i.e., equivalent to the case where the functions of the phase adjustment unit 210 and the compensation phase estimation unit 240 in fig. 2 are removed) in the electronic device shown in fig. 2, where the horizontal axis represents the total phase noise θ generated by the influence of the transceiving ends_T+θ_RThe vertical axis represents the signal-to-noise ratio (E) at different_S/N₀) Lower Bit Error Rate (BER). As described previously, since the walk term of the phase noise follows the gaussian distribution, θ determined according to equation (2)_T+θ_RIs also a random variable with a mean value of 0, and theta is set in the example shown in fig. 7_T+θ_RVary randomly within a range of (-0.2 pi, 0.2 pi). The simulation results shown in FIG. 7 are for θ according to the above setting_TAnd theta_RPhase noise theta determined from the distribution of_T+θ_RIs obtained by performing a simulation of 1000 communications per phase noise within the range of (1), wherein the reception length is 10 in turn for each communication⁵And a second pilot signal of length 10⁶The data signal of (1). In this example, the electronic device of the embodiment of the present disclosure such as shown in fig. 2 does not perform compensation phase estimation or phase adjustment, but obtains a demodulation neural network through training using only the second pilot signal in each communication, and performs demodulation processing on a data signal based on the demodulation neural network. As can be seen from fig. 7, the demodulation process performed by such an electronic device is not performedSame signal-to-noise ratio (E)_S/N₀) In the following, the Bit Error Rate (BER) varies with the phase. Therefore, it is desirable to perform phase adjustment of a received complex signal with a properly determined compensation phase to achieve optimization of demodulation performance.

Referring next to fig. 8, fig. 8 is a diagram of an example simulation result for explaining the performance of demodulation processing by an electronic device according to an embodiment of the present disclosure. Fig. 8 shows a conventional hard decision (single bit quantization of a received analog signal and conversion of the quantization result into an output result of a modulation symbol) method and an error rate performance of a demodulation process of a received complex signal using an electronic device such as the one shown in fig. 2, respectively, wherein the horizontal axis represents a signal-to-noise ratio (E)_S/N₀) And the vertical axis represents the corresponding Bit Error Rate (BER). The simulation results shown in FIG. 8 are at phase noise θ_T+θ_RIn the case of random variation according to equation (2) according to the distribution set above, the result is obtained by performing a simulation of 1000 communications for each signal-to-noise ratio, where in each communication, the conventional hard decision method reception length is 10⁶The electronic device of the embodiment of the present disclosure receives the data signal with a length of 10 in sequence³Of length 10⁵And a second pilot signal of length 10⁶The data signal of (1). In this example, the electronic device of the embodiment of the present disclosure determines the compensation phase by using the first pilot signal in each communication, obtains the demodulation neural network by training by using the second pilot signal, and then performs phase adjustment and corresponding demodulation processing on the data signal. As can be seen from fig. 8, for the case involving the phase noise varying constantly, the conventional hard decision method cannot effectively implement demodulation, but the processing performed by the electronic device of the embodiment of the present disclosure solves the channel mixing distortion problem and implements accurate demodulation of the terahertz QPSK signal by performing real-time phase compensation or adjustment in each communication and accurate demodulation of the demodulation neural network, and can make the bit error rate reach 10 when the signal-to-noise ratio is 10dB^-2。

<5. method example >

The method performed in the electronic device according to the embodiment of the present disclosure will be described next in detail. Note that these method implementations correspond to the device configuration examples described above with reference to fig. 2 to 4, and therefore the respective details and benefits of the above device configuration examples are suitably applicable to the following method embodiments.

Fig. 9 is a flow chart illustrating an example of a process of a signal processing method according to an embodiment of the present disclosure, which may be implemented, for example, by the electronic device 200 described with reference to fig. 2-4 or the example circuit implementation 300 thereof.

As shown in fig. 9, first, in step S901, a phase of a received complex signal is adjusted based on a compensation phase estimated using a pilot signal. Next, in step S902, a reception bit sequence is obtained based on the phase-adjusted complex signal through a low resolution analog-to-digital conversion process. Next, in step S903, the received bit sequence is demodulated based on the demodulation neural network obtained by training with the pilot signal to obtain modulation symbols of the received complex signal.

As an example, the complex signal and the pilot signal herein may include QPSK modulated signals.

As an example, in step S901, the local oscillator may be controlled to perform phase rotation on the received complex signal according to the compensation phase, so as to implement phase adjustment.

As an example, in step S902, a real part received bit sequence and an imaginary part received bit sequence may be obtained based on the real part and the imaginary part of the phase-adjusted complex signal, respectively, through a low resolution analog-to-digital conversion process. For example, the real and imaginary parts may be oversampled to obtain a real received bit sequence and an imaginary received bit sequence for each transmitted symbol of the received complex signal.

Further, although not shown in the drawings, the method may further include: before step S901, a compensation phase is estimated from a transmission symbol of a first pilot signal and a first received bit sequence obtained based on the received first pilot signal.

For example, the method may optionally include the following processes for estimating the compensation phase: obtaining a first received bit sequence based on the phase-adjusted, received first pilot signal by a low resolution analog-to-digital conversion process; estimating a compensation phase based on an error of the first received bit sequence with respect to the transmitted symbols of the first pilot signal; and performing phase adjustment on the received first pilot signal according to the estimated compensation phase.

Alternatively, in the process for estimating the compensation phase included in the method, the compensation phase that minimizes the error may be determined as the final compensation phase.

As an example, the first pilot signal may comprise a plurality of transmission symbols corresponding to at least two adjacent constellation points in a QPSK constellation. For example, each transmission symbol in the first pilot signal may correspond to two adjacent constellation points in a QPSK constellation. For example, the respective transmission symbols in the first pilot signal may alternately correspond to the two adjacent constellation points.

Further, although not shown in the drawings, the method may further include: before step S901, a demodulation neural network is obtained by training using a second received bit sequence obtained based on the received second pilot signal, which is marked with a modulation symbol of the second pilot signal.

For example, the method may optionally include the following processes for training a demodulating neural network: performing phase adjustment on the received second pilot signal according to the compensation phase estimated by using the first pilot signal; and obtaining a second received bit sequence based on the phase-adjusted second pilot signal through a low resolution analog-to-digital conversion process.

As an example, the transmission symbols in the second pilot signal may correspond to four constellation points in a QPSK constellation. For example, each transmission symbol in the second pilot signal may randomly correspond to the four constellation points.

According to an embodiment of the present disclosure, the subject performing the above method may be the electronic device 200 or the example circuit implementation 300 thereof according to an embodiment of the present disclosure, and thus the various aspects of the embodiments described hereinbefore with respect to the electronic device 200 or the example circuit implementation 300 thereof and the functional units thereof are applicable here.

<6. application example >

The techniques of this disclosure can be applied to a variety of products.

For example, the electronic device 200 may be implemented as any type of base station device, such as a macro eNB and a small eNB, and may also be implemented as any type of gNB (base station in a 5G system). The small eNB may be an eNB that covers a cell smaller than a macro cell, such as a pico eNB, a micro eNB, and a home (femto) eNB. Alternatively, the base station may be implemented as any other type of base station, such as a NodeB and a Base Transceiver Station (BTS). The base station may include: a main body (also referred to as a base station apparatus) configured to control wireless communication; and one or more Remote Radio Heads (RRHs) disposed at a different place from the main body.

In addition, electronic device 200 may also be implemented as any type of TRP. The TRP may have transmission and reception functions, and may receive information from or transmit information to user equipment and base station equipment, for example. In a typical example, the TRP may provide a service to the user equipment and be controlled by the base station apparatus. Further, the TRP may have a structure similar to that of the base station apparatus, or may have only a structure related to transmission and reception of information in the base station apparatus.

Further, the electronic apparatus 200 may also be various user apparatuses, which may be implemented as a mobile terminal such as a smart phone, a tablet Personal Computer (PC), a notebook PC, a portable game terminal, a portable/cryptographic dog-type mobile router, and a digital camera, or a vehicle-mounted terminal such as a car navigation apparatus. The user equipment may also be implemented as a terminal (also referred to as a Machine Type Communication (MTC) terminal) that performs machine-to-machine (M2M) communication. Further, the user equipment may be a wireless communication module (such as an integrated circuit module including a single die) mounted on each of the user equipments described above.

[ application example with respect to base station ]

(first application example)

Fig. 10 is a block diagram illustrating a first example of a schematic configuration of an eNB to which the technology of the present disclosure may be applied. The eNB 1800 includes one or more antennas 1810 and base station equipment 1820. The base station device 1820 and each antenna 1810 may be connected to each other via an RF cable.

Each of the antennas 1810 includes a single or multiple antenna elements, such as multiple antenna elements included in a multiple-input multiple-output (MIMO) antenna, and is used for the base station apparatus 1820 to transmit and receive wireless signals. As shown in fig. 10, the eNB 1800 may include multiple antennas 1810. For example, the multiple antennas 1810 may be compatible with multiple frequency bands used by the eNB 1800. Although fig. 10 shows an example in which the eNB 1800 includes multiple antennas 1810, the eNB 1800 may also include a single antenna 1810.

The base station device 1820 includes a controller 1821, memory 1822, a network interface 1823, and a wireless communication interface 1825.

The controller 1821 may be, for example, a CPU or a DSP, and operates various functions of higher layers of the base station apparatus 1820. For example, the controller 1821 generates data packets from data in signals processed by the wireless communication interface 1825 and communicates the generated packets via the network interface 1823. The controller 1821 may bundle data from the plurality of baseband processors to generate a bundle packet, and communicate the generated bundle packet. The controller 1821 may have logic functions to perform the following controls: such as radio resource control, radio bearer control, mobility management, admission control and scheduling. The control may be performed in connection with a nearby eNB or core network node. The memory 1822 includes a RAM and a ROM, and stores programs executed by the controller 1821 and various types of control data (such as a terminal list, transmission power data, and scheduling data).

The network interface 1823 is a communication interface for connecting the base station apparatus 1820 to the core network 1824. The controller 1821 may communicate with a core network node or another eNB via a network interface 1823. In this case, the eNB 1800 and a core network node or other enbs may be connected to each other through a logical interface, such as an S1 interface and an X2 interface. The network interface 1823 may also be a wired communication interface or a wireless communication interface for a wireless backhaul. If network interface 1823 is a wireless communication interface, network interface 1823 may use a higher frequency band for wireless communications than the frequency band used by wireless communication interface 1825.

The wireless communication interface 1825 supports any cellular communication scheme, such as Long Term Evolution (LTE) and LTE-advanced, and provides wireless connectivity via an antenna 1810 to terminals located in the cell of the eNB 1800. The wireless communication interface 1825 may generally include, for example, a baseband (BB) processor 1826 and RF circuitry 1827. The BB processor 1826 may perform various types of signal processing of layers such as L1, Medium Access Control (MAC), Radio Link Control (RLC), and Packet Data Convergence Protocol (PDCP), for example, encoding/decoding, modulation/demodulation, and multiplexing/demultiplexing. The BB processor 1826 may have a part or all of the above-described logic functions in place of the controller 1821. The BB processor 1826 may be a memory storing a communication control program, or a module comprising a processor and associated circuitry configured to execute a program. The update program may cause the function of the BB processor 1826 to change. The module may be a card or blade that is inserted into a slot of the base station device 1820. Alternatively, the module may be a chip mounted on a card or blade. Meanwhile, the RF circuit 1827 may include, for example, a mixer, a filter, and an amplifier, and transmit and receive a wireless signal via the antenna 1810.

As shown in fig. 10, wireless communication interface 1825 may include a plurality of BB processors 1826. For example, the plurality of BB processors 1826 may be compatible with a plurality of frequency bands used by the eNB 1800. As shown in fig. 10, wireless communication interface 1825 may include a plurality of RF circuits 1827. For example, the plurality of RF circuits 1827 may be compatible with a plurality of antenna elements. Although fig. 10 shows an example in which the wireless communication interface 1825 includes multiple BB processors 1826 and multiple RF circuits 1827, the wireless communication interface 1825 may also include a single BB processor 1826 or a single RF circuit 1827.

In the eNB 1800 shown in fig. 10, the phase adjustment unit 210, the low resolution analog-to-digital conversion unit 220, and the compensated phase estimation unit 240 in the electronic device 200 described hereinbefore with reference to fig. 2 may be implemented by a wireless communication interface 1825 or the like. Here, although not shown, for example, a high-speed flip-flop or the like may be provided in the wireless communication interface 1825 to function as the low-resolution analog-to-digital conversion unit 220. The demodulation unit 230 in the electronic apparatus 200 may be realized by the controller 1821 or the like, for example. For example, the controller 1821 may perform at least a portion of the functions of the demodulation unit 230 and the like by executing instructions stored in the memory 1822, which will not be described in detail herein.

(second application example)

Fig. 11 is a block diagram illustrating a second example of a schematic configuration of an eNB to which the technology of the present disclosure may be applied. eNB 1930 includes one or more antennas 1940, base station apparatus 1950, and RRHs 1960. The RRH 1960 and each antenna 1940 may be connected to each other via an RF cable. The base station apparatus 1950 and RRH 1960 may be connected to each other via a high-speed line such as a fiber optic cable.

Each of the antennas 1940 includes a single or multiple antenna elements (such as multiple antenna elements included in a MIMO antenna) and is used for the RRH 1960 to transmit and receive wireless signals. As shown in fig. 11, eNB 1930 may include multiple antennas 1940. For example, the plurality of antennas 1940 may be compatible with a plurality of frequency bands used by eNB 1930. Although fig. 11 shows an example in which eNB 1930 includes multiple antennas 1940, eNB 1930 may also include a single antenna 1940.

The base station device 1950 includes a controller 1951, a memory 1952, a network interface 1953, a wireless communication interface 1955, and a connection interface 1957. The controller 1951, memory 1952, and network interface 1953 are the same as the controller 1821, memory 1822, and network interface 1823 described with reference to fig. 10. The network interface 1953 is a communication interface for connecting the base station device 1950 to a core network 1954.

Wireless communication interface 1955 supports any cellular communication scheme (such as LTE and LTE-advanced) and provides wireless communication via RRH 1960 and antenna 1940 to terminals located in a sector corresponding to RRH 1960. Wireless communication interface 1955 may generally include a BB processor 1956, for example. The BB processor 1956 is the same as the BB processor 1826 described with reference to fig. 10, except that the BB processor 1956 is connected to the RF circuitry 1964 of the RRH 1960 via a connection interface 1957. As shown in fig. 11, wireless communication interface 1955 may include a plurality of BB processors 1956. For example, the plurality of BB processors 1956 may be compatible with the plurality of frequency bands used by eNB 1930. Although fig. 11 shows an example in which the wireless communication interface 1955 includes multiple BB processors 1956, the wireless communication interface 1955 may include a single BB processor 1956.

Connection interface 1957 is an interface used to connect base station device 1950 (wireless communication interface 1955) to RRHs 1960. The connection interface 1957 may also be a communication module for communication in the above-described high speed lines connecting the base station device 1950 (wireless communication interface 1955) to the RRH 1960.

RRH 1960 includes connection interface 1961 and wireless communication interface 1963.

Connection interface 1961 is an interface for connecting RRH 1960 (wireless communication interface 1963) to base station apparatus 1950. The connection interface 1961 may also be a communication module for communication in the above-described high-speed line.

Wireless communication interface 1963 transmits and receives wireless signals via antenna 1940. Wireless communication interface 1963 may generally include, for example, RF circuitry 1964. The RF circuit 1964 may include, for example, mixers, filters, and amplifiers, and transmits and receives wireless signals via the antenna 1940. As shown in fig. 11, wireless communication interface 1963 may include a plurality of RF circuits 1964. For example, multiple RF circuits 1964 may support multiple antenna elements. Although fig. 11 illustrates an example in which wireless communication interface 1963 includes multiple RF circuits 1964, wireless communication interface 1963 may also include a single RF circuit 1964.

In the eNB 1930 shown in fig. 11, the phase adjustment unit 210, the low resolution analog-to-digital conversion unit 220, and the compensated phase estimation unit 240 in the electronic apparatus 200 described hereinbefore with reference to fig. 2 may be implemented by a wireless communication interface 1963. Here, although not shown, for example, a high-speed flip-flop or the like may be provided in the wireless communication interface 1963 to function as the low-resolution analog-to-digital conversion unit 220. The demodulation unit 230 in the electronic apparatus 200 may be implemented by the controller 1951 or the like. For example, the controller 1951 may perform at least a portion of the functions of the demodulation unit 230 and the like by executing instructions stored in the memory 1952, which will not be described in detail herein.

[ application example with respect to user Equipment ]

(first application example)

Fig. 12 is a block diagram showing an example of a schematic configuration of a smartphone 2000 to which the technique of the present disclosure can be applied. The smartphone 2000 includes a processor 2001, a memory 2002, a storage device 2003, an external connection interface 2004, a camera device 2006, sensors 2007, a microphone 2008, an input device 2009, a display device 2010, a speaker 2011, a wireless communication interface 2012, one or more antenna switches 2015, one or more antennas 2016, a bus 2017, a battery 2018, and an auxiliary controller 2019.

The processor 2001 may be, for example, a CPU or a system on a chip (SoC), and controls functions of an application layer and another layer of the smartphone 2000. The memory 2002 includes a RAM and a ROM, and stores data and programs executed by the processor 2001. The storage device 2003 may include a storage medium such as a semiconductor memory and a hard disk. The external connection interface 2004 is an interface for connecting an external device such as a memory card and a Universal Serial Bus (USB) device to the smartphone 2000.

The image pickup device 2006 includes an image sensor such as a Charge Coupled Device (CCD) and a Complementary Metal Oxide Semiconductor (CMOS), and generates a captured image. The sensors 2007 may include a set of sensors such as a measurement sensor, a gyro sensor, a geomagnetic sensor, and an acceleration sensor. The microphone 2008 converts sound input to the smartphone 2000 into an audio signal. The input device 2009 includes, for example, a touch sensor, a keypad, a keyboard, a button, or a switch configured to detect a touch on the screen of the display device 2010, and receives an operation or information input from a user. The display device 2010 includes a screen, such as a Liquid Crystal Display (LCD) and an Organic Light Emitting Diode (OLED) display, and displays an output image of the smartphone 2000. The speaker 2011 converts an audio signal output from the smartphone 2000 into sound.

The wireless communication interface 2012 supports any cellular communication scheme (such as LTE and LTE-advanced) and performs wireless communication. The wireless communication interface 2012 may generally include, for example, a BB processor 2013 and RF circuitry 2014. The BB processor 2013 can perform, for example, encoding/decoding, modulation/demodulation, and multiplexing/demultiplexing, and perform various types of signal processing for wireless communication. Meanwhile, the RF circuit 2014 may include, for example, a mixer, a filter, and an amplifier, and transmit and receive a wireless signal via the antenna 2016. The wireless communication interface 2012 may be a chip module on which the BB processor 2013 and the RF circuit 2014 are integrated. As shown in fig. 12, the wireless communication interface 2012 may include a plurality of BB processors 2013 and a plurality of RF circuits 2014. Although fig. 12 shows an example in which the wireless communication interface 2012 includes multiple BB processors 2013 and multiple RF circuits 2014, the wireless communication interface 2012 may also include a single BB processor 2013 or a single RF circuit 2014.

Further, the wireless communication interface 2012 may support another type of wireless communication scheme, such as a short-range wireless communication scheme, a near field communication scheme, and a wireless Local Area Network (LAN) scheme, in addition to the cellular communication scheme. In this case, the wireless communication interface 2012 may include the BB processor 2013 and the RF circuit 2014 for each wireless communication scheme.

Each of the antenna switches 2015 switches the connection destination of the antenna 916 among a plurality of circuits (e.g., circuits for different wireless communication schemes) included in the wireless communication interface 2012.

Each of the antennas 2016 includes a single or multiple antenna elements (such as multiple antenna elements included in a MIMO antenna) and is used for transmitting and receiving wireless signals by the wireless communication interface 2012. As shown in fig. 12, the smartphone 2000 may include multiple antennas 2016. Although fig. 12 shows an example in which the smartphone 2000 includes multiple antennas 2016, the smartphone 2000 may also include a single antenna 2016.

Further, the smartphone 2000 may include an antenna 2016 for each wireless communication scheme. In this case, the antenna switch 2015 may be omitted from the configuration of the smartphone 2000.

The bus 2017 connects the processor 2001, the memory 2002, the storage device 2003, the external connection interface 2004, the image pickup device 2006, the sensor 2007, the microphone 2008, the input device 2009, the display device 2010, the speaker 2011, the wireless communication interface 2012, and the auxiliary controller 2019 to each other. The battery 2018 provides power to the various blocks of the smartphone 2000 shown in fig. 12 via a feed line, which is partially shown in the figure as a dashed line. The supplementary controller 2019 operates the minimum necessary functions of the smartphone 2000 in, for example, a sleep mode.

In the smartphone 2000 shown in fig. 12, the phase adjustment unit 210, the low-resolution analog-to-digital conversion unit 220, and the compensation phase estimation unit 240 in the electronic apparatus 200 described hereinbefore with reference to fig. 2 may be implemented by a wireless communication interface 2012. Here, although not shown, for example, a high-speed flip-flop or the like may be provided in the wireless communication interface 2012 to serve as the low-resolution analog-to-digital conversion unit 220. The demodulation unit 230 in the electronic apparatus 200 may be implemented by the processor 2001 or the auxiliary controller 2019. For example, the processor 2001 or the auxiliary controller 2019 may perform at least a part of the functions of the demodulation unit 230 and the like by executing instructions stored in the memory 2002 or the storage device 2003, which will not be described herein.

(second application example)

Fig. 13 is a block diagram showing an example of a schematic configuration of a car navigation device 2120 to which the technique of the present disclosure can be applied. Car navigation device 2120 includes a processor 2121, memory 2122, a Global Positioning System (GPS) module 2124, sensors 2125, a data interface 2126, a content player 2127, a storage medium interface 2128, an input device 2129, a display device 2130, speakers 2131, a wireless communication interface 2133, one or more antenna switches 2136, one or more antennas 2137, and a battery 2138.

The processor 2121 may be, for example, a CPU or an SoC, and controls a navigation function and another function of the car navigation device 2120. The memory 2122 includes a RAM and a ROM, and stores data and programs executed by the processor 2121.

The GPS module 2124 measures the position (such as latitude, longitude, and altitude) of the car navigation device 2120 using GPS signals received from GPS satellites. The sensors 2125 may include a set of sensors, such as a gyro sensor, a geomagnetic sensor, and an air pressure sensor. The data interface 2126 is connected to, for example, the in-vehicle network 2141 via a terminal not shown, and acquires data generated by the vehicle (such as vehicle speed data).

The content player 2127 reproduces content stored in a storage medium (such as a CD and a DVD) inserted into the storage medium interface 2128. The input device 2129 includes, for example, a touch sensor, a button, or a switch configured to detect a touch on the screen of the display device 2130, and receives an operation or information input from a user. The display device 2130 includes a screen such as an LCD or OLED display, and displays an image of a navigation function or reproduced content. The speaker 2131 outputs the sound of the navigation function or the reproduced content.

The wireless communication interface 2133 supports any cellular communication schemes (such as LTE and LTE-advanced) and performs wireless communication. The wireless communication interface 2133 may generally include, for example, a BB processor 2134 and RF circuitry 2135. The BB processor 2134 may perform, for example, encoding/decoding, modulation/demodulation, and multiplexing/demultiplexing, and perform various types of signal processing for wireless communication. Meanwhile, the RF circuit 2135 may include, for example, a mixer, a filter, and an amplifier, and transmits and receives a wireless signal via the antenna 2137. The wireless communication interface 2133 may also be one chip module on which the BB processor 2134 and the RF circuit 2135 are integrated. As shown in fig. 13, the wireless communication interface 2133 may include a plurality of BB processors 2134 and a plurality of RF circuits 2135. Although fig. 13 shows an example in which the wireless communication interface 2133 includes a plurality of BB processors 2134 and a plurality of RF circuits 2135, the wireless communication interface 2133 may also include a single BB processor 2134 or a single RF circuit 2135.

Further, the wireless communication interface 2133 may support another type of wireless communication scheme, such as a short-range wireless communication scheme, a near field communication scheme, and a wireless LAN scheme, in addition to the cellular communication scheme. In this case, the wireless communication interface 2133 may include a BB processor 2134 and RF circuitry 2135 for each wireless communication scheme.

Each of the antenna switches 2136 switches a connection destination of the antenna 2137 among a plurality of circuits (such as circuits for different wireless communication schemes) included in the wireless communication interface 2133.

Each of the antennas 2137 includes a single or multiple antenna elements (such as multiple antenna elements included in a MIMO antenna), and is used for the wireless communication interface 2133 to transmit and receive wireless signals. As shown in fig. 13, the car navigation device 2120 may include a plurality of antennas 2137. Although fig. 13 shows an example in which the car navigation device 2120 includes the plurality of antennas 2137, the car navigation device 2120 may include a single antenna 2137.

Further, the car navigation device 2120 may include an antenna 2137 for each wireless communication scheme. In this case, the antenna switch 2136 may be omitted from the configuration of the car navigation device 2120.

The battery 2138 supplies power to the respective blocks of the car navigation device 2120 shown in fig. 13 via a feeder, which is partially shown as a dotted line in the drawing. The battery 2138 accumulates electric power supplied from the vehicle.

In the car navigation device 2120 shown in fig. 13, the phase adjustment unit 210, the low resolution analog-to-digital conversion unit 220, and the compensated phase estimation unit 240 in the electronic device 200 described hereinbefore with reference to fig. 2 may be implemented by the wireless communication interface 2133. Here, although not shown, for example, a high-speed flip-flop or the like may be provided in the wireless communication interface 2133 to function as the low-resolution analog-to-digital conversion unit 220. The demodulation unit 230 in the electronic device 200 may be implemented by the processor 2121. For example, processor 2121 may perform at least a portion of the functions of demodulation unit 230 by executing instructions stored in memory 2122, which are not described herein.

The techniques of this disclosure may also be implemented as an in-vehicle system (or vehicle) 2140 that includes one or more blocks of a car navigation device 2120, an in-vehicle network 2141, and a vehicle module 2142. The vehicle module 2142 generates vehicle data (such as vehicle speed, engine speed, and fault information) and outputs the generated data to the on-board network 2141.

The preferred embodiments of the present disclosure are described above with reference to the drawings, but the present disclosure is of course not limited to the above examples. Various changes and modifications within the scope of the appended claims may be made by those skilled in the art, and it should be understood that these changes and modifications naturally will fall within the technical scope of the present disclosure.

For example, the units shown in the functional block diagrams in the figures as dashed boxes each indicate that the functional unit is optional in the corresponding apparatus, and the respective optional functional units may be combined in an appropriate manner to implement the required functions.

For example, a plurality of functions included in one unit may be implemented by separate devices in the above embodiments. Alternatively, a plurality of functions implemented by a plurality of units in the above embodiments may be implemented by separate devices, respectively. In addition, one of the above functions may be implemented by a plurality of units. Needless to say, such a configuration is included in the technical scope of the present disclosure.

In this specification, the steps described in the flowcharts include not only the processing performed in time series in the described order but also the processing performed in parallel or individually without necessarily being performed in time series. Further, even in the steps processed in time series, needless to say, the order can be changed as appropriate.

Further, the present disclosure may have a configuration as described below.

(1) An electronic device, comprising:

a processing circuit configured to:

performing phase adjustment on the received complex signal according to the compensation phase estimated by using the pilot signal;

obtaining a received bit sequence based on the phase-adjusted complex signal through a low resolution analog-to-digital conversion process; and

the received bit sequence is demodulated based on a demodulation neural network obtained by training with a pilot signal to obtain modulation symbols of the received complex signal.

(2) The electronic device of (1), wherein the complex signal and the pilot signal comprise QPSK modulated signals.

(3) The electronic device of (2), wherein the processing circuitry is further configured to:

the compensation phase is estimated from the transmitted symbols of the first pilot signal and a first received bit sequence obtained based on the received first pilot signal.

(4) The electronic device of (3), wherein the processing circuitry is further configured to:

obtaining a first received bit sequence based on the phase-adjusted, received first pilot signal by a low resolution analog-to-digital conversion process;

estimating a compensation phase based on an error of the first received bit sequence with respect to the transmitted symbols of the first pilot signal; and

the received first pilot signal is phase adjusted based on the estimated compensation phase.

(5) The electronic device of (4), wherein the processing circuit is further configured to determine a compensation phase that minimizes the error as a final compensation phase.

(6) The electronic device of (3), wherein the first pilot signal comprises a plurality of transmission symbols corresponding to at least two adjacent constellation points in a QPSK constellation.

(7) The electronic device of (3), wherein each transmission symbol in the first pilot signal corresponds to two adjacent constellation points in a QPSK constellation.

8) The electronic device of (7), wherein the respective transmission symbols in the first pilot signal alternately correspond to the two adjacent constellation points.

(9) The electronic device of (1), wherein the processing circuitry is further configured to: and controlling the local oscillator to perform phase rotation on the received complex signal according to the compensation phase so as to realize phase adjustment.

(10) The electronic device of (1), wherein the processing circuit is further configured to obtain a demodulation neural network through training using a second received bit sequence obtained based on the received second pilot signal, marked with a modulation symbol of the second pilot signal.

(11) The electronic device of (10), wherein the processing circuitry is further configured to:

performing phase adjustment on the received second pilot signal according to the compensation phase estimated by using the first pilot signal; and

a second received bit sequence is obtained based on the phase-adjusted second pilot signal through a low resolution analog-to-digital conversion process.

(12) The electronic device of (10), wherein the transmission symbols in the second pilot signal correspond to four constellation points in a QPSK constellation.

(13) The electronic device of (12), wherein each transmission symbol in the second pilot signal randomly corresponds to the four constellation points.

(14) The electronic device of any of (1) to (13), wherein the processing circuit is further configured to obtain a real part received bit sequence and an imaginary part received bit sequence based on a real part and an imaginary part of the phase-adjusted complex signal, respectively, through a low resolution analog-to-digital conversion process.

(15) The electronic device of (14), wherein the processing circuit is further configured to oversample the real and imaginary parts to obtain a real and imaginary received bit sequence for each transmitted symbol of the received complex signal.

(16) A signal processing method, comprising:

performing phase adjustment on the received complex signal according to the compensation phase estimated by using the pilot signal;

obtaining a received bit sequence based on the phase-adjusted complex signal through a low resolution analog-to-digital conversion process; and

the received bit sequence is demodulated based on a demodulation neural network obtained by training with a pilot signal to obtain modulation symbols of the received complex signal.

(17) The signal processing method as set forth in (16), wherein the complex signal and the pilot signal include QPSK modulated signals.

(18) The signal processing method according to (16), further comprising:

the compensation phase is estimated from the transmitted symbols of the first pilot signal and a first received bit sequence obtained based on the received first pilot signal.

(19) The signal processing method according to (16), further comprising:

the demodulation neural network is obtained by training with a second received bit sequence obtained based on the received second pilot signal, marked with a modulation symbol of the second pilot signal.

(20) A non-transitory computer-readable storage medium storing a program that, when executed by a processor, causes the processor to perform the method according to any one of (17) to (19).

Although the embodiments of the present disclosure have been described in detail with reference to the accompanying drawings, it should be understood that the above-described embodiments are merely illustrative of the present disclosure and do not constitute a limitation of the present disclosure. It will be apparent to those skilled in the art that various modifications and variations can be made in the above-described embodiments without departing from the spirit and scope of the disclosure. Accordingly, the scope of the disclosure is to be defined only by the claims appended hereto, and by their equivalents.

Claims (10)

1. An electronic device, comprising:

a processing circuit configured to:

performing phase adjustment on the received complex signal according to the compensation phase estimated by using the pilot signal;

obtaining a received bit sequence based on the phase-adjusted complex signal through a low resolution analog-to-digital conversion process; and

the received bit sequence is demodulated based on a demodulation neural network obtained by training with a pilot signal to obtain modulation symbols of the received complex signal.

2. The electronic device of claim 1, wherein the complex signal and the pilot signal comprise QPSK modulated signals.

3. The electronic device of claim 2, wherein the processing circuit is further configured to:

the compensation phase is estimated from the transmitted symbols of the first pilot signal and a first received bit sequence obtained based on the received first pilot signal.

4. The electronic device of claim 3, wherein the processing circuit is further configured to:

obtaining a first received bit sequence based on the phase-adjusted, received first pilot signal by a low resolution analog-to-digital conversion process;

estimating a compensation phase based on an error of the first received bit sequence with respect to the transmitted symbols of the first pilot signal; and

the received first pilot signal is phase adjusted based on the estimated compensation phase.

5. The electronic device of claim 4, wherein the processing circuit is further configured to determine a compensation phase that minimizes the error as a final compensation phase.

6. The electronic device of claim 3, wherein the first pilot signal comprises a plurality of transmission symbols corresponding to at least two adjacent constellation points in a QPSK constellation.

7. The electronic device of claim 3, wherein each transmission symbol in the first pilot signal corresponds to two adjacent constellation points in a QPSK constellation.

8. The electronic device of claim 7, wherein the respective transmission symbols in the first pilot signal alternately correspond to the two adjacent constellation points.

9. The electronic device of claim 1, wherein the processing circuit is further configured to: and controlling the local oscillator to perform phase rotation on the received complex signal according to the compensation phase so as to realize phase adjustment.

10. The electronic device of claim 1, wherein the processing circuit is further configured to obtain a demodulation neural network through training using a second received bit sequence obtained based on the received second pilot signal, marked with a modulation symbol of the second pilot signal.

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