CN114362543B - Three-phase single-stage resonant conversion device - Google Patents

Abstract

The invention provides a three-phase single-stage resonant conversion device which comprises a power unit and a control unit, wherein the control unit collects input voltage, input current and output voltage signals of the power unit and provides driving signals for a switch in the power unit. The invention describes the change condition of each variable in all switching periods in the energy conversion process of the three-phase single-stage resonant conversion device by using a numerical calculation method, and designs a control scheme of the three-phase single-stage resonant conversion device, which is simple and effective.

Description

Three-phase single-stage resonant conversion device

Technical Field

The invention belongs to the technical field of alternating current-direct current electric energy conversion, and particularly relates to a three-phase single-stage resonant conversion device.

Background

With the explosive growth of distributed energy storage, micro-grid and electric car and grid (V2G) applications, isolated AC/DC converters have attracted considerable attention in industry and academia. How to improve an AC/DC converter to obtain high conversion efficiency, a compact structure, and low cost is a research hot spot in recent years.

Compared with the current most common AC/DC two-stage strategy (the front stage is a PFC converter and the rear stage is a DC/DC converter), the single-stage AC/DC converter can achieve the aim of improving the overall efficiency and the power density of the converter by reducing the energy conversion level and removing the middle direct current bus capacity. In recent years, there are two types of single-stage AC/DC conversion strategies that have been most studied, i.e., single-stage AC/DC converters based on a phase shift control strategy and on a matrix conversion strategy, respectively. Although the strategies can realize AC/DC energy conversion through a complex control strategy, the aim that all switching tubes are soft switches in a full power range cannot be fulfilled, and popularization and application of the converters are limited.

Disclosure of Invention

The invention provides a three-phase single-stage resonance conversion device, which comprises a power unit and a control unit, wherein the control unit collects input voltage, input current and output voltage signals of the power unit and provides driving signals for a switch in the power unit, the power unit comprises a switch circuit, a resonance circuit and a rectification filter circuit, the switch circuit, the resonance circuit and the rectification filter circuit are sequentially connected in series, the input end of the switch circuit is connected with a three-phase power supply, the phase with the largest voltage absolute value is set as L phase, the phase with the smallest voltage absolute value is set as S phase, the other phase is set as M phase, the output end of the rectification filter circuit is connected with load,

the control unit comprises a voltage control loop and a current control loop, wherein the voltage control loop samples the output voltage and generates a three-phase current reference peak value after feedback adjustment with a reference voltage, the three-phase current reference peak value is input into the current control loop, the three-phase current reference peak value is multiplied by phase information of L phases and M phases to obtain L-phase reference currents and M-phase reference currents of the two phases, and the current control loop controls the L-phase and M-phase current to track the L-phase and M-phase reference currents.

The switching circuit comprises a first bridge arm, a second bridge arm and a third bridge arm, wherein the first bridge arm, the second bridge arm and the third bridge arm are connected in parallel, the first bridge arm comprises a first switch and a first capacitor, the first switch and the first capacitor are connected in series, the first switch is a bidirectional switch, the second bridge arm comprises a second switch and a second capacitor, the second switch and the second capacitor are connected in series, the second switch is a bidirectional switch, the third bridge arm comprises a third switch and a third capacitor, the third switch and the third capacitor are connected in series, the third switch is a bidirectional switch, one ends of the first capacitor, the second capacitor and the third capacitor are connected in parallel, and the other ends of the first capacitor, the second capacitor and the third capacitor are connected with three-phase alternating current through a first inductor, a second inductor and a third inductor respectively.

The resonant circuit comprises a fourth inductor, a transformer and a fourth capacitor which are sequentially connected in series, wherein a primary winding of the transformer is connected with the fourth inductor and the fourth capacitor in series, and a secondary winding of the transformer is connected with the rectifying and filtering circuit in parallel.

The L-phase reference current and the L-phase sampling current are subjected to feedback regulation to generate a switching frequency f _s 。

The M-phase reference current and the M-phase sampling current are subjected to feedback regulation to generate a duty ratio D _M 。

The L phase occupies half of the switching period, and the M phase and the S phase share the rest half period.

The invention also provides a numerical calculation method of the three-phase single-stage resonant conversion device, which comprises the following steps:

step S10: dividing a static period into N equally divided phase intervals, and selecting an LLC switching period with asymmetric input voltage in each phase interval for numerical calculation;

step S20: numerical calculation is carried out on the 1 st LLC switching period and the N th LLC switching period in the static period, and the working time delta t of three input voltages in the 1 st and the N th LLC switching periods is respectively solved _L 、Δt _M 、Δt _S And i _Lrn 、i _Lmn 、u _Crn 、u _Lm And respectively determining the operation modes of the 1 st LLC switching period and the N-th LLC switching period;

step S30: according to the operation modes and the variable values of the 1 st and the N th LLC switching periods calculated in the step S03, sequentially carrying out numerical calculation on the 2 nd to the N-1 st LLC switching periods, and respectively solving the working time delta t of three input voltages in each LLC switching period _L 、Δt _M 、Δt _S And i _Lrn 、 i _LMn 、U _crn 、U _LM And determining the operation mode of each LLC switching period;

step S40: obtaining switching frequency F at different voltage output gains and input power levels _S And D _M Is a law of variation of (c).

The step S01 further includes defining three asymmetric input voltages of the three-phase single-stage AC/DC resonant converter as U according to the voltage amplitude _L 、U _M 、U _S The U is _L Voltage direction and U of (2) _M 、U _S Is opposite to the voltage direction of, |U _L |＝|U _M |+|U _S I and U _L |＞|U _M |＞|U _S I, t in any one switching period _L ＝t _M +t _S 。

The running mode of the LLC switching period is orderly composed of three basic states of a P state, an N state and an O state, wherein,

the P state is: LLC harmonicsOutput voltage u of vibration cavity _o And input voltage u _i The directions are the same, resonant inductance L _r And a resonance capacitor C _r Keep resonance, excitation inductance L _m Is subjected to a voltage u _o Clamping;

the N state is: output voltage u of LLC resonant cavity _o And input voltage u _i Opposite direction, resonant inductance L _r And a resonance capacitor C _r Keep resonance, excitation inductance L _m The clamping voltage direction of (2) is changed;

the O state is: LLC resonant cavity and transformer secondary side direct current side disconnection, excitation inductance L _m Added to the resonant inductance L _r And a resonance capacitor C _r During resonance of (a).

The step S02 further includes sorting out each variable time domain expression and each variable equivalent equation in each LLC switching cycle requiring numerical calculation, where the variable equivalent equation follows the following rule:

1) Resonant current i in all LLC switching cycles _Lr And capacitance voltage u _Cr The continuous change is kept all the time, and the mutation at any time is avoided;

2) Energy generated by each phase of input current and input energy conservation of a corresponding LLC resonant cavity, and meanwhile, input energy and output energy conservation of the resonant cavity;

3)D _L ＝D _M +D _S =0.5, where D _L For input voltage U _L Duty cycle of D _M For inputting voltage U _M Duty cycle of D _S For input voltage U _S Duty cycle of (2);

4) The sum of the three input currents provided by the three-phase power supply to the AC/DC resonant converter is 0 during one LLC switching cycle.

The three-phase single-stage resonant conversion device controls the transmission energy of the resonant converter by adjusting the switching frequency so as to control the input current and the output voltage, and the converter realizes the soft switching of the high-frequency switch by using the resonant converter, so that the switching loss is reduced, and the higher converter efficiency is obtained.

Drawings

In order to more clearly illustrate the technical solutions of the embodiments of the present invention, the drawings that are needed to be used in the embodiments of the present invention will be briefly described below, and it is obvious that the drawings described below are only some embodiments of the present invention, and other drawings may be obtained according to these drawings for those skilled in the art.

Fig. 1 is a schematic circuit diagram of a three-phase single-stage resonant conversion device according to the present invention.

Fig. 2 is a flowchart of the numerical calculation method of the present invention.

Fig. 3 is a schematic diagram of a three-phase asymmetric input voltage.

Fig. 4 (a) to 4 (f) are schematic views of an operation mode of the resonant conversion device.

Fig. 5 (a) to 5 (c) are graphs showing output voltage gain curves.

Fig. 6 (a) to 6 (b) are graphs showing changes in output power curves.

Fig. 7 is a graph of input and output of the resonant conversion device.

Fig. 8 (a) to 8 (b) are experimental operation mode diagrams of LLC resonant cavities.

Fig. 9 is an experimental plot of switching frequency versus M-phase duty cycle.

Fig. 10 is a waveform diagram of input voltage of the resonant circuit of the present invention.

FIG. 11 illustrates the determination of the operating region of the converter within a power frequency cycle in accordance with the present invention.

Fig. 12 (a) to 12 (b) are timing diagrams of the switch in one working area of the present invention.

Detailed Description

The technical solutions of the present invention will be clearly and completely described below with reference to the accompanying drawings of the embodiments of the present invention. It will be apparent that the described embodiments are some, but not all, embodiments of the invention. All other embodiments, which can be made by a person skilled in the art without creative efforts, based on the described embodiments of the present invention fall within the protection scope of the present invention.

As shown in fig. 1, a three-phase single-stage resonant conversion device comprises a power unit 10 and a control unitThe control unit 20 collects key signals of the power unit 10 and provides driving signals for switches in the power unit 10. The power unit 10 comprises a switch circuit 11, a resonant circuit 12 and a rectifying and filtering circuit 13, wherein the switch circuit 11, the resonant circuit 12 and the rectifying and filtering circuit 13 are sequentially connected in series, the input end of the switch circuit 11 is connected with a three-phase power supply, and the output end of the rectifying and filtering circuit 13 is connected with a load R _o 。

The switch circuit 11 comprises a bridge arm 111, a bridge arm 112 and a bridge arm 113, wherein the bridge arm 111, the bridge arm 112 and the bridge arm 113 are connected in parallel. The bridge arm 111 includes a switch Sa and a capacitor Ca, where the switch Sa is a bidirectional switch, and the switch Sa includes a switch Sau and a switch Sad, and the switch Sau and the switch Sad are unidirectional switches and are connected in reverse series. The bridge arm 112 and the bridge arm 113 have the same structure. One end of the capacitor Ca, one end of the capacitor Cb and one end of the capacitor Cc are connected in parallel, and the other end of the capacitor Ca, the capacitor Cb and the capacitor Cc are connected with the three-phase alternating current Uabc through the inductor La, the inductor Lb and the inductor Lc respectively.

The resonant circuit 2 comprises inductances L connected in series in turn _r Transformer T and capacitor C _r . Primary winding of the transformer T and the inductor L _r And capacitor C _r In series, the secondary winding of the transformer T is connected in parallel with the rectifying and filtering circuit 13. In this embodiment, the rectifying and filtering circuit 13 is a switch D ₁ 、D ₂ The full wave rectifying circuit can also be a full wave or half wave rectifying circuit with other circuit structures, and the rectifying and filtering circuit 3 also comprises a capacitor C ₀ . The capacitor C ₀ And switch D ₁ 、D ₂ The output ends of the two-phase DC voltage source are connected in parallel, and the two ends are connected in parallel with a load Ro to output a DC voltage Vo.

The invention provides a numerical calculation method of a three-phase single-stage resonant conversion device, which is used for analyzing the working characteristics of the three-phase single-stage resonant conversion device, and is applied to the three-phase single-stage resonant conversion device shown in fig. 1, but is not limited to the three-phase single-stage resonant conversion device shown in fig. 1.

Typically, input intoThe three-phase input voltage in the resonant conversion device is asymmetric and the operation of each switching cycle becomes different for different forms of input voltage. In the present invention, three asymmetric input voltages are defined as U according to voltage amplitude _L 、U _M 、U _S ，U _L Represents the maximum voltage amplitude item, U, of three input voltages _s Represents the smallest voltage amplitude among three input voltages, and the rest is U _M I.e. |U _L |＞|U _M |＞|U _S I, at the same time, U _L Voltage direction and U of (2) _M 、U _S Is opposite in voltage direction and |U _L |＝|U _M |+|U _S I, U in any one switching period _L Is equal to U in operation time _M And U _s The sum of the working times of (a), t _L ＝t _M +t _S 。

In the invention, all switching cycles of the three-phase single-stage resonant conversion device can be orderly composed of three basic states, wherein the three basic states are respectively as follows:

1) P state: output voltage V _o And input voltage U _i The same direction, inductance L _r And capacitor C _r Keep resonance, excitation inductance L _m Is output with voltage V _o Clamping;

2) And N state: output voltage V _o And input voltage U _i Opposite direction, inductance L _r And capacitor C _r Keep resonance, excitation inductance L _m The clamping voltage direction of (2) is changed;

3) O state: the secondary side of the transformer T is disconnected with the direct current side of the rectifying and filtering circuit 3, and the exciting inductance L _m Added to inductance L _r And capacitor C _r During resonance of (a).

As shown in fig. 2, the method for calculating the numerical value of the three-phase single-stage resonant conversion device mainly comprises the following steps:

step S10: dividing a static period into N equally divided phase intervals, and selecting an asymmetrical switching period of input voltage in each phase interval for numerical calculation.

Dividing a static period into N equal parts, i.e. dividing the external conditions of the resonant conversion device, such as three-phase input voltage, input power and output voltage, into N equal parts, where the duration of each equal part of the phase interval is relatively short, and the external conditions of the switching periods (such as three-phase input voltage, input power and output voltage) included in each equal part of the phase interval may be considered to be almost the same, and it is presumed that the operation conditions, such as switching frequency and operation mode, of all the switching periods in each equal part of the phase interval are substantially similar.

Step S20: numerical calculation is carried out on the 1 st switching period and the N th switching period in the static period, and working time delta t of three input voltages in the 1 st and the N th switching periods is respectively solved _L 、Δt _M 、Δt _S And i _Lrn 、i _Lmn 、u _Crn 、u _Lm And respectively determining the operation modes of the 1 st switching period and the N th switching period.

Step S30: according to the operation modes and the variable values of the 1 st and the N th switching periods calculated in the step S03, sequentially carrying out numerical calculation on the 2 nd to the N-1 st switching periods, and respectively solving the working time delta t of three input voltages in each switching period _L 、Δt _M 、Δt _S And i _Lrn 、i _Lmn 、u _crn 、 u _Lm And determines the mode of operation in which each switching cycle is located.

Step S40: obtaining switching frequency f at different voltage output gains and input power levels _s And D _M Is a law of variation of (c).

Since the LLC switching frequency of the three-phase single-stage resonant conversion device is about 100kHz and is far greater than the variation frequency (about 50 Hz) of the input/output voltage and current in the resonant conversion device, the input/output voltage, current and power are assumed to be constant when each switching period is analyzed.

In step S20, the variables (including i _Lr 、i _Lm 、u _Cr 、u _Lm Etc.) and equivalent equations, the time domain expressions of the variables written in the columns can be directly obtained according to the resonance process under different states, and the equivalent equations of the variables written in the columns follow the following rules:

1) Resonant current i in all switching cycles _Lr And capacitance voltage u _cr The continuous change is kept all the time, and the mutation at any time is avoided;

2) Energy generated by each phase of input current and input energy conservation of a corresponding LLC resonant cavity, and meanwhile, the LLC resonant cavity inputs energy and outputs energy conservation;

3)D _L ＝D _M +D _S =0.5, where D _L For input voltage U _L Duty cycle of D _M For inputting voltage U _M Duty cycle of D _S For input voltage U _S Duty cycle of (2);

4) In one switching period, the sum of three input currents provided by the three-phase power supply to the resonant conversion device is 0, namely the resonant capacitor voltage u _Cr The variation is 0.

After finishing the time domain expression and equivalent equation of each variable in each switching period needing to be numerically calculated, selecting the 1 st and N th switching periods as the starting point and the end point of the numerical calculation, and inputting the input voltage U corresponding to the working points in the 1 st and N th switching periods _L 、U _M 、U _S Substituting the time domain expression and the equivalent equation of each variable of the symmetrical switching period operation modes to respectively solve the working time delta t of three input voltages in the 1 st and N th switching periods _L 、Δt _M 、Δt _S And i _Lrn 、i _Lmn 、u _Crn 、u _Lm And then determining an initial operation mode and a termination operation mode according to the calculation result. When determining the operation mode of the switching period, the calculation result needs to meet the following two conditions at the same time: (1) the operating time deltat of the three input voltages in the switching cycle _L 、Δt _M 、Δt _s Are all greater than 0; (2) resonant current i _Lr And capacitance voltage u _Cr Continuously changing all the time.

In step S30, since the time interval between the adjacent switching periods is very short, the corresponding input voltage changes very little, so when the number of the other switching periods (i.e. the 2 nd to N-1 st switching periods) is calculated, the invention considers that the operation modes of the two adjacent switching periods for numerical calculation are the same, the resonant states of the positive and negative half periods have the same composition and arrangement, and the operation states of the resonant converters corresponding to the two different input voltages in the same half period are all kept continuously changed.

And (3) presuming that an asymmetric switching period mode possibly appears in the static period according to the initial operation mode and the termination operation mode which are obtained in the step S20. In step S30, the numerical calculation is performed on the remaining switching periods (i.e., the 2 nd to N-1 st switching periods), and mainly includes the following steps:

step S31: applying the operation mode of the last switching period to the numerical calculation of the current switching period;

step S32: analyzing all the numerical calculation results of the current switching period to judge whether all the calculation results of the current switching period meet the following conditions at the same time: (1) the operating time deltat of the three input voltages in the switching cycle _L 、Δt _M 、Δt _s Are all greater than 0; (2) resonant current i _Lr And capacitance voltage u _Cr Continuously changing all the time;

step S33: if all calculation results of the current switching period meet the conditions, the operation mode of the last switching period can be applied to the numerical calculation of the current LLC switching period, and the numerical operation of the next LLC switching period can still be applied to write an equation;

step S34: if all the calculation results of the current switching period do not meet the conditions, the method is switched to the equation of the next operation mode until the correct calculation result is found.

The following describes a numerical calculation method of a three-phase single-stage resonant conversion device according to a specific embodiment, as shown in fig. 3, the numerical calculation static period of the resonant conversion device is selected as [ pi/2-pi/3 ]]，U _L 、U _M 、U _S Respectively three input voltages, |U _L |＞|U _M |＞|U _S |，U _L Voltage direction and U of (2) _M 、U _S Is opposite in voltage direction and |U _L |＝|U _M |+|U _S |，i _L 、i _M 、i _S Is the input current corresponding to the three input voltages.

The numerical calculation of the invention is carried out by adopting MATLAB software. The static period [ pi/3-pi/2 ]]Divided into n=60 equally divided phase intervals, as shown in fig. 3, when the phase angle θ is pi/3 and pi/2, the input voltages of the switching periods are symmetrical, so that two operating points (the start phase and the end phase in the static period) of pi/3 and pi/2 are selected as the starting points of numerical calculation in the whole static period, namely, the 1 st switching period and the 60 th switching period, the input voltages (U _L 、U _M 、U _S ) Substituting the phase angle theta into the equation of the same quantity of each variable of the symmetrical switching period to calculate, and determining the initial operation mode and the termination operation mode when the phase angle theta is pi/3 and pi/2. Then sequentially carrying out numerical calculation on the 2 nd to 59 th switching periods according to the initial operation mode and the termination operation mode to obtain the working time delta t of three input voltages of each switching period _L 、Δt _M 、Δt _S And i _Lrn 、i _Lmn 、u _Crn 、u _Lm And (4) equally variable, determining the operation modes of each switching period, and finally obtaining six conditions shown in fig. 4.

Under the condition that the output voltage gain is larger than 1 and the load is relatively heavy, the switching period is in a PO/P/PO mode as shown in fig. 4 (a), and the switching period is changed into a PO/PO/O mode under some phases as shown in fig. 4 (b); likewise, in the case where the output voltage gain is smaller than 1 and the load is relatively heavy, the switching period is NP/P mode, as shown in fig. 4 (c); if the load condition is relatively light, the output voltage gain is less than 1, the corresponding switching period is NOP/NOP/P mode, as shown in FIG. 4 (d); when the output voltage gain is greater than 1, the corresponding switching periods may be OPO/OP/PO mode and OPO/O mode, as shown in fig. 4 (e) and fig. 4 (f). It can be seen that in FIGS. 4 (a) to 4 (f), the positive and negative half periods are followedThe input voltage waveform is asymmetric, the duration of all resonance states is greater than 0, and the resonance current i _Lrn And resonant capacitance voltage u _Crn Are kept continuous. In addition, under the influence of DC bias voltage on the resonance capacitance voltage, and U _L Resonant current i in corresponding half period _Lrn Is lower than U _M And U _S The sum of the corresponding resonant current peaks; when the input voltage is from U _M Becomes U-shaped _S When the resonance state and the variable are kept continuous, the change trend is changed; u in different operation modes within half period _M The corresponding working times are significantly different.

The numerical relationships among the variables in the three-phase single-stage resonant conversion device of the present invention can also be obtained by the above numerical calculations, as shown in fig. 5 (a) to 5 (c) and fig. 6 (a) to 6 (b). The voltage output gain m= nUo/Urms of the three-phase single-stage resonant conversion device, where Uo is the output voltage, urms is the effective value of the input voltage, the output voltage gain M represents the output voltage condition of the resonant conversion device, and the output power Po may represent the input current level of the converter. Different L-phase input phases theta _L The switching frequency fn (fs/fr) of the switching period and the duty ratio D of the M phase are adjusted corresponding to three different input voltages _M The asymmetric LLC resonant cavity can output different output voltages and three symmetric input currents. Therefore, in the three-phase single-stage resonant conversion device, there is an equal relationship [ M, P ] _o ]＝F(θ _L ,f _n ,D _M ). As shown in fig. 5 (a) to (c), the X-axis is the switching frequency fn, the Y-axis is the duty ratio, the Z-axis is the L-phase input phase, each curve represents an output voltage gain M, any one of the X-Y planes represents an input voltage combination, and the intersection point of each curve and the X-Y plane represents the switching frequency fn and the duty ratio D of the M-phase _M The gain M of the resonant converter under action is distributed. As shown in fig. 6 (a) and (b), the three curves in each graph represent the output power at full load, half load and 10% load, respectively, where the X-axis is the switching frequency fn, the Y-axis is the M-phase duty cycle, the Z-axis is the output voltage gain M, and the L-phase input phase θ in fig. 6 (a) _L =4pi/9, L-phase input phase θ in fig. 6 (b) _L ＝7π/18。

Comparing fig. 5 (a) to 5 (c) and fig. 6 (a) to 6 (b), it can be found that: 1) In each static period, as the L-phase input phase changes from pi/2 to pi/3, the switching frequency fn and M-phase duty cycle D need to be adjusted _M To maintain the output voltage and input current of the three-phase single-stage resonant conversion device; 2) The three-phase single-stage resonant conversion device with the same output power can obtain higher output voltage gain M by reducing the switching frequency fn and the M-phase duty ratio; 3) The higher the output power, the lower the asymmetric LLC resonant cavity gain for the same switching frequency fn for the same input. In addition, it can be found that the switching frequency fn, the M-phase duty ratio, and the switching frequency fn and the M-phase duty ratio of the two start/end switching periods pi/2, pi/3 of the three-phase single-stage resonant conversion device are related. Therefore, the voltage gain of the asymmetric LLC resonant cavity of the three-phase single-stage resonant conversion device is similar to that of a traditional LLC resonant cavity, the lower the frequency is, the steeper the voltage gain curve is, and the higher the voltage gain is, the narrower the range of the switching frequency fn and the M-phase duty ratio is. However, when the switching frequency fn is too narrow, as in the curve of the voltage gain m=1.6 in fig. 5 (a), the variation range of the M-phase duty ratio is widened.

Fig. 7 to 9 show experimental waveforms of a three-phase single-stage resonant conversion device, and fig. 7 shows three-phase current i _A 、i _B 、i _C Sinusoidal variation according to control requirement, output voltage u _o Stable, each switch in the switching circuit 1 gets a soft switch; meanwhile, the LLC resonant cavity works in an NP/NP/P mode and a PO/P/PO mode as shown in fig. 8 (a-b); as shown in FIG. 9, the numerical calculation method is carried out at a temperature of 60 DEG to 90 DEG]The internal two control variables (switching frequency fs and duty cycle D _M ) The variation of (2) and the two control variables of the controller output almost coincide.

The numerical calculation method can accurately describe the change condition of each variable in all switching periods in the energy conversion process of the AC/DC converter, obtain the range and change rule of the switching frequency, the M-phase duty ratio and the output voltage which change along with the change of the phase, know the influence of the switching period and the M-phase duty ratio on the voltage gain of the LLC resonant cavity which is asymmetrically input, and the analysis is beneficial to the selection and the parameter design of each device of the three-phase single-stage AC/DC converter.

The three-phase single-stage resonant conversion device controls the transmission energy of the resonant converter by adjusting the switching frequency so as to control the input current and the output voltage, and the converter realizes the soft switching of the high-frequency switch by using the resonant converter, so that the switching loss is reduced, and the higher converter efficiency is obtained. In addition, compared with a two-stage AC/DC converter, the single-stage AC/DC converter does not need a bus capacitor, and only a single-stage structure is needed to complete energy conversion, so that higher power density can be obtained. Under the action of three time-varying input voltages, a three-phase single-stage AC/DC LLC converter utilizes gain characteristics of an LLC resonant cavity to pass through switching frequency and M-phase duty ratio D _M The regulation of the three input currents enables stable control of PFC and output voltage.

As shown in fig. 1, the control unit 20 includes a voltage control loop 26 and a current control loop 25, the voltage control loop 26 samples the output voltage Vo and performs feedback adjustment with the reference voltage Vref to generate a three-phase current reference peak value Imax, the three-phase current reference peak value Imax is input to the current control loop 25, and the three-phase current reference peak value Imax is multiplied by the phase information sin (θ _L ) Sum sin (theta) _M ) To obtain the reference current I of the two phases _Mref And I _Lref . The current control loop 25 controls the L-phase and M-phase currents i _L And i _M Tracking L-phase and M-phase reference currents I _Lref And I _Mref To achieve a three-phase supply current PFC.

Referring to fig. 10 again, the third calculation module 23 calculates the control variable f _s And D _M The following modulation is achieved:

due to u in three-phase three-wire ACDC converter _L ＝u _M +u _S ，i _L ＝i _M +i _S Therefore, the input power of L phase is dominant, L phase is selected to adjust the switching frequency to control the power transmission level, half switching period is occupied fixedly, and L phase current i _L With L-phase reference current I _Lref Generating switching frequency after feedback adjustmentf _s While the M-phase and S-phase are dedicated to distributing the input current by adjusting the duty cycle D _M PFC is realized, sharing the remaining half period. M-phase current i _M With M-phase reference current I _Mref Generating the duty ratio D after feedback adjustment _M 。

When Vo<At Vref, the error Δvo (Δvo=vref-Vo) is positive. With the help of the PI regulator, the peak current I is input _Max Will increase, its corresponding L/M phase reference current I _Lref And I _Mref And will be higher. At this time, |i _L |<I _Lref Error Δi _L (Δi _L ＝|i _L |-I _Lref ) Less than 0. With the help of the L-phase current loop PI regulator, f _s Will be adjusted lower. I i _M |<I _Mref Error Δi _M (Δi _M ＝I _Mref -|i _M I) is greater than 0, D with the help of an M-phase current loop PI regulator _M Increasing.

When Vo is greater than Vref, the error Δvo (Δvo=vref-Vo) is negative. With the help of the PI regulator, the peak current I is input _Max Will decrease, its corresponding L/M phase reference current I _Lref And I _Mref And will be smaller. At this time, |i _L |＞I _Lref Error Δi _L (Δi _L ＝|i _L |-I _Lref ) Greater than 0. With the help of the L-phase current loop PI regulator, f _s Will be adjusted higher. I i _M |<I _Mref Error Δi _M (Δi _M ＝I _Mref -|i _M I) is less than 0, D with the help of an M-phase current loop PI regulator _M And (3) reducing.

The first calculation module 21 is according to the following formula:

the L/M/S phase is determined and in connection with fig. 11, the control unit 20 then determines the inverter operating area within one power frequency period in accordance with the L/M/S phase voltage definition. Referring to FIG. 11, the working area is divided into 12 pieces, no.1-No.12, respectively. The fourth calculation module determines the driving logic of the 6 switches based on the operating areas No.1-No.12 determined by the first calculation module in combination with fig. 10. Referring to fig. 12 (a) to 12 (b), fig. 12 (a) to 12 (b) show switching timing diagrams of the region No. 3. Wherein 12 (a) is a switching timing chart and a key signal waveform chart with a switching frequency fs greater than the resonance frequency fr of the resonance circuit, and 12 (b) is a switching timing chart and a key signal waveform chart with a switching frequency fs less than the resonance frequency fr of the resonance circuit.

It should be understood that the foregoing detailed description of the present invention is provided for illustration only and is not limited to the technical solutions described in the embodiments of the present invention, and those skilled in the art should understand that the present invention may be modified or substituted for the same technical effects; as long as the use requirement is met, the invention is within the protection scope of the invention.

Claims (10)

1. A three-phase single-stage resonance conversion device comprises a power unit and a control unit, wherein the control unit collects input voltage, input current and output voltage signals of the power unit and provides driving signals for a switch in the power unit,

the power unit comprises a switch circuit, a resonant circuit and a rectifying and filtering circuit, wherein the switch circuit, the resonant circuit and the rectifying and filtering circuit are sequentially connected in series, the input end of the switch circuit is connected with a three-phase power supply, the phase with the largest absolute value of voltage is set to be L phase, the phase with the smallest absolute value of voltage is set to be S phase, the other phase is set to be M phase, the output end of the rectifying and filtering circuit is connected with a load,

the control unit comprises a voltage control loop and a current control loop, wherein the voltage control loop samples the output voltage and generates a three-phase current reference peak value after feedback adjustment with a reference voltage, the three-phase current reference peak value is input to the current control loop, the three-phase current reference peak value is multiplied by phase information of L phases and M phases to obtain L-phase reference currents and M-phase reference currents of the two phases, and the current control loop controls the L-phase and M-phase current to track the L-phase and M-phase reference currents;

the three-phase single-stage resonant conversion device executes a numerical calculation method of the three-phase single-stage resonant conversion device, and the method comprises the following steps:

step S10: dividing a static period into N equally divided phase intervals, and selecting an LLC switching period with asymmetric input voltage in each phase interval for numerical calculation;

step S20: numerical calculation is carried out on the 1 st LLC switching period and the N th LLC switching period in the static period, and the working time of three input voltages in the 1 st and N th LLC switching periods is respectively solved、/>、/>And resonant currentExciting current->Resonance capacitor voltage->Exciting inductance voltage->And respectively determining the operation modes of the 1 st LLC switching period and the N-th LLC switching period;

step S30: according to the operation modes and the variable values of the 1 st and the N th LLC switching periods calculated in the step S20, sequentially carrying out numerical calculation on the 2 nd to the N-1 st LLC switching periods, and respectively solving the working time of the three input voltages in each LLC switching period、/>、/>And resonance current->Exciting current->Resonant capacitor voltageExciting inductance voltage->And determining the operation mode of each LLC switching period;

step S40: obtaining switching frequencies at different voltage output gains and input power levelsAnd the duty cycle of the M phase->Is a law of variation of (c).

2. The three-phase single-stage resonant conversion device according to claim 1, wherein the switching circuit comprises a first bridge arm, a second bridge arm and a third bridge arm, the first bridge arm, the second bridge arm and the third bridge arm are connected in parallel, the first bridge arm comprises a first switch and a first capacitor, the first switch is a bidirectional switch, the second bridge arm comprises a second switch and a second capacitor, the second switch is connected in series with the second capacitor, the second switch is a bidirectional switch, the third bridge arm comprises a third switch and a third capacitor, the third switch is connected in series with the third capacitor, the third switch is a bidirectional switch, one ends of the first capacitor, the second capacitor and the third capacitor are connected in parallel, and the other ends of the first capacitor, the second capacitor and the third capacitor are connected with three-phase alternating current through a first inductor, a second inductor and a third inductor respectively.

3. A three-phase single-stage resonant conversion device according to claim 2, wherein the resonant circuit comprises a fourth inductor, a transformer and a fourth capacitor connected in series in sequence, a primary winding of the transformer is connected in series with the fourth inductor and the fourth capacitor, and a secondary winding of the transformer is connected in parallel with the rectifying and filtering circuit.

4. The three-phase single-stage resonant converter of claim 1, wherein said L-phase reference current and L-phase sampling current are feedback-adjusted to generate a switching frequencyf _s 。

5. The three-phase single-stage resonant converter of claim 4, wherein said M-phase reference current and M-phase sampling current are feedback-adjusted to generate a duty cycleD _M 。

6. A three-phase single-stage resonant conversion device according to claim 5, wherein the L-phase is fixed to occupy half of the switching period, and the M-phase and S-phase share the remaining half period.

7. The numerical calculation method of the three-phase single-stage resonant conversion device is characterized by comprising the following steps of:

step S10: dividing a static period into N equally divided phase intervals, and selecting an LLC switching period with asymmetric input voltage in each phase interval for numerical calculation;

step S20: numerical calculation is carried out on the 1 st LLC switching period and the N th LLC switching period in the static period, and the working time of three input voltages in the 1 st and N th LLC switching periods is respectively solved、/>、/>And resonant currentExciting current->Resonance capacitor voltage->Exciting inductance voltage->And respectively determining the operation modes of the 1 st LLC switching period and the N-th LLC switching period;

step S30: according to the operation modes and the variable values of the 1 st and the N th LLC switching periods calculated in the step S20, sequentially carrying out numerical calculation on the 2 nd to the N-1 st LLC switching periods, and respectively solving the working time of the three input voltages in each LLC switching period、/>、/>And resonance current->Exciting current->Resonant capacitor voltageExciting inductance voltage->And determining the operation mode of each LLC switching period;

step S40: obtaining switching frequencies at different voltage output gains and input power levelsAnd the duty cycle of the M phase->Is a law of variation of (c).

8. The method for calculating a value of a three-phase single-stage resonant converter according to claim 7, wherein said step S10 further comprises defining three asymmetrical input voltages of the three-phase single-stage AC/DC resonant converter as、/>、/>Said->Voltage direction and->、/>Is opposite in voltage direction,/->And is also provided withIn any one switching period +.>。

9. The method of claim 8, wherein the LLC switching cycle is operated in an operating mode comprising three basic states of P-state, N-state and O-state sequentially,

the P state is: output voltage of LLC resonant cavityAnd input voltage->The directions are the same, resonant inductance L _r And a resonance capacitor C _r Keep resonance, excitation inductance L _m Is voltage->Clamping;

the N state is: output voltage of LLC resonant cavityAnd input voltage->Opposite direction, resonant inductance L _r And a resonance capacitor C _r Keep resonance, excitation inductance L _m The clamping voltage direction of (2) is changed;

the O state is: LLC resonant cavity and transformer secondary side direct current side disconnection, excitation inductance L _m Added to the resonant inductance L _r And a resonance capacitor C _r During resonance of (a).

10. The method according to claim 9, wherein the step S20 further comprises sorting out the variable time domain expressions and the variable equivalent equations in each LLC switching cycle requiring numerical calculation, the variable equivalent equations following the following rules:

1) Resonant current in all LLC switching cyclesAnd capacitor voltage->The continuous change is kept all the time, and the mutation at any time is avoided;

2) Energy generated by each phase of input current and input energy conservation of a corresponding LLC resonant cavity, and meanwhile, the input energy and output energy conservation of the resonant cavity;

3）wherein->For input voltage +.>Duty cycle of>For input voltage +.>Duty cycle of>For input voltage +.>Duty cycle of (2);

4) The sum of the three input currents provided by the three-phase power supply to the AC/DC resonant converter is 0 during one LLC switching cycle.