CN114156890B - LCL type grid-connected inverter current control method with double inductance current changes - Google Patents
LCL type grid-connected inverter current control method with double inductance current changes Download PDFInfo
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J3/00—Circuit arrangements for ac mains or ac distribution networks
- H02J3/01—Arrangements for reducing harmonics or ripples
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J3/00—Circuit arrangements for ac mains or ac distribution networks
- H02J3/002—Flicker reduction, e.g. compensation of flicker introduced by non-linear load
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J3/00—Circuit arrangements for ac mains or ac distribution networks
- H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/12—Arrangements for reducing harmonics from ac input or output
- H02M1/126—Arrangements for reducing harmonics from ac input or output using passive filters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/483—Converters with outputs that each can have more than two voltages levels
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/493—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode the static converters being arranged for operation in parallel
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- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
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- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
- H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
- H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
- H02M7/53873—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with digital control
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Abstract
The invention discloses a current control method of a double-inductance current-variable LCL type grid-connected inverter, which realizes simultaneous grid-connected current tracking and resonance suppression by respectively decomposing and summing inductance current at the inverter side and inductance current variable at the power grid side. The method is implemented according to the following steps: step 1: the column writing single-phase equivalent circuit expression and the definition of three duty ratios of working states. Step 2: a double decomposition process; the two inductors of the LCL filter are subjected to double decomposition, and the positive half period T of the modulation wave is carried out s In step (2), a double decomposition formula of the inductor current at the inverter side and the grid-connected current is obtained; step 3: a summation process; the total current change of the two inductors in the positive half period of the modulation wave is obtained by summarizing the above formula. Step 4: an x-phase duty cycle expression is obtained. The D-D-sigma current control method provided by the invention is more suitable for LCL grid-connected inverters, and reduces the difficulty of resonance suppression while having the advantages of the D-sigma current control method.
Description
Technical Field
The invention belongs to the technical field of power electronics, and particularly relates to a current control method of a double-inductance current-variable LCL grid-connected inverter.
Background
Carbon emissions are one of the main causes of global warming, and in the carbon emissions of China, the electric power and heating sector occupies more than half of the fossil fuel combustion in the production link. To cope with climate change and energy consumption, the country proposes that "carbon dioxide emissions strive to peak before 2030, striving to achieve carbon-neutralised carbon before 2060". To achieve the "3060" carbon goal, an important measure is to develop new energy power generation technologies so as to compensate for the energy supply when gradually exiting the coal power generation project. In the field of new energy power generation, photovoltaic power generation, wind power generation and the like have been rapidly developed in recent years.
The grid-connected inverter is an important device for realizing the electric energy remittance into an alternating current grid in a power generation system, and aims to realize high efficiency, low grid-connected current harmonic wave and high reliability. Compared with a L, LC type filter, the LCL type filter has better high-frequency harmonic suppression capability, so that the LCL type filter is a more common grid-connected filter. However, the LCL type filter is a third-order system, and a resonance peak exists in the system, which may cause a large resonance current, thereby causing instability of the system. The D-sigma current control method is a grid-connected current control method based on an inverter model, and a grid-connected current duty ratio expression is directly deduced by considering inductance current change. The method avoids complex coordinate transformation, can directly control current under a three-phase abc coordinate system, and is more suitable for abnormal working conditions of power grid voltage. The application of the traditional D-sigma current control method in the LC-type grid-connected inverter is mature, but when the traditional D-sigma current control method is applied to the LCL-type grid-connected inverter, an additional damping method is needed to restrain resonance peaks, such as a resonance restraining method based on capacitive current feedback.
Disclosure of Invention
In order to solve the problems, the invention aims to provide a D-D-sigma current control method which is suitable for an LCL grid-connected inverter and simultaneously takes grid-connected current tracking and resonance suppression into consideration.
In order to achieve the above purpose, the present invention provides the following technical solutions: the LCL type grid-connected inverter current control method with double inductance current changes is implemented according to the following steps:
step 1: the method comprises the steps of writing single-phase equivalent circuit expressions and defining three duty ratios of working states;
firstly, ignoring the voltage drop and dead time influence of a power device in a positive half period of a modulation wave, and obtaining a mathematical expression of an equal circuit of x under the neutral point potential balance according to kirchhoff voltage law, wherein the mathematical expression is as follows:
in the formula (1), U dc Is input DC power voltage, U PCCx Representing the voltage at the point of common coupling PCC; inverter side inductance L xi Capacitance C x And a grid side inductance L xg An LCL filter is formed; power grid side inductance L xg Excluding grid impedance; wherein n represents the sampling time, t pon Representing U dc An operation time of/2; i.e xi I is the inverter side current xg Is the grid side current;
similarly, in the negative half period of the modulation wave, ignoring the voltage drop and dead time influence of the power device, and obtaining the mathematical expression of the equal circuit of x under the neutral point potential balance according to kirchhoff voltage law is as follows:
let the duty ratio of the grid-connected current and resonance suppression be d x ，d x Not less than 0, when the modulation wave is in positive half period, the output voltage of the inverter comprises +U dc Two voltages of/2 and 0, therefore, +U dc 2, P state, 0 is O state and-U dc And/2 is in an N state, and the duty ratios corresponding to the three output voltages are set as follows:
in the formula (3), d xP 、d xO 、d xN The x-phase duty ratios corresponding to the P state, the O state and the N state respectively;
when the modulated wave is in the negative half-cycle, the inverter output voltage includes-U dc Two voltages of/2 and 0; thus, will +U dc 2, P-state; 0, i.e., O state; -U dc And/2, namely N state, the duty ratios corresponding to the three output voltages are set as follows:
step 2: a double decomposition process;
the two inductors of the LCL filter are subjected to double decomposition, and the positive half period T of the modulation wave is carried out s In/2, the double decomposition formula of the inductor current and the grid-connected current at the inverter side is obtained as follows:
wherein T is s For switching period Δi xiP+ Representing the corresponding current variation of the x phase P working state on the side inductor of the inverter under the positive half period of the modulation wave; Δi xiO+ Representing the corresponding current variation of the x phase of the inductor on the side of the inverter under the positive half period of the modulation wave in the O working state; Δi xiN+ Representing the corresponding current variation of the x phase on the side inductor of the inverter under the positive half period of the modulation wave; Δi xgP+ Representing the corresponding current variation under the P working state of the x phase on the power grid side inductor under the positive half period of the modulation wave; Δi xgO+ Representing the corresponding current variation of the x phase on the power grid side inductor under the positive half period of the modulation wave in the O working state; Δi xgN+ Representing the corresponding current variation of the x phase on the power grid side inductor under the positive half period of the modulation wave;
in the negative half period T of the modulation wave s During period/2, the inductor current and the grid-connected current of the inverter side are deduced：
Wherein Δi xiP- Representing the corresponding current variation of the x phase P working state on the side inductor of the inverter under the negative half period of the modulation wave; Δi xiO- Representing the corresponding current variation of the x phase of the inductor at the side of the inverter under the negative half period of the modulation wave in the O working state; Δi xiN- Representing the corresponding current variation of the x phase on the side inductor of the inverter under the negative half period of the modulation wave; Δi xgP- Representing the corresponding current variation of the x phase P working state of the power grid side inductor under the negative half period of the modulation wave; Δi xgO- Representing the corresponding current variation of the x phase on the power grid side inductor under the negative half period of the modulation wave under the O working state; Δi xgN- Representing the corresponding current variation of the x phase on the power grid side inductor under the negative half period of the modulation wave;
step 3: a summation process;
the total current change of the two inductors in the positive half period of the modulation wave is obtained by summarizing the above formula and is expressed as:
the deduction is obtained by combining the formula (3) and the formula (7):
wherein Δi xi+ Representing the current variation of the x phase on the side inductor of the inverter under the positive half period of the modulation wave; Δi xg+ Representing the current change of the x phase on the inductance of the power grid side under the positive half period of the modulation wave;
likewise, by summarizing the above formula, the total current change of the two inductors in the negative half period of the modulated wave can be expressed as:
combining formula (4) and formula (9) yields formula (10):
wherein Δi xi- Representing the current variation of the x phase on the side inductor of the inverter under the negative half period of the modulation wave; Δi xg- Representing the current change of the x phase on the inductance of the power grid side under the negative half period of the modulation wave;
step 4: obtaining an x-phase duty ratio expression;
the complete expression of the current change is obtained by comprehensively modulating the two inductance current changes of positive and negative half periods of the wave:
from equation (11), the duty cycle expression of the grid-connected current in one switching period can be deduced as follows:
the amount of change in inductor current during a switching cycle should be:
i x ＝I xref (n+1)-I xref (n) (13)
wherein I is xref (n+1) is a current reference value at time n+1, I xref (n) is a current reference value at time n;
but grid-connected current can generate errors in the tracking process:
i e ＝I xref (n)-i x (n) (14)
wherein i is x (n) is the actual sampling current at time n; adding the error to the desired amount of current change yields the following expression:
i e,x ＝i x +i e ＝I xref (n+1)-i x (n) (15)
applying the above analysis to the inverter side inductor current and grid-connected current, i.e. xref (n+1)-i x (n) to replace two inductor current variations:
wherein I is xgref (n+1) and I xiref (n+1) is a reference value of the grid-connected current and the inverter-side inductor current at time n+1, respectively; i xg (n) and i xi (n) is the actual sampling value of the grid-connected current and the inverter side inductance current at the moment n respectively;
the reference value of the grid-connected current replaces the reference value of the inverter-side current, and the formula (12) is converted into:
the final duty cycle expression can be obtained by equation (17) as follows:
d x ＝K p1 [I xgref (n+1)-i xi (n)]+K p2 [I xgref (n+1)-i xg (n)]+K p3 U PCCx (18)
wherein the proportionality coefficient K p1 ＝2L xi /(U dc T s ) Scaling factor K p2 ＝2L xg /(U dc T s ) Scaling factor K p3 ＝2/U dc The method comprises the steps of carrying out a first treatment on the surface of the In the formula (18), K p2 The grid-connected current gain is represented, and the tracking precision of the current is affected; k (K) p1 The current gain of the inductor at the inverter side is expressed, and the resonance suppression effect is affected; k (K) p3 The feedforward gain of grid-connected voltage under the PCC condition is shown, and the dynamic response speed of current is influenced.
Compared with the prior art, the invention has the following beneficial effects:
according to the invention, the two inductors of the LCL filter are respectively decomposed and summed in the positive half period and the negative half period of the modulation wave, so that the duty ratio expression which simultaneously takes account of grid-connected current tracking and resonance suppression is obtained. Compared with the traditional D-sigma current control method, the D-D-sigma current control method provided by the invention is more suitable for LCL type grid-connected inverters, and reduces the difficulty of resonance suppression while having the advantages of the D-sigma current control method.
Drawings
FIG. 1 is a schematic diagram of a T-type three-level grid-connected inverter with an LCL filter of the present invention;
fig. 2 is a block diagram of a digital control implementation of the D- Σ of the present invention.
Detailed Description
The following description of the embodiments of the present invention will be made clearly and completely with reference to the accompanying drawings, in which it is apparent that the embodiments described are only some embodiments of the present invention, but not all embodiments. All other embodiments, which can be made by those skilled in the art based on the embodiments of the invention without making any inventive effort, are intended to be within the scope of the invention.
Referring to fig. 1-2, the present invention provides a technical solution: fig. 1 is a schematic diagram of a T-type three-level grid-connected inverter with an LCL filter. DC input power supply U dc With two series-connected DC filter capacitors C 1 And C 2 Parallel connection is carried out, capacitor C 1 Upper voltage of U c1 The positive bus end is marked as P and the capacitor C 2 Upper voltage of U c2 The negative bus end is marked as N, and the capacitor C 1 And capacitor C 2 The connection point O of (2) is marked as the midpoint of the direct current side; each phase bridge circuit of the T-shaped three-level inverter consists of four IGBTs with anti-parallel diodes, and the four IGBTs are respectively marked as S x1 ，S x2 ，S x3 ，S x4 ，x＝a,b,c，S x2 Emitter and S of (2) x3 Is reverse connected with the emitter of S x2 The collector electrode of (a) is connected to the midpoint O of the direct current side; s is S x1 Emitter and S of (2) x4 Is connected in series with the collector of S x1 The collector of (a) is connected with the DC side P, S x4 Is of (1)The emitter is connected with the direct current side N; s is S x3 Collector and S of (2) x1 The emitters of the three-phase alternating current voltage u are respectively output after being connected ao ，u bo ，u co ；u ao ，u bo ，u co Respectively connected with the inversion side filter inductance L ai ，L bi And L ci Is L at one end of ai ，L bi And L ci The other ends of the filter capacitors are respectively connected with a filter capacitor C a ，C b And C c One end of (2) and a net side inductance L ag ，L bg And L cg Is a member of the group; filter capacitor C a ，C b And C c The other ends of the two parts are connected together and suspended in the air; l (L) ag ，L bg And L cg The other end of the power grid is respectively connected with a, b and c phase power grid voltage U PCCa ，U PCCb ，U PCCc 。
The method is implemented according to the following four steps:
the LCL type grid-connected inverter current control method with double inductance current changes is implemented according to the following steps:
step 1: the method comprises the steps of writing single-phase equivalent circuit expressions and defining three duty ratios of working states;
firstly, ignoring the voltage drop and dead time influence of a power device in a positive half period of a modulation wave, and obtaining a mathematical expression of an equal circuit of x under the neutral point potential balance according to kirchhoff voltage law, wherein the mathematical expression is as follows:
in the formula (1), U dc Is input DC power voltage, U PCCx Representing the voltage at the point of common coupling PCC; inverter side inductance L xi Capacitance C x And a grid side inductance L xg An LCL filter is formed; power grid side inductance L xg Excluding grid impedance; wherein n represents the sampling time, t pon Representing U dc An operation time of/2; i.e xi I is the inverter side current xg Is the grid side current;
similarly, in the negative half period of the modulation wave, ignoring the voltage drop and dead time influence of the power device, and obtaining the mathematical expression of the equal circuit of x under the neutral point potential balance according to kirchhoff voltage law is as follows:
let the duty ratio of the grid-connected current and resonance suppression be d x ，d x Not less than 0, when the modulation wave is in positive half period, the output voltage of the inverter comprises +U dc Two voltages of/2 and 0, therefore, +U dc 2, P state, 0 is O state and-U dc And/2 is in an N state, and the duty ratios corresponding to the three output voltages are set as follows:
in the formula (3), d xP 、d xO 、d xN The x-phase duty ratios corresponding to the P state, the O state and the N state respectively;
when the modulated wave is in the negative half-cycle, the inverter output voltage includes-U dc Two voltages of/2 and 0; thus, will +U dc 2, P-state; 0, i.e., O state; -U dc And/2, namely N state, the duty ratios corresponding to the three output voltages are set as follows:
step 2: a double decomposition process;
the two inductors of the LCL filter are subjected to double decomposition, and the positive half period T of the modulation wave is carried out s In/2, the double decomposition formula of the inductor current and the grid-connected current at the inverter side is obtained as follows:
wherein the method comprises the steps of，T s For switching period Δi xiP+ Representing the corresponding current variation of the x phase P working state on the side inductor of the inverter under the positive half period of the modulation wave; Δi xiO+ Representing the corresponding current variation of the x phase of the inductor on the side of the inverter under the positive half period of the modulation wave in the O working state; Δi xiN+ Representing the corresponding current variation of the x phase on the side inductor of the inverter under the positive half period of the modulation wave; Δi xgP+ Representing the corresponding current variation under the P working state of the x phase on the power grid side inductor under the positive half period of the modulation wave; Δi xgO+ Representing the corresponding current variation of the x phase on the power grid side inductor under the positive half period of the modulation wave in the O working state; Δi xgN+ Representing the corresponding current variation of the x phase on the power grid side inductor under the positive half period of the modulation wave;
in the negative half period T of the modulation wave s During/2, the inverter side inductor current and the grid-tie current are derived:
wherein Δi xiP- Representing the corresponding current variation of the x phase P working state on the side inductor of the inverter under the negative half period of the modulation wave; Δi xiO- Representing the corresponding current variation of the x phase of the inductor at the side of the inverter under the negative half period of the modulation wave in the O working state; Δi xiN- Representing the corresponding current variation of the x phase on the side inductor of the inverter under the negative half period of the modulation wave; Δi xgP- Representing the corresponding current variation of the x phase P working state of the power grid side inductor under the negative half period of the modulation wave; Δi xgO- Representing the corresponding current variation of the x phase on the power grid side inductor under the negative half period of the modulation wave under the O working state; Δi xgN- Representing the corresponding current variation of the x phase on the power grid side inductor under the negative half period of the modulation wave;
step 3: a summation process;
the total current change of the two inductors in the positive half period of the modulation wave is obtained by summarizing the above formula and is expressed as:
the deduction is obtained by combining the formula (3) and the formula (7):
wherein Δi xi+ Representing the current variation of the x phase on the side inductor of the inverter under the positive half period of the modulation wave; Δi xg+ Representing the current change of the x phase on the inductance of the power grid side under the positive half period of the modulation wave;
likewise, by summarizing the above formula, the total current change of the two inductors in the negative half period of the modulated wave can be expressed as:
combining formula (4) and formula (9) yields formula (10):
wherein Δi xi- Representing the current variation of the x phase on the side inductor of the inverter under the negative half period of the modulation wave; Δi xg- Representing the current change of the x phase on the inductance of the power grid side under the negative half period of the modulation wave;
step 4: obtaining an x-phase duty ratio expression;
the complete expression of the current change is obtained by comprehensively modulating the two inductance current changes of positive and negative half periods of the wave:
from equation (11), the duty cycle expression of the grid-connected current in one switching period can be deduced as follows:
the amount of change in inductor current during a switching cycle should be:
i x ＝I xref (n+1)-I xref (n) (13)
wherein I is xref (n+1) is a current reference value at time n+1, I xref (n) is a current reference value at time n;
but grid-connected current can generate errors in the tracking process:
i e ＝I xref (n)-i x (n) (14)
wherein i is x (n) is the actual sampling current at time n; adding the error to the desired amount of current change yields the following expression:
i e,x ＝i x +i e ＝I xref (n+1)-i x (n) (15)
applying the above analysis to the inverter side inductor current and grid-connected current, i.e. xref (n+1)-i x (n) to replace two inductor current variations:
wherein I is xgref (n+1) and I xiref (n+1) is a reference value of the grid-connected current and the inverter-side inductor current at time n+1, respectively; i xg (n) and i xi (n) is the actual sampling value of the grid-connected current and the inverter side inductance current at the moment n respectively;
the reference value of the grid-connected current replaces the reference value of the inverter-side current, and the formula (12) is converted into:
the final duty cycle expression can be obtained by equation (17) as follows:
d x ＝K p1 [I xgref (n+1)-i xi (n)]+K p2 [I xgref (n+1)-i xg (n)]+K p3 U PCCx (18)
wherein the proportionality coefficient K p1 ＝2L xi /(U dc T s ) Scaling factor K p2 ＝2L xg /(U dc T s ) Scaling factor K p3 ＝2/U dc The method comprises the steps of carrying out a first treatment on the surface of the In the formula (18), K p2 The grid-connected current gain is represented, and the tracking precision of the current is affected; k (K) p1 The current gain of the inductor at the inverter side is expressed, and the resonance suppression effect is affected; k (K) p3 The feedforward gain of grid-connected voltage under the PCC condition is shown, and the dynamic response speed of current is influenced.
From the derivation of the present invention, D- Σ digital control is a three-phase current independent control that does not require coordinate transformation. Thus, when the three-phase grid is unbalanced, balanced three-phase grid-connected currents can still be obtained using the D- Σ digital control algorithm. By adopting the D-D-sigma digital control strategy, the grid-connected current can accurately track the reference value and inhibit resonance phenomenon.
Fig. 2 is a block diagram of a D- Σ digital control implementation of a T-type three-level grid-connected inverter with an LCL filter. The implementation process of each control loop in fig. 2 will be specifically described, firstly, sampling the voltage of a three-phase PCC point to perform phase-locked loop processing to obtain a power grid phase θ, and constructing a three-phase unit modulation wave of a grid-connected current reference value through θ, so that the three-phase unit modulation wave is multiplied by the amplitude of a grid-connected current given value to obtain the given value of the grid-connected current; multiplying the error between the given value of the grid-connected current and the actual current feedback value at the grid side by the proportionality coefficient K p2 Obtaining a control loop gain of the grid-connected current; multiplying the error between the given value of the grid-connected current and the actual current feedback value of the inductor at the inversion side by a proportionality coefficient K p1 Obtaining a control loop gain of resonance suppression; multiplying the grid voltage of the PCC point by a scaling factor K p3 Obtaining the control loop gain of the power grid voltage feedforward, adding the three loop gains to obtain the final modulation wave of x phases, comparing the x-phase modulation wave with a triangular carrier wave to generate a sine pulse width modulation signal,the grid driving circuit is used for controlling the switching action of the inverter, so that sine current with unit power factor is injected into the power grid.
Although embodiments of the present invention have been shown and described, it will be understood by those skilled in the art that various changes, modifications, substitutions and alterations can be made therein without departing from the principles and spirit of the invention, the scope of which is defined in the appended claims and their equivalents.
Claims (1)
1. The LCL type grid-connected inverter current control method with double inductance current changes is characterized by comprising the following steps of:
step 1: the method comprises the steps of writing single-phase equivalent circuit expressions and defining three duty ratios of working states;
firstly, ignoring the voltage drop and dead time influence of a power device in a positive half period of a modulation wave, and obtaining a mathematical expression of an equal circuit of x under the neutral point potential balance according to kirchhoff voltage law, wherein the mathematical expression is as follows:
in the formula (1), U dc Is input DC power voltage, U PCCx Representing the voltage at the point of common coupling PCC; inverter side inductance L xi Capacitance C x And a grid side inductance L xg An LCL filter is formed; power grid side inductance L xg Excluding grid impedance; wherein n represents the sampling time, t pon Representing U dc An operation time of/2; i.e xi I is the inverter side current xg Is the grid side current;
similarly, in the negative half period of the modulation wave, ignoring the voltage drop and dead time influence of the power device, and obtaining the mathematical expression of the equal circuit of x under the neutral point potential balance according to kirchhoff voltage law is as follows:
let the duty ratio of the grid-connected current and resonance suppression be d x ，d x Not less than 0, when the modulation wave is in positive half period, the output voltage of the inverter comprises +U dc Two voltages of/2 and 0, therefore, +U dc 2, P state, 0 is O state and-U dc And/2 is in an N state, and the duty ratios corresponding to the three output voltages are set as follows:
in the formula (3), d xP 、d xO 、d xN The x-phase duty ratios corresponding to the P state, the O state and the N state respectively;
when the modulated wave is in the negative half-cycle, the inverter output voltage includes-U dc Two voltages of/2 and 0; thus, will +U dc 2, P-state; 0, i.e., O state; -U dc And/2, namely N state, the duty ratios corresponding to the three output voltages are set as follows:
step 2: a double decomposition process;
the two inductors of the LCL filter are subjected to double decomposition, and the positive half period T of the modulation wave is carried out s In/2, the double decomposition formula of the inductor current and the grid-connected current at the inverter side is obtained as follows:
wherein T is s For switching period Δi xiP+ Representing the corresponding current variation of the x phase P working state on the side inductor of the inverter under the positive half period of the modulation wave; Δi xiO+ Representing the corresponding current in the O working state of the x phase on the side inductor of the inverter under the positive half period of the modulation waveA variation amount; Δi xiN+ Representing the corresponding current variation of the x phase on the side inductor of the inverter under the positive half period of the modulation wave; Δi xgP+ Representing the corresponding current variation under the P working state of the x phase on the power grid side inductor under the positive half period of the modulation wave; Δi xgO+ Representing the corresponding current variation of the x phase on the power grid side inductor under the positive half period of the modulation wave in the O working state; Δi xgN+ Representing the corresponding current variation of the x phase on the power grid side inductor under the positive half period of the modulation wave;
in the negative half period T of the modulation wave s During/2, the inverter side inductor current and the grid-tie current are derived:
wherein Δi xiP- Representing the corresponding current variation of the x phase P working state on the side inductor of the inverter under the negative half period of the modulation wave; Δi xiO- Representing the corresponding current variation of the x phase of the inductor at the side of the inverter under the negative half period of the modulation wave in the O working state; Δi xiN- Representing the corresponding current variation of the x phase on the side inductor of the inverter under the negative half period of the modulation wave; Δi xgP- Representing the corresponding current variation of the x phase P working state of the power grid side inductor under the negative half period of the modulation wave; Δi xgO- Representing the corresponding current variation of the x phase on the power grid side inductor under the negative half period of the modulation wave under the O working state; Δi xgN- Representing the corresponding current variation of the x phase on the power grid side inductor under the negative half period of the modulation wave;
step 3: a summation process;
the total current change of the two inductors in the positive half period of the modulation wave is obtained by summarizing the above formula and is expressed as:
the deduction is obtained by combining the formula (3) and the formula (7):
wherein Δi xi+ Representing the current variation of the x phase on the side inductor of the inverter under the positive half period of the modulation wave; Δi xg+ Representing the current change of the x phase on the inductance of the power grid side under the positive half period of the modulation wave;
likewise, by summarizing the above formula, the total current change of the two inductors in the negative half period of the modulated wave can be expressed as:
combining formula (4) and formula (9) yields formula (10):
wherein Δi xi- Representing the current variation of the x phase on the side inductor of the inverter under the negative half period of the modulation wave; Δi xg- Representing the current change of the x phase on the inductance of the power grid side under the negative half period of the modulation wave;
step 4: obtaining an x-phase duty ratio expression;
the complete expression of the current change is obtained by comprehensively modulating the two inductance current changes of positive and negative half periods of the wave:
from equation (11), the duty cycle expression of the grid-connected current in one switching period can be deduced as follows:
the amount of change in inductor current during a switching cycle should be:
i x ＝I xref (n+1)-I xref (n) (13)
wherein I is xref (n+1) is a current reference value at time n+1, I xref (n) is a current reference value at time n;
but grid-connected current can generate errors in the tracking process:
i e ＝I xref (n)-i x (n) (14)
wherein i is x (n) is the actual sampling current at time n; adding the error to the desired amount of current change yields the following expression:
i e,x ＝i x +i e ＝I xref (n+1)-i x (n) (15)
applying the above analysis to the inverter side inductor current and grid-connected current, i.e. xref (n+1)-i x (n) to replace two inductor current variations:
wherein I is xgref (n+1) and I xiref (n+1) is a reference value of the grid-connected current and the inverter-side inductor current at time n+1, respectively; i xg (n) and i xi (n) is the actual sampling value of the grid-connected current and the inverter side inductance current at the moment n respectively;
the reference value of the grid-connected current replaces the reference value of the inverter-side current, and the formula (12) is converted into:
the final duty cycle expression can be obtained by equation (17) as follows:
d x ＝K p1 [I xgref (n+1)-i xi (n)]+K p2 [I xgref (n+1)-i xg (n)]+K p3 U PCCx (18)
wherein the proportionality coefficient K p1 ＝2L xi /(U dc T s ) Scaling factor K p2 ＝2L xg /(U dc T s ) Scaling factor K p3 ＝2/U dc The method comprises the steps of carrying out a first treatment on the surface of the In the formula (18), K p2 The grid-connected current gain is represented, and the tracking precision of the current is affected; k (K) p1 The current gain of the inductor at the inverter side is expressed, and the resonance suppression effect is affected; k (K) p3 The feedforward gain of grid-connected voltage under the PCC condition is shown, and the dynamic response speed of current is influenced.
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