CN114142751B - Three-phase CSR proportional integral resonance control method under unbalanced power grid voltage - Google Patents

Three-phase CSR proportional integral resonance control method under unbalanced power grid voltage Download PDF

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CN114142751B
CN114142751B CN202111529927.0A CN202111529927A CN114142751B CN 114142751 B CN114142751 B CN 114142751B CN 202111529927 A CN202111529927 A CN 202111529927A CN 114142751 B CN114142751 B CN 114142751B
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csr
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CN114142751A (en
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郭强
黄勇军
李海啸
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Chongqing University of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/2173—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a biphase or polyphase circuit arrangement
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Supply And Distribution Of Alternating Current (AREA)

Abstract

The invention discloses a three-phase CSR proportional integral resonance control method under unbalanced power grid voltage to obtain direct-current side load voltage u o And a DC side output current i dc The method comprises the steps of carrying out a first treatment on the surface of the Load voltage u on DC side o And a DC side load voltage reference valueAfter subtracting to obtain voltage error signal, obtaining DC reference value component by PI controlAnd then outputs current i with direct current side dc Comparing to obtain a current error signal, and obtaining a feedback term through the PIR controllerSimultaneously introducing an active damping loop control item for inhibiting LC resonance at an alternating current side and a power factor compensation item for ensuring stable unit power factor operation of CSR, and adding the three items into a grid side current i under a d-q synchronous rotation coordinate system d And i q And converted to obtain a modulated signal i α And i β And after passing through the wave trap, the three-phase CSR is controlled by adopting space vector pulse width modulation. The invention has the advantages of simple and efficient control algorithm, small calculated amount, capability of omitting a network side current sensor, effective reduction of system cost and the like.

Description

Three-phase CSR proportional integral resonance control method under unbalanced power grid voltage
Technical Field
The invention relates to the technical field of power grids, in particular to a three-phase CSR proportional-integral resonance control method under unbalanced power grid voltage.
Background
With the application and development of high-power current converting technology, PWM rectifiers are receiving more and more attention from industry and academia. Compared with voltage source rectifiers (voltage source rectifier, VSR), the current source rectifiers (current source rectifier, CSR) have the advantages of high reliability, wide-range voltage reduction output, no need of a pre-charging circuit, easiness in parallel operation and the like when in direct current short circuit, and are increasingly applied to the fields of data center power supply systems, multi-electric aircraft power distribution systems, battery pack charging systems, renewable new energy systems and the like.
At present, the research of CSR is mainly focused on the voltage balance condition of a three-phase power grid. However, in an actual system, the working condition of unbalanced power grid voltage is common, and double low-frequency pulsation and third harmonic wave of current at the grid side appear at the direct current side under the working condition of unbalanced power grid voltage can deteriorate the CSR performance, so that researchers at home and abroad sequentially put forward some control strategies.
For example, the specific harmonic order of the power grid current suppression item is determined in the cost function provided by model predictive control, so that the use of a sensor is reduced, the CSR can still keep the power grid current highly sinusoidal under the power grid voltage unbalance working condition, but the control algorithm is complex, and the calculated amount is large.
Disclosure of Invention
Aiming at the defects in the prior art, the invention aims to solve the technical problems that: how to provide a control algorithm is simple and efficient, the calculated amount is small, a network side current sensor can be omitted, and the three-phase CSR proportional-integral resonance control method under unbalanced power grid voltage can effectively reduce the system cost.
In order to solve the technical problems, the invention adopts the following technical scheme:
a three-phase CSR proportional-integral resonance control method under unbalanced power grid voltage is characterized by comprising the following steps:
s1, obtaining direct-current side load voltage u o And a DC side output current i dc ；
S2, voltage outer loop control: load voltage u on DC side o And a DC side load voltage reference valueSubtracting to obtain a voltage error signal, and obtaining a direct current reference value component by PI control of the voltage error signal>
S3, current inner loop control: to direct current reference value componentAnd direct current side output current i dc Comparing to obtain a current error signal, obtaining +.>Will->Adding the net side current i under d-q synchronous rotation coordinate system as feedback item d And i q ；
S4, will i d And i q Obtaining a modulation signal i under two-phase stationary coordinates through coordinate conversion α And i β And the space vector pulse width modulation is adopted to control the current source type PWM rectifier.
As an optimization, before the step S4, the net side current i in the d-q synchronous rotation coordinate system d And i q Adding power factor compensation termAnd->
Wherein: c (C) ac Is an alternating current side filter capacitor, u q And u d The capacitor voltage is filtered for the alternating current side under the d-q synchronous rotation coordinate system.
As a further optimization, the net side current i in the d-q synchronous rotation coordinate system is before the step S4 d And i q Adding active damping loop control itemAnd->Obtaining the net side current i under d-q synchronous rotation coordinate system d And i q The method comprises the following steps of:
wherein,and->AC side capacitor voltage u under d-q synchronous rotation coordinate q And u d Filtering by high-pass filter and then the source damping feedback gain is multiplied to obtain:
wherein: k (K) Cv Is an active damping feedback gain;
the expression of the high-pass filter is:
wherein: omega h Is a high pass filter cut-off frequency.
As an optimization, the PIR controller is:
wherein: k (k) p 、k i 、k r Respectively proportional, integral, resonant gain coefficients.
In the step S3, the transfer function of the current inner loop control is:
wherein: a, a 4 ＝K m k p R L C dc ；a 3 ＝K m k p +K m (k i +k r )R L C dc ；a 2 ＝K m (k i +k r )+K m 4ω 2 k p R L C dc ；a 1 ＝K m (4ω 2 k p +4ω 2 k i R L C dc )；a 0 ＝K m 4ω 2 k i ；b 5 ＝R L L dc C dc ；b 4 ＝L dc ；b 3 ＝4ω 2 R L L dc C dc +R L ；b 2 ＝4ω 2 L dc ；b 1 ＝4ω 2 R L ；b 0 ＝0；
Wherein K is m For the system voltage gain, R L Is a load resistor, C dc Is a direct-current side capacitor L dc Is a direct current side inductance.
Further, in the step S4, a wave trap is used to eliminate the modulated signal i α And i β The trap is:
wherein: k (K) 1 Is the controller coefficient.
Further, in the step S4, the space vector pulse width modulation adopts a twelve-sector modulation strategy, and the vector sequence of each sector is a bilateral symmetry current vector sequence.
In conclusion, the method has the advantages of simple and efficient control algorithm, small calculated amount, capability of omitting a network side current sensor, effective reduction of system cost and the like.
Drawings
Fig. 1 is a topology diagram of a current source PWM rectifier.
Fig. 2 is a block diagram of a three-phase CSR control strategy.
Fig. 3 is a single-phase (a-phase) vector diagram of the CSR input side in non-unity power factor operation.
Fig. 4 is a single-phase (a-phase) vector diagram of the CSR input side at unity power factor operation.
FIG. 5 is a schematic diagram of a capacitive voltage feedback active damping strategy.
Fig. 6 is a diagram of an ac-side LC filter Bode.
Fig. 7 is a block diagram of a CSR system control loop.
Fig. 8 is a current inner loop control block diagram.
FIG. 9 is a plot of the open loop transfer function Bode of current at different kr.
Fig. 10 is a graph of a current open-closed loop transfer function Bode.
Fig. 11 is a 12-sector division diagram.
Fig. 12 is a Bode diagram of a trap designed according to the present invention.
Fig. 13 is a schematic diagram of a three-phase unbalanced grid voltage.
Fig. 14 shows a-phase grid voltage, grid side current and dc side load voltage in a conventional double closed loop simulation.
Fig. 15 is a graph of a harmonic analysis of the network side current under a conventional double closed loop simulation.
Fig. 16 shows a-phase grid voltage, grid side current and dc side load voltage under simulation of the inventive scheme.
Fig. 17 is a graph of harmonic analysis of network side current under simulation of the inventive scheme.
Fig. 18 is a graph of the active, reactive and power factors of the present invention under simulation.
FIG. 19 is a simulation result of the dynamic response of the method of the present invention.
FIG. 20 is a model-based code generation control block diagram.
Fig. 21 is a three-phase unbalanced grid voltage.
Fig. 22 shows a-phase grid voltage, grid-side current and dc-side load voltage in CSR experiments with conventional dual closed-loop control.
Fig. 23 is a graph of harmonic analysis of network side current under CSR experiments with conventional dual closed loop control.
FIGS. 24 to 26 are graphs showing the results of CSR experiments performed by the method of the present invention.
Fig. 27 is a graph of harmonic analysis of network side current when u=100deg.V.
FIG. 28 is a schematic of the results of dynamic response experiments.
Detailed Description
The present invention will be described in further detail with reference to examples.
In the embodiment, a mathematical model of the CSR in a three-phase static coordinate system is firstly established, and the reasons for the existence of double low-frequency pulsation on the direct current side and the occurrence of third harmonic on the alternating current side when the power grid voltage is unbalanced are obtained through analysis. Providing a direct current side to adopt double closed loop control, wherein the current inner loop is directly controlled by a proportional integral resonance (proportional integral resonance, PIR) controller to inhibit double low frequency pulsation of the direct current output side; an alternating-current side capacitance current compensation method is used for eliminating the influence of alternating-current side capacitance current on the system power factor, and simultaneously, active damping control is introduced into the alternating-current side capacitance voltage feedback to inhibit the parallel resonance of the alternating-current side LC; designing a trap to eliminate third harmonic in a modulation link, and realizing high sine of the current at the network side; finally, the method proposed in this embodiment is verified through simulation and experiment.
1. Three-phase CSR system operation analysis
The topology of a three-phase current source PWM rectifier is shown in fig. 1. Wherein e k For the grid voltage, i k For the net side current flow,i sk for rectifier ac side current, k=a, b, c; l (L) ac 、C ac Filtering inductance and capacitance of rectifier AC side, L dc 、C dc Respectively a direct-current side inductor and a capacitor, D is a freewheeling diode, R L For the load resistance, u dc 、i dc Output voltage and current from DC side, u o 、i o The load voltage and current, respectively.
Establishing a mathematical model of CSR under a three-phase static coordinate system, and obtaining according to kirchhoff voltage and current law:
wherein: u (u) ck For ac side filter capacitor voltage, sigma k As a switching function.
The unbalanced grid voltage can be expressed as:
wherein: e (E) p 、E n The magnitudes of the positive sequence component and the negative sequence component, respectively; omega is the fundamental wave angular frequency of the power grid voltage; alpha p 、α n The initial phase angles of the positive and negative sequence components, respectively.
Converting the three-phase power grid voltage through Park conversion to obtain voltage e under d-q synchronous rotation coordinates d 、e q The net side current is converted to i by Park d 、i q The network side power can be expressed as:
the net side instantaneous power expression is:
wherein:
wherein:respectively positive sequence and negative sequence components of the voltage; />The positive and negative sequence components of the current, respectively.
As can be seen from equation (4), both the network side active and reactive power contain second harmonic components at unbalanced network voltages. Neglecting the system loss, the system power conservation can know that the direct-current side voltage and current will also have a second harmonic component. Neglecting the effect of the dc side capacitance, the dc side output voltage, current can be expressed as:
wherein: u (U) dc As the average value of the DC side voltage, u dc1 、u dc2 The amplitudes of the cosine and sine components of the second voltage harmonic are respectively; i dc Is the average value of direct current side current, i dc1 、i dc2 The amplitudes of the cosine and sine components of the second current harmonic are respectively obtained.
Only the switching function basis wave quantity is considered in a steady state:
σ k (t)＝msin(ωt+β) (6)
wherein: m is a modulation factor, m is more than or equal to 0 and less than or equal to 1, and beta is the initial phase angle of the fundamental wave.
The further expression for the net side current is:
from equation (7), the second harmonic of the dc side output current causes the net side current to contain the third harmonic.
2. CSR control strategy under unbalanced grid voltage
At present, under the working condition of three-phase unbalanced power grid voltage, a common control strategy of CSR is to start from two angles of positive and negative sequence separation of power grid voltage and constant active power of an input side, improve the current waveform of the grid side by inhibiting the negative sequence component of the current of the grid side, and remove low-frequency pulsation of the direct current side. However, the control strategy needs to extract positive and negative sequence components of the power grid voltage, and the extraction is complex and the calculated amount is large through a phase-locked loop and other methods.
In view of this problem, this embodiment proposes a novel control scheme without extracting positive and negative sequence components of the grid voltage, as shown in fig. 2, by designing a control loop to directly suppress the direct-current output side double frequency component, and then according to equation (7), it can be known that when the direct-current side double frequency component is attenuated, the third harmonic quantity of the grid side current is also reduced, thereby realizing the grid side current high sinusoidal.
The control principle is as follows: DC side load voltage u o And a DC side load voltage reference valueSubtracting to obtain a voltage error signal, and obtaining a direct current reference value component +.>Continue to output current i from DC side dc Comparing to obtain a current error signal, obtaining +.>And the direct-current side double frequency pulsation suppression is realized.
The function of the three-phase CSR network side LC is to filter out harmonic waves of network side current, but the network side voltage and current generate phase differences due to the influence of filter capacitor current, so that the system cannot operate under unit power factor.
As shown in fig. 3, taking a phase a as an example, e a 、i a Representing the voltage and current at the network side, i sa Represents an alternating-current side current, u la Represents the voltage across the ac side filter inductance, u ca 、i ca The voltage and current of the two ends of the alternating current side filter capacitor are represented, the input power factor angle existing between the network side voltage and current when CSR operates in a non-unit power factor mode is represented by delta, and the angle between the network side voltage and the alternating current side is represented by deltaAnd (3) representing. It can be seen from FIG. 4 that in order to increase the input power factor of the system, it is necessary to increase the unit power factor operation of the system>And delta is reduced. The operation principle of the three-phase CSR can be obtained by:
wherein: e (E) m Represented as net side voltage effective value.
From formula (8), it can be seen that in U dc And E is m While remaining unchanged, increasing the modulation factor m increases. The present embodiment thus designs a compensation current method by reducing i ca So that i sa Increase, thereby increasing m and->The input power factor of the system is improved. From formula (1), CSR can be obtained at d-q coordinates:
namely:
in order to reduce the effect of system noise, the differential term in equation (10) above, i.e cd ＝-ωC ac u q 、i cq ＝ωC ac u d When designing the compensation currentThe filter capacitance current i can be effectively reduced ca And at the same time, the system input power factor is increased.
Meanwhile, the three-phase CSR is also susceptible to PWM harmonic waves or system step response, so that the network side LC circuit generates parallel resonance. In order to solve the problem, the embodiment introduces a high-pass filter to obtain harmonic components of the alternating-current side filter capacitor near the resonant frequency, and effectively inhibits resonance through the capacitor voltage feedback active damping loop. The control block diagram is shown in FIG. 5, in which K Cv Representing the corresponding feedback gain.
The HPF transfer function is taken as s/(s+ω) h ) The transfer function of the net side current versus the ac side current is:
wherein: a is that 1 ＝1，A 0 ＝ω h ；B 3 ＝L ac C ac ，B 2 ＝K Cv L ac +L ac C ac ω h ，B 1 ＝1，B 0 ＝ω h 。
By reasonably setting the feedback gain, the damping coefficient and oscillation frequency of the system can be changed, and as can be seen from FIG. 6, the feedback gain K is increased within the range of (0,0.65) Cv The LC resonance spike will gradually decay and gradually exhibit an over-damped characteristic. Comprehensively consider, select K Cv =0.25 can satisfy resonance suppression and does not cause saturation of the controller output.
Introducing a power factor compensation termAnd->Active damping Loop control term +.>And->And (3) comprehensively obtaining:
obtaining a modulation signal i under two-phase stationary coordinates through coordinate transformation α And i β 。
3. Loop controller design
The DC side of the three-phase CSR adopts a double closed loop control strategy, as shown in FIG. 7, wherein G v (s)，G i (s) respectively a voltage outer loop PI controller and a current inner loop PIR controller, K m For system voltage gain, take K m ＝1.5E m 。
The PIR controller is connected with a resonance controller in parallel on the basis of the PI controller. In the PIR controller, PI is used for tracking control of the direct current signal, and R is used for tracking control of the low frequency ripple signal, which realizes no-difference tracking of the direct current signal and the low frequency ripple signal at the same time. It should be noted that, in the PIR controller design of the present embodiment, the control objective of R is to implement the non-difference tracking of the direct-current side frequency doubling component, so the value of the component in the corresponding R should be twice the fundamental angular frequency of the grid voltage, i.e. 2ω. At this point the PIR controller expression:
wherein: k (k) p 、k i 、k r Respectively proportional, integral, resonant gain coefficients.
The current inner loop control block diagram is shown in fig. 8:
the current inner loop open loop transfer function is obtained by utilizing the Meissen formula:
the finishing formula (14) can be obtained:
wherein: a, a 4 ＝K m k p R L C dc ；a 3 ＝K m k p +K m (k i +k r )R L C dc ；a 2 ＝K m (k i +k r )+K m 4ω 2 k p R L C dc ；a 1 ＝K m (4ω 2 k p +4ω 2 k i R L C dc )；a 0 ＝K m 4ω 2 k i ；b 5 ＝R L L dc C dc ；b 4 ＝L dc ；b 3 ＝4ω 2 R L L dc C dc +R L ；b 2 ＝4ω 2 L dc ；b 1 ＝4ω 2 R L ；b 0 ＝0。
Determining k in PIR controller p 、k i The parameters are then substituted into equation (15), when k r The current open loop transfer function Bode obtained when taking different values is shown in fig. 9.
It can be seen that with k r The stronger the system tracking doubling capability is, and the whole control system can further realize the inhibition of direct current side doubling pulsation, however, the overlarge k is r Will affect other frequencies, comprehensively consider and select k in order to ensure the overall performance of the system r 100.
The final current inner loop open-closed loop transfer function Bode chart is shown in fig. 10, at the moment, the phase margin is 87.3 degrees, the cut-off frequency is 1910Hz, the system bandwidth is 2100Hz, and the system bandwidth is about 1/10 of the switching frequency, so that the system performance requirement is met.
4. System modulation strategy
SVM is the most commonly used modulation method for CSR, and the switching state and sector division of conventional six-sector modulation are shown in table 1.
TABLE 1 switch State and sector division
However, the conventional six-sector modulation strategy has the defect of high switching power loss of the system, and for this problem, the twelve-sector modulation strategy is adopted in this embodiment, and the space vector distribution is shown in fig. 11. As shown in Table 2, the average switching voltage of the system can be reduced by using the bilateral symmetry current vector sequence, so that the power loss is reduced, and the three-phase CSR operation efficiency is improved.
TABLE 2 Current space vector sequences for sectors
From the foregoing, it can be seen that the modulation signal i obtained by the coordinate transformation is due to the double frequency ripple of the current inner loop output signal α And i β The third harmonic will be contained and a trap can be used to eliminate the third harmonic. At this time, the value of the corresponding component of the wave trap should be three times of the fundamental wave angular frequency of the power grid voltage, namely 3 omega, and the expression is as follows:
wherein: k (K) 1 Is the controller coefficient.
A Bode diagram of the corresponding trap is shown in fig. 12. It can be seen that the trap is able to attenuate the input signal significantly at around 3 omega, i.e. 150Hz, without any effect at other frequencies.
5. Simulation and experimental result analysis
5.1 simulation verification
Based on Matlab/Simulink, a three-phase CSR simulation model is built, and steady-state and dynamic operation performances of the system are analyzed, so that the correctness of the proposed method is verified. The three-phase unbalanced power grid voltage parameter is e a ＝156∠0°，e b ＝131∠-115°，e c =131++125°, voltage waveform as shown in fig. 13, system simulation parameters as shown in table 3:
TABLE 3 main parameters of CSR
When the conventional double closed loop is adopted, the simulation result of the system in steady state operation is shown in fig. 14 and 15, wherein the load reference voltage is 100V.
It can be seen that although the grid voltage and the grid-side current remain in phase, they cannot achieve suppression of the direct-side load voltage double frequency ripple, while the grid-side current total harmonic distortion (Total Harmonic Distortion, THD) is 6.35%, which does not meet GB-T/14549-1993 and IEEE519 standards.
Fig. 16 to fig. 18 show simulation results when the current inner loop adopts PIR control, the system grid voltage and the grid-side current keep the same phase, the active power is kept constant, the reactive power has smaller double frequency pulsation, and the system power factors in the stage all reach more than 0.985, so that the double frequency pulsation of the direct-current side load voltage is effectively inhibited; the net side current THD is 2.23% and meets the GB-T/14549-1993 and IEEE519 standards.
Fig. 19 shows a simulated waveform for a step change in load from 5.6Ω to 11.2Ω. The result shows that the network side current has no obvious distortion in the abrupt change process, the direct current side load voltage can complete the tracking of a given reference value within 20ms, and the system is verified to have good dynamic performance.
5.2, experiment verification
In order to verify the feasibility of the control strategy provided by the embodiment, a CSR experimental platform is built, as shown in fig. 20. The power switching tube and the series diode adopt Indeluxe FF100R12RT4 and Ai Saisi MEA75-12DA respectively, the control chip adopts a Texas instrument TMS320F28335, and the logic conversion unit adopts a Siraisin XC95288. The generation and downloading process of the control codes is automatically completed through Matlab and a Composer Studio (CCS V5), and experimental parameters are the same as simulation parameters.
The programmable power supply Chroma-61702 is adopted to simulate the unbalanced state of the power grid voltage, and the waveform of the power grid voltage is shown in figure 21.
When the traditional double closed-loop control is adopted, the waveforms of the a-phase power grid voltage, the grid side current and the direct current side load voltage are shown in fig. 22 and 23 when the system runs stably, and the fact that the power grid voltage and the grid side current are kept in the same phase can be seen, but the inhibition of double frequency pulsation of the direct current side load voltage can not be realized, meanwhile, the THD value of the grid side current reaches 7.92%, and the THD value is basically consistent with a simulation result.
Fig. 24-26 show experimental results of CSR steady state operation at different reference voltages when the current inner loop is PIR controlled. Wherein, e in FIG. 24 a ＝156V，In FIG. 25 e a ＝156V，/>In FIG. 26 e a ＝156V，/>
Under different operation conditions, the system a-phase power grid voltage and the grid-side current can both keep the same phase, the direct-current-side load voltage can accurately and rapidly track the reference voltage value, and meanwhile, the direct-current-side load voltage doubling pulsation can be effectively restrained, and experiments show that the control strategy provided by the embodiment has good steady-state performance on the three-phase CSR under different output powers under the unbalanced three-phase power grid voltage condition.
The proposed control strategy is at e a ＝156V、The harmonic analysis of the current at the time network side is shown in fig. 27, and is basically consistent with the simulation result, the THD of the current at the network side is only 2.78%, and the standards of GB-T/14549-1993 and IEEE519 are met, so that the correctness of the control strategy and the parameter design of the controller provided by the embodiment is verified.
The experimental results of the grid voltage, grid side current and dc side load voltage obtained by the step change of the load from 5.6Ω to 11.2Ω are shown in fig. 28.
The control strategy provided by the embodiment has good dynamic performance on the three-phase CSR under the unbalanced working condition of the three-phase grid voltage, and the experimental result shows that the control strategy is consistent with the simulation result, the direct-current side load voltage can track the given value within 20ms, the grid side current has no obvious distortion in the abrupt change process.
The embodiment provides a direct-current-side-based current inner loop PIR control strategy, and introduces a fixed frequency trap, so that the problems that a direct current side has double frequency pulsation and a network side current has third harmonic when a three-phase CSR system is unbalanced in power grid voltage are solved. The capacitive current compensation is adopted to solve the problem that the power factor is influenced by the capacitive current at the network side, and the system power factor is effectively improved. From simulation and experimental verification, the following conclusions were drawn:
1) The traditional direct current double closed loop structure has obvious direct current side double frequency pulsation under the unbalanced working condition of the power grid voltage, and the third harmonic of the current at the grid side is larger, and meanwhile, the power factor is lower.
2) Under three different output powers, the direct current side pulsation can be well restrained, the net side current THD value is less than 3%, the unit power factor operation is realized, and meanwhile, the adjusting time is less than 20ms under the load abrupt change working condition, and the dynamic performance is good.
3) Compared with the traditional control scheme, the method provided by the embodiment does not need positive and negative sequence component calculation, is simple and efficient in algorithm, omits a network side current sensor, and effectively reduces system cost.
The foregoing description of the preferred embodiments of the invention is not intended to be limiting, but rather is intended to cover all modifications, equivalents, and alternatives falling within the spirit and principles of the invention.

Claims (5)

1. A three-phase CSR proportional-integral resonance control method under unbalanced power grid voltage is characterized by comprising the following steps:
s1, obtaining direct-current side load voltage u o And a DC side output current i dc ；
S2, voltage outer loop control: load voltage u on DC side o And a DC side load voltage reference valueSubtracting to obtain a voltage error signal, and obtaining a direct current reference value component by PI control of the voltage error signal>
S3, current inner loop control: to direct current reference value componentAnd direct current side output current i dc Comparing to obtain a current error signal, obtaining +.>Will->Adding the net side current i under d-q synchronous rotation coordinate system as feedback item d And i q ；
S4, will i d And i q Obtaining a modulation signal i under two-phase stationary coordinates through coordinate conversion α And i β After passing through the wave trap, the current source type PWM rectifier is controlled by adopting space vector pulse width modulation;
before the step S4, the net side current i under the d-q synchronous rotation coordinate system d And i q Adding power factor compensation termAnd->
Wherein: c (C) ac Is an alternating current side filter capacitor, u q And u d The voltage of the filter capacitor at the alternating current side under the d-q synchronous rotation coordinate system;
before the step S4, the net side current i under the d-q synchronous rotation coordinate system d And i q Adding active damping loop control itemAnd->Obtaining the net side current i under d-q synchronous rotation coordinate system d And i q The method comprises the following steps of:
wherein,and->AC side under d-q synchronous rotation coordinatesCapacitor voltage u q And u d Filtering by high-pass filter and then the source damping feedback gain is multiplied to obtain:
wherein: k (K) Cv Is an active damping feedback gain;
the expression of the high-pass filter is:
wherein: omega h Is a high pass filter cut-off frequency.
2. The method for three-phase CSR proportional-integral-resonance control under unbalanced grid voltage of claim 1, wherein the PIR controller is:
wherein: k (k) p 、k i 、k r Respectively proportional, integral, resonant gain coefficients.
3. The method for controlling three-phase CSR proportional-integral resonance under unbalanced network voltage according to claim 2, wherein in step S3, the transfer function of the current inner loop control is:
wherein: a, a 4 ＝K m k p R L C dc ；a 3 ＝K m k p +K m (k i +k r )R L C dc ；a 2 ＝K m (k i +k r )+K m 4ω 2 k p R L C dc ；a 1 ＝K m (4ω 2 k p +4ω 2 k i R L C dc )；a 0 ＝K m 4ω 2 k i ；b 5 ＝R L L dc C dc ；b 4 ＝L dc ；b 3 ＝4ω 2 R L L dc C dc +R L ；b 2 ＝4ω 2 L dc ；b 1 ＝4ω 2 R L ；b 0 ＝0；
Wherein K is m For the system voltage gain, R L Is a load resistor, C dc Is a direct-current side capacitor L dc Is a direct current side inductance.
4. The method for three-phase CSR proportional-integral-resonance control at unbalanced network voltages of claim 1, wherein the trap is used to cancel the modulated signal i α And i β The trap is:
wherein: k (K) 1 Is the controller coefficient.
5. The method for controlling three-phase CSR proportional-integral resonance under unbalanced network voltage according to claim 1, wherein in step S4, the space vector pulse width modulation adopts a twelve-sector modulation strategy, and the vector sequence of each sector is a bilateral symmetry current vector sequence.
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