CN114079412A - Motor prediction control method based on phase voltage duty ratio calculation - Google Patents

Motor prediction control method based on phase voltage duty ratio calculation Download PDF

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CN114079412A
CN114079412A CN202111374652.8A CN202111374652A CN114079412A CN 114079412 A CN114079412 A CN 114079412A CN 202111374652 A CN202111374652 A CN 202111374652A CN 114079412 A CN114079412 A CN 114079412A
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motor
voltage
axis
current
torque
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CN114079412B (en
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黄银国
郑博轩
赵美蓉
郑叶龙
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Tianjin University
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Tianjin University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/20—Estimation of torque
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

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  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

The invention discloses a motor prediction control method based on phase voltage duty ratio calculation, which comprises the following steps: acquisition Udc、Ia、Ib、IcMeasuring the motor speed omegarAnd a rotation angle θ; calculating deltai、δj、δ0Verifying the reasonableness of the torque, eliminating unsatisfied torque and calculating the torque Te(k +1) and flux linkage prediction value
Figure DDA0003363408350000011
Solving the objective function epsilon (U) to make the control effectOptimal solution delta for the fruit closest to the reference quantityopi、δopj、δop(ii) a Calculating ud(k+1|k)、uq(k +1| k), converted to uα(k+1|k)、uβ(k +1| k); determining the sector where the voltage vector is located according to theta, and determining uα(k+1|k)、uβ(k +1| k) is converted into the components U of the two standard voltage vectors that make up the sector boundaryb1、Ub2Calculating the corresponding effective time Tb1、Tb2(ii) a And determining a switching time node of the three-phase inverter by a seven-segment SVPWM (space vector pulse width modulation) method, establishing a comparator to control the switching state of the three-phase inverter, and supplying a three-phase voltage to a driving motor to operate according to a set control effect. The invention can improve the stability of motor control and reduce overshoot in the process of predictive control.

Description

Motor prediction control method based on phase voltage duty ratio calculation
Technical Field
The invention relates to the technical direction of motor control, belongs to the field of electromechanics, and particularly relates to a motor prediction control method based on phase voltage duty ratio calculation.
Background
In a mechanical transmission system, torque is one of the most typical mechanical quantities for the system performance of production equipment, and is widely applied to equipment such as motor load simulation and torque wrenches. The traditional electric torque generation mode is traditional direct torque control, and the direct control of the motor torque is realized by selecting a proper voltage vector, but the control stability is poor, the torque and the current pulsation are large during stabilization, and the increasingly-improved high-precision torque control cannot be met. Model Predictive Control (MPC) is an online optimization Control algorithm generated in industrial process Control, with the development of micro-Control technology, MPC can realize short-period Control processes such as servo drive, etc., and because it can select the optimal Control parameters at the next moment according to a target function, online correction is realized, and the MPC has high dynamic response Control performance. However, voltage vector prediction still exists because the control period and the voltage vector amplitude are constant, and the torque obtained by prediction control has obvious pulsation.
Disclosure of Invention
The invention aims to overcome the defects in the prior art and provides a motor prediction control method based on phase voltage duty ratio calculation. The control method can further improve the stability of motor control and reduce overshoot caused by fixed voltage vector amplitude in the process of predictive control.
The purpose of the invention is realized by the following technical scheme.
The invention relates to a motor prediction control method based on phase voltage duty ratio calculation, which comprises the following processes:
the first step is as follows: data acquisition
The AC power supply is switched on, AC-DC conversion of signals is realized through the rectifier module, direct current voltage signals are obtained to supply power for each module in the motor monitoring control system, and DC-AC conversion of the signals is realized through the inverter module in the motor monitoring control system to supply power for the motor; DC voltage U is collected on DC bus of motor by DC voltage sensordcRespectively collecting three-phase current signals I by using direct current sensors on three-phase current signal wires of the motora、Ib、Ic(ii) a Counting information is obtained through a photoelectric incremental encoder arranged on the motor, and the rotating speed omega of the motor is decoded and measuredrAnd a rotation angle θ;
the second step is that: duty cycle calculation
According to the three-phase current signal I obtained by collectiona、Ib、IcAnd a rotation angle theta, and obtaining an instantaneous current i by using Park conversiond(k) And iq(k) Let id(l)＝id(k)，iq(l)＝iq(k) Calculating the corresponding delta by substituting the formula (1) and the formula (2) into the formula (3)iAnd deltajThen, the zero vector duty ratio delta is calculated according to the formula (4)0；
Figure BDA0003363408330000021
id *(l+1)＝0 (2)
Figure BDA0003363408330000022
δ0＝1-max(δi,δj) (4)
Wherein id(l) And iq(l) The components i of the measured current in the l period of the current loop on the d-axis and q-axis, respectivelyd(k) And iq(k) The components of the current value of the torque loop in the k period on the d axis and the q axis are respectively, wherein k is nl, and n is the number of loops required for calibration; i.e. id *(l+1)、iq *(l +1) is the period i of the current loop l +1, respectivelyd(l+1)、iqA reference amount of (l + 1); u shapedi、UdjAre respectively a voltage vector Ui、UjComponent of projection on d-axis, Uqi、UqjAre respectively a voltage vector Ui、UjComponent of projection on the q-axis, δi、δjAre respectively corresponding voltage vectors Ui、UjDuty cycle during modulation; t issiIs the control period of the current loop; l issIs the stator inductance; p is a radical ofnThe number of pole pairs of the motor is;
Figure BDA0003363408330000023
is a permanent magnet flux linkage; t ise *(l +1) is a reference torque value at the current loop l +1 cycle time;
to delta according to the formula (5)i、δj、δ0The rationality of (3) is verified, the possibility that the equation (5) is not satisfied is eliminated, and the remaining torque T at the time of the torque loop k +1 cycle is calculated from the equations (6) to (10)e(k +1) and flux linkage prediction value
Figure BDA0003363408330000031
Figure BDA0003363408330000032
Figure BDA0003363408330000033
Figure BDA0003363408330000034
Figure BDA0003363408330000035
Figure BDA0003363408330000036
Figure BDA0003363408330000037
Wherein id(k+1)、iq(k +1) is the component of the current value in the torque loop k +1 cycle on the d-axis and q-axis, respectively, TsωIs the control period of the torque loop, L is the stator inductance,
Figure BDA0003363408330000038
the components of the flux linkage values on the d-axis and the q-axis, T, respectively, during the k +1 period of the torque ringe(k +1) is a predicted value of the k +1 period of the torque,
Figure BDA0003363408330000039
is the predicted value of k +1 period of flux linkage;
an optimal solution delta for making the control effect closest to the reference quantity is obtained according to the objective function epsilon (U)op、δopj、δop0；
Figure BDA00033634083300000310
Wherein, Te *(k +1) is the reference torque value at the moment of the torque loop k +1 cycle, β is the influence factor of the balancing torque and the flux linkage,
Figure BDA00033634083300000311
is the reference flux linkage value at the moment of the torque ring k +1 cycle;
the third step: PWM modulation
According to the optimal solution deltaop、δopj、δop0The decomposition voltage vector u under the d-q coordinate axis in the k +1 period is calculated by combining with the corresponding voltage vectord(l+1)、uq(l+1)：
Figure BDA00033634083300000312
Wherein, UopdiIs the optimal solution deltaopiCorresponding voltage vector UiComponent on d-axis, UopdjIs the optimal solution deltaopjCorresponding voltage vector UjComponent on d-axis, UopqiIs the optimal solution deltaopCorresponding voltage vector UiComponent on the q-axis, UopqjIs the optimal solution deltaojCorresponding voltage vector UjA component on the q-axis;
combining a motor rotation angle theta obtained by a photoelectric incremental encoder arranged on the motor, and converting a decomposition voltage vector u under a d-q coordinate axis by Anti-Parkd(k+1)、uq(k +1) into a Voltage vector u under an alpha-beta fixed coordinate Systemα(k +1) and uβ(k+1)；
Determining the sector of the voltage vector according to the rotation angle theta, and adding uα(k+1)、uβ(k +1) is converted into the component U of the two standard voltage vectors that make up the sector boundaryb1、Ub2And calculating a voltage vector U according to the formula (13)b1、Ub2Corresponding effective time Tb1、Tb2；
Figure BDA0003363408330000041
Figure BDA0003363408330000042
Wherein, Ub1、Ub2Clockwise and counterclockwise boundary voltage components of the sector, respectively;
according to the effective time T of the determined sector voltage vectorb1、Tb2And determining a switching time node of the three-phase inverter by a seven-segment SVPWM pulse width modulation method, and establishing a comparator to control the switching state of the three-phase inverter, so that the three-phase voltage driving motor operates according to a set control effect.
The motor monitoring control system comprises an alternating current power supply, a rectifying circuit, a motor, a DSP chip, an inverter, a direct current voltage sensor and a direct current sensor; the motor adopts an AC servo motor 80CB075C-500000 of Sen and Chu company series, the motor is provided with a photoelectric incremental encoder, and a DSP chip selects TMS320F28335, so that the motor has an AD conversion function; the direct-current voltage sensor is an HV300GB Hall voltage sensor and is connected to a direct-current bus of the motor; the direct current sensors are CHB-25NP/SP9 closed-loop Hall current sensors, three direct current sensors are arranged and are respectively connected to three-phase current signal wires of the motor; the output ends of the photoelectric incremental encoder, the direct-current voltage sensor and the direct-current sensor are connected with the input end of the DSP chip; the inverter is FSBB30CH60, the input end of the inverter is connected with the output end of the DSP chip, and the output end of the inverter is connected with the motor; the rectification circuit adopts an uncontrollable silicon rectification module, the input end of the uncontrollable silicon rectification module is connected with an alternating current power supply, and the output end of the uncontrollable silicon rectification module is respectively connected with each component needing power supply in the motor monitoring control system.
According to the on-off of 6 power switches of the three-phase inverter, 8 different switch states can be generated in total, and 2 zero voltage vectors U with the same effect are included0、U7Voltage vectors U of the same magnitude as 61、U2、U3、 U4、U5、U6In the second step, the voltage vector Ui、UjAll can take U0、U1、U2、U3、U4、U5、U6、 U7。
The complex plane α - β is divided into six sectors according to the voltage space vector: in an alpha-beta coordinate system, the counterclockwise direction is positive, when the included angle theta between the clockwise direction and the alpha axis is 0-60 degrees, the clockwise direction is divided into a sector I, when the included angle theta is 60-120 degrees, the counterclockwise direction is divided into a sector II, when the included angle theta is 120-180 degrees, the clockwise direction is divided into a sector III, when the included angle theta is 180-240 degrees, the counterclockwise direction is divided into a sector IV, when the included angle theta is 240-300 degrees, the clockwise direction is divided into a sector V, and when the included angle theta is 300-360 degrees, the clockwise direction is divided into a sector VI.
Compared with the prior art, the technical scheme of the invention has the following beneficial effects:
(1) according to the invention, after the two effective vectors and the zero vector are synthesized, the direction and the amplitude of the synthesized voltage vector can be changed, and the current control performance of motor control is improved.
(2) The invention provides a duty ratio prediction calculation method by utilizing the relation between the double closed-loop control cycles, realizes the prediction model control based on the duty ratio, avoids the overshoot phenomenon caused by the traditional voltage vector in the prediction control, further improves the dynamic control performance in the motor prediction control, and reduces the torque pulsation in the control.
(3) The invention introduces a judgment mechanism of the reasonability of the duty ratio, screens out unreasonable duty ratio in the prediction model, reduces the data calculation amount in the prediction model, saves the calculation time and improves the dynamic performance of the control model.
Drawings
FIG. 1 is a d-q coordinate system diagram;
FIG. 2 is a diagram of voltage vector versus sector;
FIG. 3 is a schematic view of a motor monitoring control system;
fig. 4 is a flow chart of a motor monitoring control system.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more apparent, embodiments of the present invention are described in further detail below. It should be understood that the detailed description and specific examples, while indicating the present invention, are given by way of illustration and explanation only, not limitation.
The invention relates to a motor prediction control method based on phase voltage duty ratio calculation, which comprises the following processes:
the first step is as follows: data acquisition
And the AC power supply is switched on, AC-DC conversion of signals is realized through the rectifying module, direct current voltage signals are obtained to supply power for each module in the motor monitoring control system, and DC-AC conversion of signals is realized through the inverter module in the motor monitoring control system to supply power for the motor. DC voltage U is collected on DC bus of motor by DC voltage sensordcRespectively collecting three-phase current signals I by using direct current sensors on three-phase current signal wires of the motora、Ib、Ic. Counting information is obtained through a photoelectric incremental encoder arranged on the motor, and the rotating speed omega of the motor is decoded and measuredrAnd a rotation angle theta.
As shown in fig. 3 and 4, the motor monitoring and controlling system includes an ac power supply, a rectifying circuit, a motor, a DSP chip, an inverter, a dc voltage sensor, and a dc current sensor. The motor adopts an AC servo motor 80CB075C-500000 of Senjian company series, the motor is provided with a photoelectric incremental encoder, and a DSP chip selects TMS320F28335, so that the motor has an AD conversion function. The direct-current voltage sensor is an HV300GB Hall voltage sensor and is connected to a direct-current bus of the motor. The direct current sensors are CHB-25NP/SP9 closed-loop Hall current sensors, three direct current sensors are arranged, and the three direct current sensors are respectively connected to three-phase current signal wires of the motor. And the output ends of the photoelectric incremental encoder, the direct-current voltage sensor and the direct-current sensor are connected with the input end of the DSP chip. The inverter is FSBB30CH60, the input end of the inverter is connected with the output end of the DSP chip, and the output end of the inverter is connected with the motor. The rectification circuit adopts an uncontrollable silicon rectification module, the input end of the uncontrollable silicon rectification module is connected with an alternating current power supply, and the output end of the uncontrollable silicon rectification module is respectively connected with each component needing power supply in the motor monitoring control system.
The second step is that: duty cycle calculation
The relationship between the stator flux linkage of the motor and the three-phase coordinate system is shown in figure 1,
Figure BDA0003363408330000061
is the stator flux linkage of the motor,
Figure BDA0003363408330000062
the component of the stator flux linkage on the d-axis,
Figure BDA0003363408330000063
the component of the stator flux linkage on the q axis is shown, and delta is the included angle between the stator flux linkage of the motor and the d axis.
The stator flux linkage of the permanent magnet synchronous motor can be expressed in a projection form in a d-q coordinate system as follows:
Figure BDA0003363408330000064
Figure BDA0003363408330000065
wherein, the stator current equation under the d-q coordinate system can be obtained according to the formula (1) and the formula (2) as follows:
Figure BDA0003363408330000066
Figure BDA0003363408330000067
wherein idComponent of current in d-axis, iqComponent of current in q-axis, Ld、LqStator inductances of d-axis and q-axis respectively,
Figure BDA0003363408330000068
is a permanent magnet flux linkage. The electromagnetic torque T is obtained by combining the formula (3) and the formula (4)eThe equation of (c):
Figure BDA0003363408330000069
wherein p isnFor the number of motor pole pairs, substituting equation (3) and equation (4) into equation (5) can further result in:
Figure BDA00033634083300000610
from equation (6), it can be seen that the electromagnetic torque of the motor includes two parts: electromagnetic torque generated by magnetic field interaction between the stator and rotor of the motor; reluctance torque generated by salient pole structures of the motor. For a surface-mounted three-phase permanent magnet synchronous motor, the stator inductance Ld＝Lq＝LsTherefore, the torque increment equation of the motor can be taken:
Figure BDA0003363408330000071
wherein, Delta TeAnd delta is the variation of the motor torque and the rotor variation angle in one cycle respectively. In a control period, because the mechanical time constant is far greater than the electrical time constant, the position change of the rotor of the motor is small, so that the torque angle or the stable amplitude value can be rapidly changed by controlling the stator flux linkage, and the torque can be rapidly changed. According to the relation between the motor torque and the flux linkage, the change of the flux linkage equation can directly control the motion state of the motor. The flux linkage equation of the motor is as follows:
Figure BDA0003363408330000072
in the formula us、Rs、isThe stator voltage, the stator resistance and the stator current are respectively, and if the influence of the stator resistance is neglected, the flux linkage formula of the motor can be simplified into the integral of the stator voltage with respect to time. Finally, the inverter is used for controlling the change of the three-phase voltage of the motor, so that the output torque of the motor can be changed, and the aim of torque control is fulfilled.
According to the on-off of 6 power switches of the three-phase inverter, 8 different switch states can be generated in total, and 2 zero voltage vectors U with the same effect are included0、U7Voltage vectors U of the same magnitude as 61、U2、U3、 U4、U5、U6As shown in table 1. Meanwhile, the complex plane (α - β coordinate system) may be divided into six sectors according to the voltage space vector, as shown in table 2. The correspondence of the voltage vectors to the sectors is shown in fig. 2.
TABLE 1 Voltage vector vs. switch diagram
Figure BDA0003363408330000073
Table 2 sector division table
Figure BDA0003363408330000074
According to the principle of voltage vector synthesis, the stator voltage vector u can be obtainedsDecomposed into voltage vectors Ui、Uj、 U0And its action time Ti、Tj、T0The product relationship is shown in equation (9). In this case, any voltage vector within the hexagonal area composed of six voltage vectors can be generated by the three-phase inverter.
Figure BDA0003363408330000081
Under the assumption that the eddy current, the hysteresis loss and the rotor damping winding of the motor are not calculated, the stator magnetic field is in sinusoidal distribution, and the induced electromotive force of the stator three-phase winding is a sine wave, the mathematical model of the permanent magnet synchronous motor under the d-q coordinate system can be expressed as follows:
Figure BDA0003363408330000082
wherein, ω isrIs the electrical angular velocity of the rotor, udIs the component on the d-axis of the motor stator voltage, uqIs the component on the q-axis of the motor stator voltage. The voltage of each control period of the motor is controlled by a three-phase inverter which consists of 6 power switching devices, and the upper power switching device and the lower power switching device are conducted in a complementary mode.
In the motor control process, the torque loop in the double closed loop control has a period which is usually longer than the control period of the current loop because of the need of carrying out torque calibration, so that the set value of the current can be considered to be kept unchanged in the shorter control period of the current loop.
Figure BDA0003363408330000083
Wherein id(l) And iq(l) The components of the measuring current on the d axis and the q axis during the l period of the current loop are respectively; i.e. id *(l+1)、iq *(l +1) is the period i of the current loop l +1, respectivelyd(l+1)、iqA reference amount of (l + 1); u shapedi、 UdjAre respectively a voltage vector Ui、UjA projection component on the d-axis; u shapeqi、UqjAre respectively a voltage vector Ui、UjA projection component on the q-axis; deltai、δjAre respectively corresponding voltage vectors Ui、UjDuty cycle during modulation; t issiIs the control period of the current loop; l issIs the stator inductance.
The effective duty cycle delta in the corresponding partition can be solved according to the equation setiAnd deltajWherein the value of the duty cycle should be equal to or greater than 0 and equal to or less than 1. Zero vector duty cycle delta0It should be expressed as:
δ0＝1-max(δi,δj) (12)
because of the adoption of idThe control scheme is 0, so that the i of the current loop at the moment of the l +1 th cycle is calculated according to the torque reference set by the motorq *(l+1)。
Figure BDA0003363408330000084
id *(l+1)＝0 (14)
Wherein p isnThe number of pole pairs of the motor is;
Figure BDA0003363408330000091
is a permanent magnet flux linkage; t ise *(l +1) is the reference torque value at the moment of the cycle of current loop l + 1.
According to the three-phase current signal I obtained by collectiona、Ib、IcAnd a rotation angle theta, and obtaining an instantaneous current i by using Park conversiond(k) And iq(k) Let id(l)＝id(k)，iq(l)＝iq(k)，id(k) And iq(k) The current values at the time of k cycles of the torque loop are components on the d axis and the q axis, respectively, where k is nl and n is the number of loops required for calibration. Substituting equation (11) along with equations (13) and (14) calculates the corresponding δiAnd deltajThen, the zero vector duty ratio delta is calculated according to the formula (12)0。
To δ according to equation (15)i、δj、δ0The rationality of (c) is verified, the possibility that the equation (15) is not satisfied is excluded, and the remaining torque T at the time of the torque loop k +1 cycle is calculated from the equations (16) to (20)e(k +1) and flux linkage prediction value
Figure BDA0003363408330000092
Figure BDA0003363408330000093
Figure BDA0003363408330000094
Figure BDA0003363408330000095
Figure BDA0003363408330000096
Figure BDA0003363408330000097
Figure BDA0003363408330000098
Wherein id(k+1)、iq(k +1) is the component of the current value in the torque loop k +1 cycle on the d-axis and q-axis, respectively, TsωIs the control period of the torque loop, L is the stator inductance,
Figure BDA0003363408330000099
the components of the flux linkage values on the d-axis and the q-axis, T, respectively, during the k +1 period of the torque ringe(k +1) is a predicted value of the k +1 period of the torque,
Figure BDA00033634083300000910
is a predictor of the k +1 period of the flux linkage.
According to the control requirements, an objective function epsilon (U) can be constructed, and the control accuracy conditions of different voltage vectors are evaluated:
Figure BDA00033634083300000911
wherein, Te *(k +1) is the reference torque value at the moment of the torque loop k +1 cycle, β is the influence factor of the balancing torque and the flux linkage,
Figure BDA0003363408330000101
at the moment of the torque ring k +1 cycleThe flux linkage value is referred to.
An optimal solution δ that brings the control effect closest to the reference amount is found from equation (21)opi、δopj、δop0That is, a predicted value T of the k +1 period of the torque and flux linkage for minimizing the objective function ε (U) is obtainede(k +1) and
Figure BDA0003363408330000102
further determining the corresponding duty ratio delta of the predicted valueopi、δopj、δop0。
The third step: PWM modulation
According to the optimal solution deltaop、δop、δop0The decomposition voltage vector u under the d-q coordinate axis in the k +1 period is calculated by combining with the corresponding voltage vectord(k+1)、uq(k+1)：
Figure BDA0003363408330000103
Wherein, UopdiIs the optimal solution deltaopiCorresponding voltage vector UiComponent on d-axis, UopdjIs the optimal solution deltaopjCorresponding voltage vector UjComponent on d-axis, UopqiIs the optimal solution deltaopiCorresponding voltage vector UiComponent on the q-axis, UopqjIs the optimal solution deltaopjCorresponding voltage vector UjThe component on the q-axis.
Combining a motor rotation angle theta obtained by a photoelectric incremental encoder arranged on the motor, and converting a decomposition voltage vector u under a d-q coordinate axis by Anti-Parkd(k+1)、uq(k +1) into a Voltage vector u under an alpha-beta fixed coordinate Systemα(k +1) and uβ(k+1)。
Determining the sector of the voltage vector according to the rotation angle theta, and adding uα(k+1)、uβ(k +1) is converted into the component U of the two standard voltage vectors that make up the sector boundaryb1、Ub2And calculating a voltage vector U according to equation (23)b1、Ub2Corresponding effective time Tb1、Tb2。
Figure BDA0003363408330000104
Figure BDA0003363408330000105
Wherein, Ub1、Ub2Are the clockwise and counterclockwise boundary voltage components of the sector, respectively.
Taking the voltage vector in the I-th sector as an example, u can be calculatedα(k +1) and uβ(k +1) at U1And U2The effective time of (3) is:
Figure BDA0003363408330000111
wherein, T1、T2Are respectively a voltage vector U1、U2The effective time of (a).
Determining the effective time T of the sector voltage vectorb1、Tb2And then, the switching time node of the three-phase inverter can be determined by selecting a seven-segment SVPWM method, and a comparator is established to control the switching state of the three-phase inverter, so that the three-phase voltage driving motor can be supplied to operate according to the set control effect.
While the present invention has been described in terms of its functions and operations with reference to the accompanying drawings, it is to be understood that the invention is not limited to the precise functions and operations described above, and that the above-described embodiments are illustrative rather than restrictive, and that various changes may be made therein by those skilled in the art without departing from the spirit and scope of the invention as defined by the appended claims.

Claims (4)

1. A motor predictive control method based on phase voltage duty ratio calculation is characterized by comprising the following processes:
the first step is as follows: data acquisition
The AC power supply is switched on, AC-DC conversion of signals is realized through the rectifier module, direct current voltage signals are obtained to supply power for each module in the motor monitoring control system, and DC-AC conversion of the signals is realized through the inverter module in the motor monitoring control system to supply power for the motor; DC voltage U is collected on DC bus of motor by DC voltage sensordcRespectively collecting three-phase current signals I by using direct current sensors on three-phase current signal wires of the motora、Ib、Ic(ii) a Counting information is obtained through a photoelectric incremental encoder arranged on the motor, and the rotating speed omega of the motor is obtained through decodingrAnd a rotation angle θ;
the second step is that: duty cycle calculation
According to the three-phase current signal I obtained by collectiona、Ib、IcAnd a rotation angle theta, and obtaining an instantaneous current i by using Park conversiond(k) And iq(k) Let id(l)＝id(k)，iq(l)＝iq(k) Calculating the corresponding delta by substituting the formula (1) and the formula (2) into the formula (3)iAnd deltajThen, the zero vector duty ratio delta is calculated according to the formula (4)0；
Figure FDA0003363408320000011
id *(l+1)＝0 (2)
Figure FDA0003363408320000012
δ0＝1-max(δi，δj) (4)
Wherein id(l) And iq(l) The components i of the measured current in the l period of the current loop on the d-axis and q-axis, respectivelyd(k) And iq(k) Are respectively torque ringsThe components of the current value in the k period on the d axis and the q axis are determined, wherein k is nl, and n is the number of rings required by calibration; i.e. id *(l+1)、iq *(l +1) is the period i of the current loop l +1, respectivelyd(l+1)、iqA reference amount of (l + 1); u shapedi、UdjAre respectively a voltage vector Ui、UjProjection component on d-axis, Uqi、UqjAre respectively a voltage vector Ui、UjComponent of projection on the q-axis, δi、δjAre respectively corresponding voltage vectors Ui、UjDuty cycle during modulation; t issiIs the control period of the current loop; l issIs the stator inductance; p is a radical ofnThe number of pole pairs of the motor is;
Figure FDA0003363408320000021
is a permanent magnet flux linkage; t ise *(l +1) is the reference torque value at the moment of the cycle of the current loop l + 1;
to delta according to the formula (5)i、δj、δ0The rationality of (3) is verified, the possibility that the equation (5) is not satisfied is excluded, and the remaining torque T at the time of the torque loop k +1 cycle is calculated from the equations (6) to (10)e(k +1) and flux linkage prediction value
Figure FDA0003363408320000022
Figure FDA0003363408320000023
Figure FDA0003363408320000024
Figure FDA0003363408320000025
Figure FDA0003363408320000026
Figure FDA0003363408320000027
Figure FDA0003363408320000028
Wherein id(k+1)、iq(k +1) is the component of the current value in the torque loop k +1 cycle on the d-axis and q-axis, respectively, TsωIs the control period of the torque loop, L is the stator inductance,
Figure FDA0003363408320000029
the components of the flux linkage values on the d-axis and the q-axis at the period of the torque loop k +1, Te(k +1) is a predicted value of the k +1 period of the torque,
Figure FDA00033634083200000210
is the predicted value of k +1 period of flux linkage;
an optimal solution delta for making the control effect closest to the reference quantity is obtained according to the objective function epsilon (U)opi、δopj、δop0；
Figure FDA00033634083200000211
Wherein, Te *(k +1) is the reference torque value at the moment of the torque loop k +1 cycle, β is the influence factor of the equilibrium torque and flux linkage,
Figure FDA00033634083200000212
is the reference flux linkage value at the moment of the torque ring k +1 cycle;
the third step: PWM modulation
According to mostOptimal solution of deltaop、δopj、δopThe decomposition voltage vector u under the d-q coordinate axis in the k +1 period is calculated by combining with the corresponding voltage vectord(k+1)、uq(k+1)：
Figure FDA0003363408320000031
Wherein, UopdiIs the optimal solution deltaopiCorresponding voltage vector UiComponent on d-axis, UopdjIs the optimal solution deltaopjCorresponding voltage vector UjComponent on d-axis, UopqiIs the optimal solution deltaopiCorresponding voltage vector UiComponent on the q-axis, UopqjIs the optimal solution deltaopjCorresponding voltage vector UjA component on the q-axis;
combining a motor rotation angle theta obtained by a photoelectric incremental encoder arranged on the motor, and converting a decomposition voltage vector u under a d-q coordinate axis by Anti-Parkd(k+1)、uq(k +1) into a Voltage vector u in an α - β fixed coordinate Systemα(k +1) and uβ(k+1)；
Determining the sector of the voltage vector according to the rotation angle theta, and adding uα(k+1)、uβ(k +1) is converted into the component U of the two standard voltage vectors that make up the sector boundaryb1、Ub2And calculating a voltage vector U according to the formula (13)b1、Ub2Corresponding effective time Tb1、Tb2；
Figure FDA0003363408320000032
Figure FDA0003363408320000033
Wherein, Ub1、Ub2Clockwise and counterclockwise boundary voltage components of the sector, respectively;
according to the effective time T of the determined sector voltage vectorb1、Tb2And selecting a switching time node of the three-phase inverter determined by a seven-segment SVPWM method, and establishing a comparator to control the switching state of the three-phase inverter, so that the three-phase voltage driving motor operates according to a set control effect.
2. The motor predictive control method based on phase voltage duty cycle calculation according to claim 1, characterized in that the motor monitoring control system comprises an alternating current power supply, a rectification circuit, a motor, a DSP chip, an inverter, a direct current voltage sensor and a direct current sensor; the motor adopts an AC servo motor 80CB075C-500000 of Sen and Chu company series, the motor is provided with a photoelectric incremental encoder, and a DSP chip selects TMS320F28335, so that the motor has an AD conversion function; the direct-current voltage sensor is an HV300GB Hall voltage sensor and is connected to a direct-current bus of the motor; the direct current sensors are CHB-25NP/SP9 closed-loop Hall current sensors, three direct current sensors are arranged and are respectively connected to three-phase current signal wires of the motor; the output ends of the photoelectric incremental encoder, the direct-current voltage sensor and the direct-current sensor are connected with the input end of the DSP chip; the inverter is FSBB30CH60, the input end of the inverter is connected with the output end of the DSP chip, and the output end of the inverter is connected with the motor; the rectification circuit adopts an uncontrollable silicon rectification module, the input end of the uncontrollable silicon rectification module is connected with an alternating current power supply, and the output end of the uncontrollable silicon rectification module is respectively connected with each component needing power supply in the motor monitoring control system.
3. The method as claimed in claim 1, wherein the motor predictive control method based on phase voltage duty ratio calculation is characterized in that 8 different switch states can be generated according to the on and off of 6 power switches of the three-phase inverter, and the switch states comprise 2 zero voltage vectors U with the same effect0、U7Voltage vectors U of the same magnitude as 61、U2、U3、U4、U5、U6In the second step, the voltage vector Ui、UiAll can take U0、U1、U2、U3、U4、U5、U6、U7。
4. The method of claim 1, wherein the complex plane α - β is divided into six sectors according to voltage space vectors: in an alpha-beta coordinate system, the anticlockwise direction is positive, when the included angle theta between the anticlockwise direction and an alpha axis is 0-60 degrees, the anticlockwise direction is divided into a sector I, when the included angle theta is 60-120 degrees, the anticlockwise direction is divided into a sector II, when the included angle theta is 120-180 degrees, the anticlockwise direction is divided into a sector III, when the included angle theta is 180-240 degrees, the anticlockwise direction is divided into a sector IV, when the included angle theta is 240-300 degrees, the anticlockwise direction is divided into a sector V, and when the included angle theta is 300-360 degrees, the anticlockwise direction is divided into a sector VI.
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