CN114079405B - Non-cascade predictive speed synchronous control method without weight coefficient setting suitable for double permanent magnet motor system - Google Patents
Non-cascade predictive speed synchronous control method without weight coefficient setting suitable for double permanent magnet motor system Download PDFInfo
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P5/00—Arrangements specially adapted for regulating or controlling the speed or torque of two or more electric motors
- H02P5/46—Arrangements specially adapted for regulating or controlling the speed or torque of two or more electric motors for speed regulation of two or more dynamo-electric motors in relation to one another
- H02P5/50—Arrangements specially adapted for regulating or controlling the speed or torque of two or more electric motors for speed regulation of two or more dynamo-electric motors in relation to one another by comparing electrical values representing the speeds
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
- H02P21/0017—Model reference adaptation, e.g. MRAS or MRAC, useful for control or parameter estimation
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
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Abstract
The invention discloses a non-cascade predictive speed synchronous control method without a weight coefficient, which is suitable for a double permanent magnet motor system, and based on the idea of unified modeling, a motion equation, a voltage equation and a mathematical model of a two-level voltage source inverter of a driving motor are combined to establish a unified model; then predicting a reference voltage vector with current and rotation speed information; in order to solve the problem of weight coefficient setting in the traditional cost function, simplify algorithm complexity, determine an improved cost function according to the proportion of synchronous errors and tracking errors, and select an optimal voltage vector and calculate vector acting time. And then considering the actual user requirement, integrating the current limitation into the selection of the alternative voltage vector to realize the limitation of the current. And finally, the dynamic response speed and the synchronous tracking precision of the system can be improved, and the system has the advantages of intuitive modeling, simple structure and the like.
Description
Technical Field
The invention relates to the technical field of double permanent magnet motor system control, in particular to a non-cascade predictive speed synchronous control method without weight coefficient setting, which is suitable for a double permanent magnet motor system.
Background
In the multi-motor speed synchronous control system, the quality of the synchronous performance directly influences the reliability of high-performance industrial systems such as printing, spinning, papermaking, electric automobiles and the like and the quality of produced products. Therefore, how to improve the performance of the multi-motor synchronous control system and the synchronous control precision is very important.
At present, the synchronous control structure of multiple motors is mainly divided into two types: the non-coupling structure mainly comprises parallel control and master-slave control, and in the parallel control structure, the motors have no coupling effect, so that the improvement of the synchronous performance of the motors can only be realized by reducing the tracking error of each motor, and the synchronous performance is poor. The coupling structure comprises cross coupling control, virtual total axis control, offset coupling control and the like. In addition, in order to improve the motor speed control performance, a learner applies a modern control theory such as sliding mode variable structure control, active disturbance rejection control, neural network control, internal model control and the like to a multi-motor speed synchronous control system and obtains remarkable results. The essence of the control structures is to improve the synchronous performance and tracking performance of the multi-motor rotating speed synchronous control system by improving the rotating speed controller or the cross coupling controller of a single motor.
But this cascade-type control structure itself has some limitations: 1) The multiple control loops greatly limit the dynamic response speed of the system; problems such as motion inertia, untimely response and the like can influence the speed synchronization precision and the tracking precision; 2) The cascade control is generally to control a synchronization error which has occurred, and cannot realize advance control before the occurrence of the synchronization error. One important method for achieving predictive control of synchronization errors before they occur and improving their dynamic response speed is model predictive control.
Model predictive control (Model predictive control, MPC) is suitable for use in a multi-motor control system and has the advantage of simplicity and flexibility in design in solving the problems involving complex constraint multivariable control. However, in the existing control set model prediction speed control method, weight coefficient setting is needed, and most of the control set model prediction speed control methods adopt an empirical method or a trial-and-error method, so that the control difficulty is increased, and a great amount of time is consumed in trial and error. Therefore, the research on the non-cascade predictive speed synchronous control method without the weight coefficient setting of the double permanent magnet motor system has important significance.
Disclosure of Invention
The invention aims to improve the dynamic response speed and the synchronous tracking precision of a double permanent magnet motor system, and provides a non-cascade prediction speed synchronous control method without a weight coefficient, which is suitable for the double permanent magnet motor system from the viewpoint of a limited control set. Firstly, based on the idea of unified modeling, combining a motion equation and a voltage equation of a driving motor with a mathematical model of a two-level voltage source inverter to establish a unified model; then predicting a reference voltage vector with current and rotation speed information; in order to solve the problem of weight coefficient setting in the traditional cost function, simplify algorithm complexity, determine an improved cost function according to the ratio of the synchronous error to the tracking error, and select alternative voltage vectors and optimize the cost function. And then considering the actual user requirement, integrating the current limitation into the selection of the alternative voltage vector to realize the limitation of the current. Different from the traditional synchronous control strategy, the non-cascade prediction speed synchronous control method without the weight coefficient setting can improve the dynamic response speed and the synchronous tracking precision of the system, and has the advantages of visual modeling, simple structure and the like.
The technical scheme adopted for realizing the purpose of the invention is as follows:
a non-cascade predictive speed synchronous control method without weight coefficient setting suitable for a double permanent magnet motor system comprises the following steps:
step 1, measuring current and rotor position signals of two motors, and observing the rotating speed and load torque of each motor at the current moment through an extended Kalman filter observer;
step 2, one-step delay compensation: the optimal voltage vector selected at the previous moment acts on the current moment to realize the effect of the optimal voltage vector at the middle moment of the control period, so that the time delay is reduced;
step 3, predicting a reference voltage vector, namely obtaining a predicted reference voltage vector through form transformation according to a discrete voltage equation and a motor motion equation which are obtained after delay compensation;
step 4, calculating the position angle of the reference voltage vector, determining an improved cost function according to the ratio of the contour error to the tracking error, and selecting an optimal voltage vector and corresponding vector acting time according to the cost function;
and 5, judging the relation between the selected optimal voltage vector and the voltage circle through a current limiting module to obtain the voltage vector finally acting on each inverter, wherein the current limiting module converts the maximum current amplitude limit to the voltage vector alpha-beta axis to draw the voltage circle.
In the above technical solution, in the step 3, the reference voltage vector prediction is:
in the formula (1), u di * Is the d-axis reference voltage vector; u (u) qi * A q-axis reference voltage vector; omega i * For a given rotor mechanical angular velocity; psi phi type si * For a given stator flux linkage; j (J) i The motor rotational inertia; t (T) Li Is the load torque; b (B) i Is the viscous friction factor of the motor; omega i Is the mechanical angular velocity of the rotor; r is R si 、L si Respectively a stator resistor and a stator inductor; psi phi type fi Is a permanent magnet flux linkage; omega ei Is the electric angular velocity of the motor rotor, and omega ei ＝p i ω i ，p i Is the pole pair number; k (K) ti Is the torque coefficient of the permanent magnet synchronous motor and K ti ＝1.5p i ψ fi ；T si Is the current sampling period; t (T) smi Is the sampling period of the rotating speed, and T smi ＝10T si The method comprises the steps of carrying out a first treatment on the surface of the k represents the sampling time of the current; t represents the sampling time of the rotation speed, i qi I is the q-axis component of the stator current di Is the d-axis component of the stator current.
In the above technical solution, in the step 4, the cost function is determined by the following method:
when epsilon/e<σ max Epsilon and e represent the magnitudes of the synchronization error and the tracking error, sigma, respectively max For limiting the ratio of synchronization error to tracking error, the tracking performance is mainly considered, the synchronization error is reduced by reducing the tracking error, and an improved cost function is adopted
J＝|u * -u j | (3)
In the formula (3), u j Represents an alternative voltage vector, j=0, 1,2,3,4,5,6,7.
When epsilon/e>σ max At this time, mainly consider the synchronization performance, directly add the synchronization error into the cost function, and adopt the improved cost function as
J＝ε 2 (k+1) (4)
In the above technical scheme, in the step 4, |ε/e|<σ max The optimal voltage vector selection method is as follows:
the effective vector nearest to the reference voltage vector is selected as a first alternative voltage vector, and two vectors adjacent to the first alternative voltage vector and a sum zero vector are selected as a second alternative voltage vector. Table 1 shows the alternative voltage vectors corresponding to each sector, and the action time of the two voltage vectors is calculated by the cost function J minimization principle
Wherein u is i1 For the selected first alternative voltage vector, u i2 For the selected second alternative voltage vector, t i1 For the time of action, t, of the first alternative voltage vector i2 Is the time of action of the second alternative voltage vector.
By u i1 、u i2 Synthesizing an optimal voltage vector according to the corresponding action time: u (u) opt ＝u i1 t i1 +u i2 t i2 。
Table 1 alternative voltage vectors for each sector
θ refi Is the position angle of the reference voltage vector.
Alternatively, in step 4, |ε/e|>σ max The optimal voltage vector selection method is as follows: in order to take the influence of the synchronization error into consideration, the possibility of combining the x-axis voltage vector and the y-axis voltage vector is increased as much as possible, but the method cannot be violated with the target with the minimum tracking error. Thus, the first and second substrates are bonded together,and taking the first alternative voltage vector and the second alternative voltage vector as alternative voltage vectors together so as to calculate and optimize a cost function to determine an optimal voltage vector. And calculating the corresponding current and rotating speed, substituting the current and rotating speed into the cost function for comparison, selecting the alternative voltage vector with the minimum cost function as the optimal voltage vector, and giving the alternative voltage vector corresponding to each sector in the table 2.
Table 2 alternative voltage vectors for each sector
θ refi Is the position angle of the reference voltage vector.
In the above technical solution, in the step 5, the voltage circle drawn by converting the maximum current amplitude limitation to the voltage vector α - β axis is:
[u αi (k+1)+Γ i1 ] 2 +[u βi (k+1)+Γ i2 ] 2 <(L si /T si ·I maxi ) 2
and is also provided with
u αi 、u βi Respectively the alpha and beta axis components of the stator voltage, theta ei For the electrical angle of the motor rotor.
Four cases are discussed in terms of the position of the voltage circle in the α - β coordinate system:
case1: when the formula (6) is satisfied, the selected optimal voltage vector falls within the voltage circle, and the current limit is satisfied at this time, and the optimal voltage vector is not adjusted.
When the formula (6) is not satisfied, the following three cases are classified:
case2: when the origin is located in the circle, the selected optimal voltage vector has an intersection point with the voltage circle, and a new optimal voltage vector u 'is synthesized by the selected voltage vector and the zero vector' opt 。
When the origin is located atWhen the circle is out of the circle, two conditions are adopted, namely Case3 and Case4, the distances from the circle center to the optimal voltage vector are distinguished, and when the distances from the circle center to the optimal voltage vector are smaller than the radius of the voltage circle, the distances from the circle center to the optimal voltage vector are Case3; and when the distance from the center of the circle to the optimal voltage vector is larger than the radius of the voltage circle, the distance is Case4. Radius of voltage circle L si T si ·I maxi At this time, let the expression of the optimal voltage vector in the alpha-beta coordinate system be A i u αi +B i u βi =0, and k i ＝-B i /A i Wherein: a is that i 、B i The ratio coefficients of the alpha axis and the beta axis are respectively, and the center of a voltage circle is (-gamma) i1 ，-Г i2 ) Then the distance from the center of the circle to the optimal voltage vector can be expressed as
Case3: the selected optimal effective voltage vector has two intersection points with the voltage circle, the larger intersection point coordinate is obtained by the method, and a new optimal voltage vector u' is recombined according to the method of Case2 " opti 。
Case4: the selected optimal effective voltage vector has no intersection point with the voltage circle, and in order to meet the current limiting condition, a zero vector is adopted as the optimal voltage vector to act on the whole control period.
Compared with the prior art, the invention has the beneficial effects that:
1. the invention provides a non-cascade prediction speed synchronous control method without weight coefficient tuning, which is suitable for a double permanent magnet motor system and can improve the dynamic response speed and synchronous tracking precision of the system.
2. The invention determines the improved cost function according to the ratio of the synchronous error to the tracking error, selects the optimal voltage vector and calculates the vector acting time, can solve the problem of weight coefficient setting in the traditional cost function, and simplifies the complexity of the algorithm.
3. The invention integrates current limitation into the selection of the alternative voltage vector, realizes the limitation of the current by improving the selected optimal voltage vector, and can solve the problem of overcurrent in an actual system.
Drawings
Fig. 1 is a block diagram of a predictive synchronous control architecture for a dual permanent magnet motor system.
FIG. 2 is a flow chart of a predictive synchronous control algorithm for a dual permanent magnet motor system.
Fig. 3 is a block diagram of a two-level voltage source inverter.
Fig. 4 is a block diagram of a current limiting module.
Detailed Description
The present invention will be described in further detail with reference to specific examples. It should be understood that the specific embodiments described herein are for purposes of illustration only and are not intended to limit the scope of the invention.
Example 1
The invention provides a non-cascade predictive speed synchronous control method without weight coefficient setting, which is applicable to a double permanent magnet motor system from the perspective of a limited control set. Firstly, based on the idea of unified modeling, combining a motion equation and a voltage equation of a driving motor with a mathematical model of a two-level voltage source inverter to establish a unified model; then predicting a reference voltage vector with current and rotation speed information; in order to solve the problem of weight coefficient setting in the traditional cost function, simplify algorithm complexity, determine an improved cost function according to the proportion of synchronous errors and tracking errors, and select an optimal voltage vector and calculate vector acting time. And then considering the actual user requirement, integrating the current limitation into the selection of the alternative voltage vector to realize the limitation of the current. And finally, the dynamic response speed and the synchronous tracking precision of the system can be improved, and the system has the advantages of intuitive modeling, simple structure and the like.
For the non-cascade predictive speed synchronous control strategy of the non-weighting coefficient setting of the double-permanent magnet motor system, the predictive synchronous control structure block diagram of the double-permanent magnet motor system is shown in figure 1. Aiming at a series of problems in the traditional cascade synchronous control method, the invention combines the advantages of intuitive modeling of model predictive control, quick dynamic response and the like, directly adds the synchronous error into a cost function to carry out optimal control, and simultaneously realizes the predictive control of the synchronous error. The method can establish a unified control framework to control two motors, predict reference voltage vectors with current and rotating speed information, determine improved cost functions according to the ratio of synchronous errors to tracking errors, and select optimal voltage vectors and calculate vector acting time. And then considering the actual user demand, integrating the current limitation into the selection of the alternative voltage vector, and realizing the limitation of the current by improving the selected optimal voltage vector.
The flow chart of the predictive and synchronous control algorithm of the double permanent magnet motor system is shown in fig. 2. Mainly comprises the following 5 steps:
1) Stator current is measured and rotational speed and load torque are observed by an extended kalman filter observer.
2) One step delay compensation.
3) And (5) predicting a reference voltage vector.
4) And calculating the position of the reference voltage vector, and selecting the optimal voltage vector and the corresponding acting time.
5) And judging the relation between the selected optimal voltage vector and the voltage circle through the current limiting module to obtain the voltage vector finally acting on the inverter.
Example 2
For a non-cascade predictive speed synchronous control method for the non-weighting coefficient setting of a double permanent magnet motor system, the predictive synchronous controller of the double permanent magnet motor system is specifically as follows:
1. mathematical modeling of a double permanent magnet synchronous motor system:
the motion equation and the voltage equation of the permanent magnet synchronous motor are as follows
Wherein J is i The motor rotational inertia; t (T) ei Is electromagnetic torque; t (T) Li Is the load torque; τ i Is the driving torque; b (B) i Is the viscous friction factor of the motor; omega i Is the mechanical angular velocity of the rotor; r is R si 、L qi And L di Respectively a stator resistance, a quadrature axis inductance and a direct axis inductance; u (u) di 、u qi 、i di And i qi D, q-axis components of the stator voltage and stator current, respectively; psi phi type fi Is a permanent magnet flux linkage; omega ei Is the electric angular velocity of the motor rotor, and omega ei ＝p i ω i ，p i Is polar logarithmic.
The mathematical model of the two-level voltage source inverter is as follows:
the two-level voltage source inverter comprises 2 3 Basic voltage vectors (6 effective vectors V) 1 -V 6 And two zero vectors V 0 ，V 7 )。
In d-q coordinate system, use u i Represents the output voltage vector, and has
In U dci A DC side voltage of 2L-VSI; t (T) i For rotating the transformation matrix, S i Is a switch state column vector, and
S i ＝[S i1 S i3 S i5 ] T (5)
wherein S is i1 、S i3 、S i5 And the switch state of the upper bridge arm of the 2L-VSI is shown.
2. Design of a predictive synchronous controller of a double permanent magnet motor system:
(1) One-step delay compensation:
in order to improve the current prediction precision, the formula (2) is expressed by a state equation, and an improved two-step Euler integral method is adopted, wherein a discretization model is formed by
Wherein T is si Is the sampling period.
(2) Reference voltage vector prediction:
obtaining q-axis current reference value through motion equationIs that
Wherein T is smi Is the sampling period of the rotating speed, and T smi ＝10T si ，T si Is the current sampling period; t represents the sampling time of the rotation speed, and omega i (t+1) represents a rotational speed predicted value.
Substituting equation (7) into the discretized voltage equation, and obtaining the predicted reference voltage vector by transformation form as
Wherein u is di * Is the d-axis reference voltage vector; u (u) qi * A q-axis reference voltage vector; omega i * For a given rotor mechanical angular velocity; psi phi type si * For a given stator flux linkage; k represents the sampling instant of the current.
(3) Determination of a cost function:
discussion of two cases:
1) When epsilon/e<σ max Epsilon and e represent the magnitudes of the synchronization error and the tracking error, sigma, respectively max For limiting the ratio of synchronization error to tracking error, the tracking performance is mainly considered, the synchronization error is reduced by reducing the tracking error, and an improved cost function is adopted
J＝|u * -u j | (10)
Wherein u is j Represents an alternative voltage vector, j=0, 1,2,3,4,5,6,7.
2) When epsilon/e>σ max At this time, mainly consider the synchronization performance, directly add the synchronization error into the cost function, and adopt the improved cost function as
J＝ε 2 (k+1) (11)
(4) Selection of an optimal voltage vector:
obtaining a reference voltage vector d, q-axis component by the method (8)And->To compare the reference voltage vector with 8 base voltage vectors, the d-q axis reference voltage vector is converted to the alpha-beta axis as
Thus, the reference voltage vector position angle θ is obtained refi Is that
According to the obtained reference voltage vector position angle, the sector to which the reference voltage vector belongs is determined by combining the partition structure diagram of the two-level voltage source inverter as shown in fig. 3.
The selection of the optimal voltage vector is discussed in two cases:
first case:
when epsilon/e<σ max When the effective vector closest to the reference voltage vector is selected as the first voltage vector, two vectors adjacent to the first voltage vector and the sum zero vector are selected as the second alternative vector. Table 1 shows the alternative voltage vectors corresponding to each sector, by the cost function minimization principle, toObtaining the action time of two voltage vectors as
Wherein u is i1 For the selected first alternative voltage vector, u i2 For the selected second alternative voltage vector, t i1 For the time of action, t, of the first alternative voltage vector i2 Is the time of action of the second alternative voltage vector. Specific reference may be made to X G.Zhang, B.S.Hou.double Vectors Model Predictive Torque Control Without Weighting Factor Based on Voltage Tracking Error [ J]IEEE Trans.Power Electron, 2018,33 (3): 2368-2380. Will not be described in detail herein.
By u i1 、u i2 And corresponding time of action to synthesize an optimal voltage vector, i.e., u opt ＝u i1 t i1 +u i2 t i2 。
Table 1 alternative voltage vectors for each sector
Second case:
when epsilon/e>σ max In order to take the influence of the synchronization error into consideration, the possibility of combining the x-axis voltage vector and the y-axis voltage vector is increased as much as possible, but the method cannot be violated with the target with the minimum tracking error. Thus, the first vector and the second vector selected previously are taken together as the candidate voltage vector, and calculatedAnd substituting corresponding current and rotating speed information into the cost function for comparison, and selecting an alternative voltage vector with the minimum cost function as an optimal voltage vector. Table 2 gives the alternative voltage vectors for each sector.
Table 2 alternative voltage vectors for each sector
(5) A current limiting module:
the maximum current amplitude limit is converted to the alpha-beta axis of the voltage vector, namely, the current limit is integrated into the selection of alternative voltage vectors, and the current limit is realized by improving the selected optimal voltage vector. The voltage circles drawn on the voltage vector alpha-beta axis are:
[u αi (k+1)+Γ i1 ] 2 +[u βi (k+1)+Γ i2 ] 2 <(L si /T si ·I maxi ) 2
and is also provided with
u αi 、u βi Respectively the alpha and beta axis components of the stator voltage, theta ei For the electrical angle of the motor rotor.
Four cases are discussed in terms of the position of the voltage circle in the α - β coordinate system, as shown in fig. 4.
Case1: when the expression (15) is satisfied, the selected optimal voltage vector falls within a voltage circle, i.e., a circle represented by a green line in the figure, and at this time, the current limit is satisfied, and the optimal voltage vector is not adjusted.
When the formula (15) is not satisfied, the following three cases are classified:
case2: when the origin is located in the circle, i.e. the circle represented by the blue line in the figure, the selected optimal voltage vector has an intersection with the voltage circle, and a new optimal voltage vector u 'is synthesized by the selected voltage vector and the zero vector' opt 。
When the origin is located outside the circle, there are two cases, namely, a circle (Case 3) represented by a pink line in the figure and a circle (Case 4) represented by an orange line in the figure, at this time, the two cases are distinguished by the distance from the center of the circle to the optimal voltage vector, and when the distance from the center of the circle to the optimal voltage vector is smaller than the radius of the circle, the two cases are Case3; and when the distance from the circle center to the optimal voltage vector is larger than the radius of the circle, the distance is Case4. At this time, let the expression of the optimal voltage vector in the alpha-beta coordinate system be A i u αi +B i u βi =0, and k i ＝-B i /A i The center of the voltage circle is (-f) i1 ，-Г i2 ) Then the distance from the center of the circle to the optimal voltage vector can be expressed as
Case3: the selected optimal effective voltage vector has two intersection points with the voltage circle, the larger intersection point coordinate is obtained by the method, and a new optimal voltage vector u' is recombined according to the method of Case2 " opti 。
Case4: the selected optimal effective voltage vector has no intersection point with the voltage circle, and in order to meet the current limiting condition, a zero vector is adopted as the optimal voltage vector to act on the whole control period.
The foregoing is merely a preferred embodiment of the present invention and it should be noted that modifications and adaptations to those skilled in the art may be made without departing from the principles of the present invention, which are intended to be comprehended within the scope of the present invention.
Claims (9)
1. The non-cascade predictive speed synchronous control method without weight coefficient setting suitable for the double permanent magnet motor system is characterized by comprising the following steps of:
step 1, measuring current and rotor position signals of two motors, and observing the rotating speed and load torque of each motor at the current moment through an extended Kalman filter observer;
step 2, one-step delay compensation: the optimal voltage vector selected at the last moment acts on the current moment;
step 3, reference voltage vector prediction: obtaining a predicted reference voltage vector through form transformation according to a discrete voltage equation and a motor motion equation which are obtained after delay compensation;
step 4, calculating the position angle of the reference voltage vector, determining an improved cost function according to the ratio of the contour error to the tracking error, and selecting an optimal voltage vector and corresponding vector acting time according to the cost function;
step 5, the selected optimal voltage vector passes through a current limiting module, and the relation between the selected optimal voltage vector and a voltage circle is judged to obtain the voltage vector which finally acts on each inverter, wherein the current limiting module converts the maximum current amplitude limit to the voltage vector alpha-beta axis to draw the voltage circle;
in the step 3, the reference voltage vector prediction is:
in the formula (1), u di * Is the d-axis reference voltage vector; u (u) qi * A q-axis reference voltage vector; omega i * For a given rotor mechanical angular velocity; psi phi type si * For a given stator flux linkage; j (J) i The motor rotational inertia; t (T) Li Is the load torque; b (B) i Is the viscous friction factor of the motor; omega i Is the mechanical angular velocity of the rotor; r is R si 、L si Respectively a stator resistor and a stator inductor; psi phi type fi Is a permanent magnet flux linkage; omega ei Is the electric angular velocity of the motor rotor, and omega ei ＝p i ω i ，p i Is the pole pair number; k (K) ti Is the torque coefficient of the permanent magnet synchronous motor and K ti ＝1.5p i ψ fi ；T si Is the current sampling period; t (T) smi Is the sampling period of the rotating speed, and T smi ＝10T si The method comprises the steps of carrying out a first treatment on the surface of the k represents the sampling time of the current; t represents the sampling time of the rotation speed, i qi I is the q-axis component of the stator current di Is the d-axis component of the stator current.
2. The non-cascade predictive speed synchronization control method without weighting coefficient tuning of claim 1, wherein in said step 4, the cost function is determined by:
epsilon and e represent the magnitudes of the synchronization error and the tracking error, sigma, respectively max A limit value which is the ratio of the synchronization error to the tracking error:
when epsilon/e<σ max In the time-course of which the first and second contact surfaces,
J＝|u * -u j | (3)
in the formula (3), u j Represents an alternative voltage vector, j=0, 1,2,3,4,5,6,7;
when epsilon/e>σ max In the time-course of which the first and second contact surfaces,
J＝ε 2 (k+1) (4)。
3. the non-cascade predictive speed synchronization control method without weighting factor tuning of claim 2, wherein in said step 4, when |ε/e|<σ max The optimal voltage vector selection method is as follows:
selecting an effective vector nearest to the reference voltage vector as a first alternative voltage vector, selecting two vectors adjacent to the first alternative voltage vector and a zero vector as a second alternative voltage vector, and obtaining the action time of the two voltage vectors as a second alternative voltage vector through a cost function J minimization principle
Wherein u is i1 For the first selected alternative voltage vector,u i2 for the selected second alternative voltage vector, t i1 For the time of action, t, of the first alternative voltage vector i2 The time of action of the second alternative voltage vector;
by u i1 、u i2 Synthesizing an optimal voltage vector according to the corresponding action time: u (u) opt ＝u i1 t i1 +u i2 t i2 。
4. The non-cascade predictive speed synchronization control method without weighting factor tuning of claim 3, wherein the candidate voltage vectors corresponding to each sector are as shown in table 1:
table 1 alternative voltage vectors for each sector
θ refi Is the position angle of the reference voltage vector.
5. The non-cascade predictive speed synchronization control method without weighting factor tuning of claim 2, wherein in said step 4, when |ε/e|>σ max The optimal voltage vector selection method is as follows:
selecting a valid vector nearest to the reference voltage vector as a first alternative voltage vector, and selecting two vectors adjacent to the first alternative voltage vector and a sum zero vector as a second alternative voltage vector;
the first alternative voltage vector and the second alternative voltage vector are taken as alternative voltage vectors together, corresponding current and rotating speed are calculated and substituted into the cost function for comparison, and the alternative voltage vector with the smallest cost function is selected as the optimal voltage vector.
6. The non-cascade predictive speed synchronization control method without weighting coefficient setting of claim 5, wherein the candidate voltage vectors corresponding to each sector are as shown in table 2:
table 2 alternative voltage vectors for each sector
θ refi Is the position angle of the reference voltage vector.
7. The non-cascade predictive speed synchronization control method without weighting coefficient setting as set forth in claim 4 or 6, wherein a position angle of the reference voltage vector is calculated by:
calculating a reference voltage vector d, q-axis component from a reference voltage vectorAnd->Converting d-q axis reference voltage vector to alpha-beta axis as
Thus, the reference voltage vector position angle θ is obtained refi Is that
8. The non-cascade predictive speed synchronization control method without weighting coefficient setting according to claim 1, wherein the voltage circle drawn by converting the current maximum amplitude limitation to the voltage vector α - β axis in the step 5 is:
u αi 、u βi respectively the alpha and beta axis components of the stator voltage, theta ei For the electrical angle of the motor rotor.
9. The non-cascade predictive speed synchronization control method without weighting coefficient setting of claim 8, wherein the position of the voltage circle in the α - β coordinate system is discussed in four cases:
case1: when the formula (6) is satisfied, the optimal voltage vector is not adjusted;
when the formula (6) is not satisfied, the following three types are classified:
case2: when the origin is located in the circle, the optimal voltage vector and the voltage circle have an intersection point, and a new optimal voltage vector u 'is synthesized by the optimal voltage vector and the zero vector' opt ；
When the origin is located outside the circle, distinguishing the origin by the distance from the circle center to the optimal voltage vector, and when the distance from the circle center to the optimal voltage vector is smaller than the radius of the voltage circle, determining the origin as Case3; when the distance from the center of the circle to the optimal voltage vector is larger than the radius of the voltage circle, the distance is Case4;
radius of voltage circle L si /T si ·I maxi Let the expression of the optimal voltage vector in the alpha-beta coordinate system be A i u αi +B i u βi =0, and k i ＝-B i /A i Wherein: a is that i 、B i The ratio coefficients of the alpha axis and the beta axis are respectively, and the center of a voltage circle is (-gamma) i1 ，-Г i2 ) The distance from the center of the circle to the optimal voltage vector is expressed as
Case3: the selected optimal effective voltage vector has two intersection points with the voltage circle, the larger intersection point coordinate is obtained by the method, and a new optimal voltage vector u' is recombined according to the method of Case2 " opti ；
Case4: the selected optimal effective voltage vector has no intersection point with the voltage circle, and a zero vector is adopted as the optimal voltage vector to act on the whole control period.
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