CN113972877B - Simplified permanent magnet synchronous motor model prediction current control method - Google Patents
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
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Abstract
The invention discloses a simplified permanent magnet synchronous motor model prediction current control method. The conventional model predictive current control requires a cost function to evaluate the voltage vector corresponding to each switching state of the inverter, resulting in complex system calculation. Furthermore, only one vector acts during one control period, making the steady state performance of the system poor. In order to solve the problems, the invention only needs to evaluate three non-adjacent non-zero vectors by using a cost function, and determines two non-zero vectors in a complete control set according to the magnitude relation of the cost function corresponding to the three non-adjacent non-zero vectors. The final action vector can be obtained by calculating the two non-zero vectors through the current dead beat principle twice, and the range of the final action vector covers the whole hexagon. Compared with the traditional model predictive current control, the invention effectively reduces the calculation load and improves the steady-state performance of the system.
Description
Technical Field
The invention relates to the field of model predictive current control of permanent magnet synchronous motors, in particular to a simplified model predictive current control method. The method is beneficial to reducing the calculation load of the system and improving the steady-state performance of the system.
Background
The permanent magnet synchronous motor has the advantages of high efficiency, small volume, simple structure and the like, and is applied to the fields of transportation, aerospace, national defense and the like.
At present, the traditional vector control and direct torque control are mainly adopted in the field of drive control of a permanent magnet synchronous motor. Vector control involves complex coordinate transformations and PI parameters of the speed outer loop and the current inner loop require complex tuning procedures, whereas direct torque control, while responding quickly to torque changes, has poor steady state performance. With the continuous enhancement of the computing power of modern microprocessors, model predictive control is beginning to be applied to the field of permanent magnet synchronous motor control.
The model predictive control has the advantages of simple control structure, capability of combining different constraint conditions and the like. The model predictive control of the permanent magnet synchronous motor can be divided into model predictive torque control and model predictive current control. The model predictive current control of the permanent magnet synchronous motor utilizes the discretization characteristic of the output vector of the inverter, calculates the predictive current value of the corresponding vector of each inverter switching state, then respectively brings the predictive current value into a preset cost function, and selects the switching state corresponding to the minimum value of the cost function as the control quantity of the next period of the inverter. This places a significant computational burden on the system, since the current value for each vector needs to be predicted and taken into the cost function calculation during the selection of the vector. Only one vector is applied per control period, which makes the error in current tracking large, resulting in poor steady state performance of the system.
Disclosure of Invention
Aiming at the two problems of large calculation amount and poor steady-state performance of the permanent magnet synchronous motor model predictive control, the invention provides a simplified permanent magnet synchronous motor model predictive control method. Two non-zero vectors are selected in the complete control set by comparing the magnitudes of the corresponding cost functions of the three non-adjacent non-zero vectors. The two non-zero vectors are calculated through the current dead beat principle twice, so that the final action vector with the range covering the whole hexagon can be obtained.
In order to achieve the aim of the invention, the invention adopts the following technical scheme:
The simplified permanent magnet synchronous motor model prediction current control method comprises the following steps:
step 1: firstly, deducing a mathematical model and a discretized predictive current control model of a permanent magnet synchronous motor under a rotating coordinate system;
step 2: secondly, calculating cost functions corresponding to three non-adjacent non-zero vectors respectively, and selecting two non-zero vectors in the complete control set according to the magnitude relation of the cost functions;
step 3: and (3) calculating the action time of the two non-zero vectors and the action time of the zero vector obtained in the step (2) according to the principle of dead beat of the current twice, so that the final action vector range can cover the whole hexagon.
Further, the specific process of the step 1 is as follows:
the voltage equation of the permanent magnet synchronous motor under the dq coordinate system is as follows:
Wherein u d and u q are voltages of the motor in the dq coordinate system respectively; i d and i q are currents in dq coordinate system, respectively; l d and L q are inductances in dq coordinate system respectively; phi f is the permanent magnet flux linkage amplitude; r is motor phase resistance; omega e is the electrical frequency of the motor;
the state equation of a permanent magnet synchronous motor can be expressed as:
using a forward differential discretization formula, the current equation of the permanent magnet synchronous motor is:
Wherein i d (k) and i q (k) are currents in a dq coordinate system at the time of k respectively; i d (k+1) and i q (k+1) are currents in the dq coordinate system at the time of k+1, respectively; u d (k) and u q (k) are voltages in the dq coordinate system at time k, respectively; e d (k) and e q (k) are respectively estimated values of back electromotive force at k time in dq coordinate system, and T s is sampling time.
Further, the specific process of step 2 is as follows:
The basic idea of the permanent magnet synchronous motor model predictive current control is to select an optimal voltage vector through a cost function, and then take the switching state of the optimal voltage vector as the control quantity of an inverter, wherein the cost function g of the permanent magnet synchronous motor model predictive current control can be expressed as:
Wherein i d * and i q * are respectively reference currents under a dq coordinate system, d-axis reference currents are set to be zero, and q-axis reference currents are obtained by a PI controller;
The predicted current values of three non-adjacent non-zero vectors at the time k+1 of u 1、u3、u5 are calculated respectively, wherein u 1 represents a vector corresponding to the inverter switching state of 100, u 3 represents a vector corresponding to the inverter switching state of 010, u 5 represents a vector corresponding to the inverter switching state of 001, and the components of the predicted current values of the three vectors under the dq coordinate system can be expressed as:
Wherein i d 1(k+1)、iq 1 (k+1) represents the components of the predicted current value of the non-zero vector u 1 at the time of k+1 in the dq coordinate system, i d 3(k+1)、iq 3 (k+1) represents the components of the predicted current value of the non-zero vector u 3 at the time of k+1 in the dq coordinate system, and i d 5(k+1)、iq 5 (k+1) represents the components of the predicted current value of the non-zero vector u 5 at the time of k+1 in the dq coordinate system, respectively; u d 1、uq 1 represents the component of the non-zero vector u 1 in the dq coordinate system, u d 3、uq 3 represents the component of the non-zero vector u 3 in the dq coordinate system, and u d 5、uq 5 represents the component of the non-zero vector u 5 in the dq coordinate system;
The predicted current values of the three non-adjacent non-zero vectors of u 1、u3、u5 at the time k+1 are brought into a cost function calculation, so that three different cost function values of g 1、g3、g5 can be obtained; wherein g 1 represents a value of a value calculated by substituting a predicted current value corresponding to the non-zero vector u 1 into the value function, g 3 represents a value of a value calculated by substituting a predicted current value corresponding to the non-zero vector u 3 into the value function, and g 5 represents a value of a value calculated by substituting a predicted current value corresponding to the non-zero vector u 5 into the value function;
Comparing the magnitudes of the three cost function values, if g 1＜g3＜g5, selecting a vector u 1、u2 in the control set; if g 1＜g5＜g3, then vector u 1、u6 is selected in the control set; if g 3＜g1＜g5, then vector u 2、u3 is selected in the control set; if g 3＜g5＜g1, then vector u 3、u4 is selected in the control set; if g 5＜g1＜g3, then vector u 5、u6 is selected in the control set; if g 5＜g3＜g1, then vector u 4、u5 is selected in the control set; where u 2 represents a vector corresponding to an inverter switching state of 110, u 4 represents a vector corresponding to an inverter switching state of 011, and u 6 represents a vector corresponding to an inverter switching state of 101.
Further, the specific steps of step 3 are as follows:
according to the current dead beat principle, the current i s (k+1) at the end of one control period can be expressed as:
is(k+1)＝is(k)+sid1Ts+sj(1-d1)Ts
Wherein i s (k) represents the current value at time k, i s (k+1) represents the current value at time k+1, u i and u j represent the two non-zero vectors selected in step 2, s i represents the current change rate of the motor under the action of the non-zero vector u i, s j represents the current change rate of the motor under the action of the non-zero vector u j, d 1 is the proportion of the action time of u i in one control period to the whole control period, and 1-d 1 represents the proportion of the action time of u j in one control period to the whole control period; the solution for d 1 can be expressed as:
Wherein i s * denotes a reference current value;
The resultant vector of u i and u j after being allocated by a certain duty cycle can be expressed as:
usyn＝d1×ui+(1-d1)×uj
After the primary current dead beat calculation, the range of the synthesized vector is limited to the connecting line of the two vectors, in order to further expand the range of the acting vector, the primary current dead beat calculation is performed again, the range of the acting vector is expanded by introducing a zero vector, and the secondary current dead beat calculation can be expressed as:
is(k+1)＝is(k)+ssynd2Ts+s0(1-d2)Ts＝is *
where s syn represents the current change rate of the motor under the action of the composite vector u syn, s 0 represents the current change rate of the motor under the action of the zero vector, d 2 is the proportion of the action time of the composite vector u syn in one control period to the whole control period, and 1-d 2 represents the proportion of the action time of the zero vector in one control period to the whole control period, and the solution of d 2 can be expressed as follows:
The resultant vector of u syn and u 0、u7 after being allocated by a certain duty cycle can be expressed as:
Where u applied represents a vector that is ultimately applied to the motor by the inverter, u 0 represents a vector corresponding to an inverter switching state of 000, and u 7 represents a vector corresponding to an inverter switching state of 111. Through the dead beat calculation of the current for two times, the final action vector u applied is synthesized by two non-zero vectors and zero vectors through a certain proportion, the range of the final action vector can cover the whole hexagonal area, the current tracking error is greatly reduced, and the steady state performance of the system is improved.
The beneficial effects of the invention are as follows:
1. By comparing the magnitudes of the corresponding cost functions of the three non-adjacent non-zero vectors, the invention can determine the two non-zero vectors in the complete control set, avoids evaluating the vectors corresponding to all the switch states, and reduces the calculated amount of the system.
2. According to the invention, through the calculation of the dead beat principle of twice currents, the final acting vector is synthesized by two non-zero vectors and a zero vector, the vector range covers the whole hexagon, and the steady-state performance of the system is improved.
3. The invention can further improve the competitiveness of model predictive current control in the field of permanent magnet synchronous motor control.
Drawings
FIG. 1 is a simplified block diagram of a permanent magnet synchronous motor model predictive current control method;
FIG. 2 is a schematic diagram of three non-adjacent non-zero vectors;
FIG. 3 is a schematic view of the range of action vectors; wherein, (a) is a primary current dead beat post action vector range and (b) is a secondary current dead beat post action vector range;
FIG. 4 is a schematic diagram of a three-phase current waveform; wherein, (a) predicts three-phase current waveform for traditional model, and (b) predicts three-phase current waveform for the invention;
FIG. 5 is a schematic diagram of phase current harmonic content; wherein, (a) is the current harmonic content of the traditional model predictive current control phase and (b) is the current harmonic content of the phase of the invention.
Detailed description of the preferred embodiments
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention.
Embodiments of the present invention are described in detail below, examples of which are illustrated in the accompanying drawings, wherein like or similar reference numerals refer to like or similar elements or elements having like or similar functions throughout. The embodiments described below by referring to the drawings are illustrative only and are not to be construed as limiting the invention.
As shown in the structural block diagram of FIG. 1, the invention is a simplified permanent magnet synchronous motor model prediction current control method, which mainly comprises a high-efficiency vector selection method and a method for reducing current tracking errors by expanding the range of action vectors.
The invention takes a three-phase permanent magnet synchronous motor as a control object, and carries out simplified model prediction current control on the three-phase permanent magnet synchronous motor, and the specific measures are as follows:
and step 1, deducing a mathematical model and a discretized predictive current control model of the permanent magnet synchronous motor.
The voltage equation of the permanent magnet synchronous motor under the dq coordinate system is as follows:
Wherein u d and u q are voltages of the motor in the dq coordinate system respectively; i d and i q are currents in dq coordinate system, respectively; l d and L q are inductances in dq coordinate system respectively; phi f is the permanent magnet flux linkage amplitude; r is motor phase resistance; omega e is the electrical frequency of the motor.
The state equation of a permanent magnet synchronous motor can be expressed as:
using a forward differential discretization formula, the current equation of the permanent magnet synchronous motor is:
Wherein i d (k) and i q (k) are currents in a dq coordinate system at the time of k respectively; i d (k+1) and i q (k+1) are currents in the dq coordinate system at the time of k+1, respectively; u d (k) and u q (k) are voltages in the dq coordinate system at time k, respectively; e d (k) and e q (k) are respectively estimated values of back electromotive force at k time in dq coordinate system, and T s is sampling time.
And 2, selecting two non-zero vectors in the complete control set according to the magnitude relation of the corresponding cost functions of the three non-adjacent non-zero vectors.
The basic idea of the permanent magnet synchronous motor model predictive current control is to select an optimal voltage vector through a cost function, and then take the switching state of the optimal voltage vector as the control quantity of the inverter. The cost function g of the permanent magnet synchronous motor model predictive current control can be expressed as:
Wherein i d * and i q * are respectively reference currents in dq coordinate system, d-axis reference current is set to zero, q-axis reference current is obtained by PI controller.
The predicted current values at time k +1 for the three non-adjacent non-zero vectors of u 1、u3、u5 are calculated separately, as shown in figure 2. The components id 1、iq 1、id 3、iq 3、id 5、iq 5 of the predicted current values of these three vectors in the dq coordinate system can be expressed as:
The predicted current values of the three non-adjacent non-zero vectors of u 1、u3、u5 at the time k+1 are brought into a cost function calculation to obtain three different cost function values of g 1、g3、g5.
The magnitudes of the three cost function values are compared. If g 1＜g3＜g5, then vector u 1、u2 is selected in the control set. If g 1＜g5＜g3, then vector u 1、u6 is selected in the control set. If g 3＜g1＜g5, then vector u 2、u3 is selected in the control set. If g 3＜g5＜g1, then vector u 3、u4 is selected in the control set. If g 5＜g1＜g3, then vector u 5、u6 is selected in the control set. If g 5＜g3＜g1, then vector u 4、u5 is selected in the control set.
And 3, calculating the action time of the two non-zero vectors and the action time of the zero vector obtained in the step 2 according to the principle of dead beat of the two currents, so that the final action vector range can cover the whole hexagon.
According to the current dead beat principle, the current i s (k+1) at the end of one control period can be expressed as:
is(k+1)＝is(k)+sid1Ts+sj(1-d1)Ts
Where u i and u j represent the two non-zero vectors selected in step 2. d 1 is the ratio of the active time of u i in one control period to the whole control period, and 1-d 1 are the ratios of u j in one control period to the whole control period. The solution for d 1 can be expressed as:
The resultant vector of u i and u j after being allocated by a certain duty cycle can be expressed as:
usyn＝d1×ui+(1-d1)×uj
After one-time current dead beat calculation, the range of the synthesized vector is limited to the two-vector line, as shown in fig. 3 (a). In order to further expand the range of the acting vector, the current dead beat calculation is performed once again, and the range of the acting vector is expanded by introducing a zero vector, and the second current dead beat calculation can be expressed as:
is(k+1)＝is(k)+ssynd2Ts+s0(1-d2)Ts＝is *
d 2 is the ratio of the active time of u syn in one control period to the whole control period, and 1-d 2 represent the ratio of the active time of zero vector in one control period to the whole control period. The solution for d 2 can be expressed as:
The resultant vector of u syn and u 0、u7 after being allocated by a certain duty cycle can be expressed as:
Where u applied represents the vector that is ultimately applied to the motor by the inverter, and after the dead beat calculation of the second current, the resultant vector range is extended into the triangle of the two non-zero vectors and their wiring, as shown in fig. 3 (b). As the combination of the two non-zero vectors has six conditions, the range of the final action vector can cover the whole hexagonal area, the current tracking error is greatly reduced, and the steady-state performance of the system is improved.
FIG. 4 (a) is a graph of a conventional model predictive current control three-phase current waveform with a THD of 3.11% for the phase current, as shown in FIG. 5 (a); fig. 4 (b) is a three-phase current waveform diagram of the present invention, and the THD of the phase current is 0.87%, as shown in fig. 5 (b). The simplified model prediction current control current tracking performance is good, and the steady state performance of the system is improved.
In the description of the present specification, reference to the terms "one embodiment," "some embodiments," "illustrative embodiments," "examples," "specific examples," or "some examples," etc., means that a particular feature, structure, material, or characteristic described in connection with the embodiment or example is included in at least one embodiment or example of the invention. In this specification, schematic representations of the above terms do not necessarily refer to the same embodiments or examples. Furthermore, the particular features, structures, materials, or characteristics described may be combined in any suitable manner in any one or more embodiments or examples.
While embodiments of the present invention have been shown and described, it will be understood by those of ordinary skill in the art that: many changes, modifications, substitutions and variations may be made to the embodiments without departing from the spirit and principles of the invention, the scope of which is defined by the claims and their equivalents.
Claims (3)
1. The simplified permanent magnet synchronous motor model prediction current control method is characterized by comprising the following steps of:
step 1: firstly, deducing a mathematical model and a discretized predictive current control model of a permanent magnet synchronous motor under a rotating coordinate system;
step 2: secondly, calculating cost functions corresponding to three non-adjacent non-zero vectors respectively, and selecting two non-zero vectors in the complete control set according to the magnitude relation of the cost functions;
step 3: according to the principle of dead beat of the current for two times, calculating the action time of the two non-zero vectors and the action time of the zero vector obtained in the step 2, so that the final action vector range can cover the whole hexagon;
The specific process of the step 2 is as follows:
The basic idea of the permanent magnet synchronous motor model predictive current control is to select an optimal voltage vector through a cost function, and then take the switching state of the optimal voltage vector as the control quantity of an inverter, wherein the cost function g of the permanent magnet synchronous motor model predictive current control can be expressed as:
Wherein i d * and i q * are respectively reference currents under a dq coordinate system, d-axis reference currents are set to be zero, and q-axis reference currents are obtained by a PI controller;
respectively calculating predicted current values of three non-adjacent non-zero vectors of u 1、u3、u5 at the time k+1, wherein u 1 represents a vector corresponding to the inverter switching state of 100, u 3 represents a vector corresponding to the inverter switching state of 010, and u 5 represents a vector corresponding to the inverter switching state of 001;
The predicted current values of the three non-adjacent non-zero vectors of u 1、u3、u5 at the time k+1 are brought into a cost function calculation, so that three different cost function values of g 1、g3、g5 can be obtained; wherein g 1 represents a value of a value calculated by substituting a predicted current value corresponding to the non-zero vector u 1 into the value function, g 3 represents a value of a value calculated by substituting a predicted current value corresponding to the non-zero vector u 3 into the value function, and g 5 represents a value of a value calculated by substituting a predicted current value corresponding to the non-zero vector u 5 into the value function;
Comparing the magnitudes of the three cost function values, if g 1＜g3＜g5, selecting a vector u 1、u2 in the control set; if g 1＜g5＜g3, then vector u 1、u6 is selected in the control set; if g 3＜g1＜g5, then vector u 2、u3 is selected in the control set; if g 3＜g5＜g1, then vector u 3、u4 is selected in the control set; if g 5＜g1＜g3, then vector u 5、u6 is selected in the control set; if g 5＜g3＜g1, then vector u 4、u5 is selected in the control set; where u 2 represents a vector corresponding to an inverter switching state of 110, u 4 represents a vector corresponding to an inverter switching state of 011, and u 6 represents a vector corresponding to an inverter switching state of 101;
The components of the predicted current values of three non-adjacent non-zero vectors in the dq coordinate system can be expressed as:
Wherein i d 1(k+1)、iq 1 (k+1) represents the components of the predicted current value of the non-zero vector u 1 at the time of k+1 in the dq coordinate system, i d 3(k+1)、iq 3 (k+1) represents the components of the predicted current value of the non-zero vector u 3 at the time of k+1 in the dq coordinate system, and i d 5(k+1)、iq 5 (k+1) represents the components of the predicted current value of the non-zero vector u 5 at the time of k+1 in the dq coordinate system, respectively; u d 1、uq 1 represents the component of the non-zero vector u 1 in the dq coordinate system, u d 3、uq 3 represents the component of the non-zero vector u 3 in the dq coordinate system, and u d 5、uq 5 represents the component of the non-zero vector u 5 in the dq coordinate system;
the specific steps of the step 3 are as follows:
according to the current dead beat principle, the current i s (k+1) at the end of one control period can be expressed as:
is(k+1)＝is(k)+sid1Ts+sj(1-d1)Ts
Wherein i s (k) represents the current value at time k, i s (k+1) represents the current value at time k+1, u i and u j represent the two non-zero vectors selected in step 2, s i represents the current change rate of the motor under the action of the non-zero vector u i, s j represents the current change rate of the motor under the action of the non-zero vector u j, d 1 is the proportion of the action time of u i in one control period to the whole control period, and 1-d 1 represents the proportion of the action time of u j in one control period to the whole control period; the solution for d 1 can be expressed as:
Wherein i s * denotes a reference current value;
The resultant vector of u i and u j after being allocated by a certain duty cycle can be expressed as:
usyn＝d1×ui+(1-d1)×uj
After the primary current dead beat calculation, the range of the synthesized vector is limited to the connecting line of the two vectors, in order to further expand the range of the acting vector, the primary current dead beat calculation is performed again, the range of the acting vector is expanded by introducing a zero vector, and the secondary current dead beat calculation can be expressed as:
is(k+1)＝is(k)+ssynd2Ts+s0(1-d2)Ts＝is*
where s syn represents the current change rate of the motor under the action of the composite vector u syn, s 0 represents the current change rate of the motor under the action of the zero vector, d 2 is the proportion of the action time of the composite vector u syn in one control period to the whole control period, and 1-d 2 represents the proportion of the action time of the zero vector in one control period to the whole control period, and the solution of d 2 can be expressed as follows:
The resultant vector of u syn and u 0、u7 after being allocated by a certain duty cycle can be expressed as:
Where u applied represents a vector that is ultimately applied to the motor by the inverter, u 0 represents a vector corresponding to an inverter switching state of 000, and u 7 represents a vector corresponding to an inverter switching state of 111.
2. The simplified permanent magnet synchronous motor model predictive current control method according to claim 1, wherein the specific process of step 1 is as follows:
the voltage equation of the permanent magnet synchronous motor under the dq coordinate system is as follows:
Wherein u d and u q are voltages of the motor in the dq coordinate system respectively; i d and i q are currents in dq coordinate system, respectively; l d and L q are inductances in dq coordinate system respectively; phi f is the permanent magnet flux linkage amplitude; r is motor phase resistance; omega e is the electrical frequency of the motor;
the state equation of a permanent magnet synchronous motor can be expressed as:
using a forward differential discretization formula, the current equation of the permanent magnet synchronous motor is:
Wherein i d (k) and i q (k) are currents in a dq coordinate system at the time of k respectively; i d (k+1) and i q (k+1) are currents in the dq coordinate system at the time of k+1, respectively; u d (k) and u q (k) are voltages in the dq coordinate system at time k, respectively; e d (k) and e q (k) are respectively estimated values of back electromotive force at k time in dq coordinate system, and T s is sampling time.
3. The simplified permanent magnet synchronous motor model predictive current control method of claim 1, further comprising, through two times of dead beat calculation of current, synthesizing a final action vector u applied from two non-zero vectors and a zero vector by a certain proportion, wherein the range of the final action vector can cover the whole hexagonal area, and the current tracking error is greatly reduced, so that the steady state performance of the system is improved.
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| CN111600522A (en) * | 2020-05-08 | 2020-08-28 | 北方工业大学 | Motor model prediction current control method and device, electronic equipment and medium |
| CN112054736A (en) * | 2020-09-11 | 2020-12-08 | 南通大学 | Permanent magnet synchronous motor model prediction current overmodulation control method for optimizing zone modulation |
| CN112803861A (en) * | 2021-03-19 | 2021-05-14 | 哈尔滨理工大学 | Zero-vector-free algorithm for predictive control of three-vector model of permanent magnet synchronous motor |
| CN113098349A (en) * | 2021-04-28 | 2021-07-09 | 杭州电子科技大学 | Discrete space vector modulation permanent magnet synchronous motor model prediction control method |
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| CN111600522A (en) * | 2020-05-08 | 2020-08-28 | 北方工业大学 | Motor model prediction current control method and device, electronic equipment and medium |
| CN112054736A (en) * | 2020-09-11 | 2020-12-08 | 南通大学 | Permanent magnet synchronous motor model prediction current overmodulation control method for optimizing zone modulation |
| CN112803861A (en) * | 2021-03-19 | 2021-05-14 | 哈尔滨理工大学 | Zero-vector-free algorithm for predictive control of three-vector model of permanent magnet synchronous motor |
| CN113098349A (en) * | 2021-04-28 | 2021-07-09 | 杭州电子科技大学 | Discrete space vector modulation permanent magnet synchronous motor model prediction control method |
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