CN113872479A

CN113872479A - Permanent magnet synchronous motor controller with bus current estimation function and driving equipment

Info

Publication number: CN113872479A
Application number: CN202111119949.XA
Authority: CN
Inventors: 徐晖; 王满江; 张大双; 周建刚; 普刚; 钟亮
Original assignee: Dongfeng Trucks Co ltd
Current assignee: Dongfeng Trucks Co ltd
Priority date: 2021-09-24
Filing date: 2021-09-24
Publication date: 2021-12-31
Anticipated expiration: 2041-09-24
Also published as: CN113872479B

Abstract

The invention discloses a permanent magnet synchronous motor controller with a bus current estimation function and driving equipment. The device comprises an angle acquisition module, a position acquisition module and a control module, wherein the angle acquisition module is used for acquiring the position angle of a motor rotor; the current sampling module is used for collecting three-phase current of the motor; the coordinate transformation module is used for carrying out coordinate transformation on the three-phase current according to the position angle; the rotating speed loop module is used for determining given current according to the set rotating speed and the feedback rotating speed; the current loop module is used for determining a given voltage according to the feedback current and the given current; the SVPWM module is used for calculating the duty ratio of the three-phase square wave according to the static voltage; the inverter module is used for outputting three-phase voltage according to the duty ratio of the three-phase square wave; and the bus current calculating module is used for calculating the bus current according to the three-phase current and the three-phase square wave duty ratio. The motor controller estimates the direct current bus current by a method of reconstructing three-phase current, and the estimation method is simple and accurate in result.

Description

Permanent magnet synchronous motor controller with bus current estimation function and driving equipment

Technical Field

The invention belongs to the technical field of new energy automobiles, and particularly relates to a permanent magnet synchronous motor controller with a bus current estimation function and driving equipment.

Background

The permanent magnet synchronous motor (PMSM for short) has the outstanding advantages of high power density, accurate positioning, wide speed regulation range, stable low-speed operation, small torque pulsation and the like, and is widely applied to the field of new energy vehicles.

In order to realize the closed-loop control of the permanent magnet synchronous motor of the electric compressor, the phase current of the stator winding needs to be detected in real time, and there are generally three methods: (1) detecting by using an alternating current transformer; (2) detecting by using a Hall sensor; (3) and detecting by using a precise sampling resistor. The alternating current transformer isolates strong current and weak current, has strong anti-interference capability and small influence by temperature, but has larger volume. The Hall sensor also has strong and weak electric isolation, strong anti-interference capability, but is greatly influenced by temperature and has larger volume. The precise sampling resistor has no isolation function, has poor anti-interference capability, has high requirements on the design and wiring of the controller, has smaller size and is suitable for the condition that the installation size of the controller is limited.

The automobile has strict requirements on the size of each part, and any increase in the size of the part has an influence on the size and layout of the whole automobile. In order to reduce the size of the controller, it is appropriate to use a precision sampling resistor to detect the current. To reduce the size, neither the control channel nor the detection channel is provided with isolation. Thus, the strong and weak currents must be grounded, and the anti-interference problem must be considered during design and wiring.

Theoretical analysis shows that under the condition of normal switching of the IGBT, the phase current of the stator winding cannot be correctly detected only by using two sampling resistors. In order to correctly detect the phase current, the dead time and the follow current function of the follow current diode are reasonably utilized to turn off all the upper bridge arms, so that the phase current can be correctly detected only by two precise sampling resistors, and the size of an electronic circuit board of the controller is effectively reduced. However, in the dead time, when the upper arm of the inverter is completely turned off, the current on the dc bus is zero, and the dc bus current cannot be detected, so that the overload protection cannot be realized.

The total power P is U · I, where U is the dc bus voltage and I is the dc bus current. Since the dc bus current cannot be detected, the total power cannot be calculated.

Disclosure of Invention

The present invention is directed to solve the above-mentioned drawbacks of the prior art, and provides a pmsm controller and a driving device with a bus current estimation function, which can estimate the bus current more accurately.

The technical scheme adopted by the invention is as follows: a permanent magnet synchronous motor controller with bus current estimation function comprises

The angle acquisition module is used for acquiring the position angle of the motor rotor, sending the position angle to the coordinate transformation module, carrying out differential processing on the position angle to obtain a feedback rotating speed, and sending the feedback rotating speed to the rotating speed ring module

The current sampling module is used for collecting three-phase currents Ia, Ib and Ic of the motor and sending the three-phase currents to the coordinate transformation module;

the coordinate transformation module is used for carrying out coordinate transformation on the three-phase current according to the position angle to obtain feedback currents Id and Iq under a rotating coordinate system and sending the feedback currents Id and Iq to the current loop module; the SVPWM module is used for converting the coordinate of the given voltage to obtain the static voltage under a two-phase static coordinate system and sending the static voltage to the SVPWM module

A rotating speed loop module used for determining the given torque according to the set rotating speed and the feedback rotating speed, obtaining the given current ID and IQ after looking up the table and sending the given current ID and IQ to the current loop module,

the current loop module is used for determining given voltages Ud and Uq according to the feedback current and the given current, sending the given voltages Ud and Uq to the coordinate transformation module, and performing inverse transformation on the coordinate transformation module;

an SVPWM module for calculating the duty ratio of the three-phase square wave according to the static voltage Ualpha and the Ubeta and sending the duty ratio to the inverter and the bus current calculating module

The inverter module is used for outputting three-phase voltage according to the duty ratio of the three-phase square wave;

and the bus current calculating module is used for calculating the bus current according to the three-phase current and the three-phase square wave duty ratio.

Further, the bus current was calculated by the following formula

I_dc＝-(CMPa*I_a+CMPb*I_b+CMPc*I_c)/T_pwm

Wherein, I_dcIs the bus current; CMPa, CMPb and CMPc are ABC three-phase square wave comparison values respectively; ia. Ib and Ic are ABC three-phase currents respectively; t is_pwmHalf the carrier period.

Further, the sampling time of the CWPA, the CMPb and the CMPc is between the time after the square wave comparison value of the previous period and the time before the dead zone compensation algorithm of the previous period.

Further, after the bus current is calculated by the bus current calculation module, the bus current is filtered to obtain an estimated bus current, and then the estimated bus current is compensated to obtain a final compensated bus current.

Further, the bus current is compensated for in the following manner

I'_dc＝I_dc*K

Wherein, I'_dcTo compensate for bus current; i is_dcIs the bus current; k is a compensation coefficient.

Further, the K is determined by the following formula:

k is (dead zone ratio + modulation factor)/modulation factor.

Further, the angle sampling module is an angle encoder.

Further, the current sampling module is a hall current sensor.

Further, the coordinate transformation module includes

The Clarke transformation module is used for transforming the three-phase static coordinate system to the two-phase static coordinate system;

the Park transformation module is used for transforming the two-phase static coordinate system to the two-phase rotating coordinate system;

and the IPark transformation module is used for transforming the two-phase rotating coordinate system to the two-phase static coordinate system.

A steering apparatus comprising a permanent magnet synchronous motor controller as claimed in any preceding claim.

The invention has the beneficial effects that: the motor controller estimates the direct current bus current by a method of reconstructing three-phase current, estimates the bus current by analyzing the relationship between the bus current and the three-phase current under each switching vector, and has simple estimation method and accurate result. By using the estimation result, overload protection and estimation of total power consumption can be further realized, and control over the vehicle can be better realized.

Drawings

Fig. 1 is a schematic diagram of a motor controller according to the present invention.

Detailed Description

The following further describes embodiments of the present invention with reference to the drawings. It should be noted that the description of the embodiments is provided to help understanding of the present invention, but the present invention is not limited thereto. In addition, the technical features involved in the embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.

In the description of the present invention, it is to be understood that the terms "upper", "lower", "front", "rear", "left", "right", "top", "bottom", "inner", "outer", and the like, indicate orientations or positional relationships based on the orientations or positional relationships shown in the drawings, are merely for convenience in describing the present invention and simplifying the description, and do not indicate or imply that the device or element being referred to must have a particular orientation, be constructed and operated in a particular orientation, and thus, should not be construed as limiting the present invention.

Where the terms "comprising", "having" and "including" are used in this specification, there may be another part or parts unless otherwise stated, and the terms used may generally be in the singular but may also be in the plural.

It should be noted that although the terms "first," "second," "top," "bottom," "side," "other," "end," "other end," and the like may be used and used in this specification to describe various components, these components and parts should not be limited by these terms. These terms are only used to distinguish one element or section from another element or section. For example, a first element could be termed a second element, and, similarly, a second element could be termed a first element, with the top and bottom elements being interchangeable or switchable with one another, where appropriate, without departing from the scope of the present description; the components at one end and the other end may be of the same or different properties to each other.

Further, in constituting the component, although it is not explicitly described, it is understood that a certain error region is necessarily included.

In describing positional relationships, for example, when positional sequences are described as being "on.. above", "over.. below", "below", and "next", unless such words or terms are used as "exactly" or "directly", they may include cases where there is no contact or contact therebetween. If a first element is referred to as being "on" a second element, that does not mean that the first element must be above the second element in the figures. The upper and lower portions of the member will change depending on the angle of view and the change in orientation. Thus, in the drawings or in actual construction, if a first element is referred to as being "on" a second element, it can be said that the first element is "under" the second element and the first element is "over" the second element. In describing temporal relationships, unless "exactly" or "directly" is used, the description of "after", "subsequently", and "before" may include instances where there is no discontinuity between steps.

The features of the various embodiments of the present invention may be partially or fully combined or spliced with each other and performed in a variety of different configurations as would be well understood by those skilled in the art. Embodiments of the invention may be performed independently of each other or may be performed together in an interdependent relationship.

As shown in FIG. 1, the present invention provides a PMSM controller with bus current estimation function, comprising

In the above scheme, the bus current is calculated by the following formula

I_dc＝-(CMPa，I_a+CMpb*I_b+CMpc*I_c)/T_pwm

Wherein, I_dcIs the bus current; CMPa, CMPb and CMPc are ABC three-phase square wave comparison values respectively; ia. Ib and Ic are ABC three-phase power respectivelyA stream; t is_pwmHalf the carrier period. And the value taking time of the CMPa, the CMPb and the CMPc is between the time after the square wave comparison value of the previous period and the time before the dead zone compensation algorithm of the previous period.

In the above scheme, after the bus current calculation module calculates the bus current, the bus current is filtered to obtain an estimated bus current, and then the estimated bus current is compensated to obtain a final compensated bus current.

In the above scheme, the bus current is compensated in the following manner

I'_dc＝I_dc*K

Wherein, I'_dcTo compensate for bus current; i is_dcIs the bus current; k is a compensation coefficient. The K is determined by the following formula:

k is (dead zone ratio + modulation factor)/modulation factor.

In the above scheme, the angle sampling module is an angle encoder.

In the above scheme, the current sampling module is a hall current sensor.

In the above solution, the coordinate transformation module comprises

After the set rotating speed SPDset of a Permanent Magnet Synchronous Motor (PMSM) is set, a motor controller transmits the deviation amount of a rotating speed feedback value and a rotating speed set value to a speed ring; the speed loop PI regulator processes the input error signal according to the set PI control parameter and converts the input error signal into a target torque model; obtaining target current signals ID and IQ of a DQ axis through a look-up table, and outputting the target current signals ID and IQ to a current loop regulator as given values; obtaining a three-phase current value through current sampling, then obtaining feedback values id and iq of the DQ axis current through CLARK conversion and PARK conversion, and outputting the feedback values id and iq as feedback to a current loop regulator; the current loop regulator converts the current signal into the voltage UD and UQ of the quadrature axis for output after the PI regulation; after an IPARK conversion link, Ualfa and Ubeta are obtained; and the signals are transmitted to an SVPWM module to generate 6 paths of duty ratio signals to drive a PMSM to operate. The detailed functions of the modules are respectively as follows:

an angle sampling module: the angle signal theta is collected through the encoder, and then the feedback rotating speed signal spd can be obtained through a differential link. An incremental encoder is commonly used, and an output pulse signal of a rotary encoder can be directly input to a PLC (programmable logic controller), and the pulse signal of the PLC is counted by using a high-speed counter of the PLC to obtain a measurement result. The rotary encoders of different models have different phase numbers of output pulses, and some rotary encoders output A, B, Z three-phase pulses, some rotary encoders output A, B two-phase pulses, and the simplest rotary encoders output only a phase A. The encoder has 5 leads, 3 of which are pulse output lines, 1 of which is a COM terminal line, and 1 of which is a power supply line (OC gate output type). The power supply of the encoder can be an external power supply, and can also directly use the DC24V power supply of the PLC. The "-" terminal of the power supply is connected with the COM terminal of the encoder, and the "+" terminal is connected with the power supply terminal of the encoder. The COM end of the encoder is connected with the COM end of the PLC input, the A, B, Z two-phase pulse output line is directly connected with the input end of the PLC, A, B is pulses with a phase difference of 90 degrees, a Z-phase signal only has one pulse in one rotation of the encoder and is usually used as a basis for a zero point, and the response time of the PLC input needs to be noticed during connection. The rotary encoder is also provided with a shielding wire, and the shielding wire is grounded when the rotary encoder is used, so that the anti-interference performance is improved.

A current sampling module: two-phase current was collected by hall sensor.

A coordinate transformation module: the coordinate systems to be used in vector control systems can be divided into two categories: one is a stationary coordinate system, including a three-phase stationary coordinate system ABC and a two-phase stationary coordinate system a β; the other type is a rotating coordinate system, which is divided into a rotor rotating coordinate system and a stator rotating coordinate system, and a rotor rotating coordinate system dq is used herein. The transformation of the ABC three-phase stationary coordinate system to the two-phase stationary coordinate system a β is generally referred to as Clarke transformation, and the transformation of a β to the two-phase rotating coordinate system dq is generally referred to as Park transformation. The following specifically describes the specific processes of the two coordinate transformations. However, in practical simulation, a three-phase stationary natural coordinate system is directly transformed into a coordinate system synchronously rotating with the rotor, which is called Park transformation. The results before and after the coordinate transformation can be seen from the simulations below.

Clarke transformation module:

in order to simplify the operation, an a axis in a two-phase static coordinate system is defined to coincide with a phase winding of a stator, and a beta axis leads the a axis by 90 space electrical angles anticlockwise. Obtaining a transformation matrix C3s/2s according to the constant amplitude transformation principle as follows:

the transformation of physical quantities in the three-phase stationary coordinate system into the two-phase stationary coordinate system according to the above formula can be expressed as:

a Park transformation module:

and defining a two-phase coordinate system dq which rotates at a synchronous speed in space, wherein the d axis is superposed with the axis of the rotor magnetic pole, the q axis leads the d axis by 90-degree space electrical angle anticlockwise, and the included angle between the d axis and the A-phase stator winding is theta. The transformation matrix C2 s/2r can also be obtained as:

the physical quantities that can be obtained in a two-phase rotating coordinate system can be expressed as:

a current loop module:

in a closed-loop control system, a current loop belongs to an inner loop, the function of the current loop is to enable the current of a motor to follow the change of given current, and the current loop has important influence on the rapidity and the accuracy of system response. According to a voltage balance equation of the permanent magnet synchronous motor, the influence of quadrature-direct axis coupling and back electromotive force is not considered, and an idealized current loop model of a PI controller is added.

A rotating speed ring module:

the rotating speed ring belongs to an outer ring and has the function of enabling the rotating speed of the motor to follow the change of the given rotating speed. The input is the rotational speed deviation, and the output is the given torque.

And an SVPWM module:

the theoretical basis of SVPWM is the principle of mean value equivalence, i.e. the mean value of a basic voltage vector is made equal to a given voltage vector by combining the basic voltage vectors during a switching cycle. At a certain moment, the voltage vector rotates into a certain area, which can be obtained by two adjacent non-zero vectors and different combinations of zero vectors in time that make up this area. The action time of the two vectors is applied for a plurality of times in a sampling period, thereby controlling the action time of each voltage vector, enabling the voltage space vector to approach to rotate according to a circular track, approaching to an ideal magnetic flux circle through the actual magnetic flux generated by different switching states of the inverter, and determining the switching state of the inverter according to the comparison result of the two, thereby forming a PWM waveform. And the current magnitude of the three phases of the motor is controlled by controlling the complementary PWM signals of the upper bridge and the lower bridge of the 3 paths. The output is three comparison values CMPa, CMPb and CMPc.

In the inverter circuit, the voltage on a direct current bus is Udc, three-phase voltages output by an inverter are UA, UB and UC, the three-phase voltages are respectively applied to a plane coordinate system with a spatial difference of 120 degrees, three voltage space vectors are defined as UA (t), UB (t) and UC (t), the directions of the three voltage space vectors are always on respective axes, the magnitudes of the three voltage space vectors change with time according to a sine law, and the time phases are different by 120 degrees. Assuming Um is the effective value of the phase voltage and f is the power frequency, then:

the resultant space vector u (t) of the three-phase voltage space vector addition can be expressed as:

it can be seen that u (t) is a rotating space vector, whose amplitude is constant, and is a phase voltage peak value, and rotates at a constant speed in the counterclockwise direction at an angular frequency ω -2 pi f. The purpose of the SVPWM algorithm is to represent the u (t) vector rotating in space using the switching states of the three-phase bridge.

Since the inverter has 6 switching tubes in total for three-phase arms, in order to study the space voltage vector output by the inverter when the upper and lower arms of each phase are combined with different switches, a specific switching function Sx (x ═ a, b, c) is defined as follows:

all possible combinations of (Sa, Sb, Sc) are eight in total, including 6 non-zero vectors Ul (001), U2(010), U3(011), U4(100), U5(101), U6(110), and two zero vectors U0(000), U7(111), and the following is an analysis taking one of the switch combinations as an example, assuming Sx (x ═ a, b, c) ═ 100, so the phase voltages can be expressed as: (phase voltage is the voltage of each phase relative to the motor intermediate connection point)

The same can be said that the phase voltages of the three phases in other switch states. In addition, the line voltage is the voltage difference between two phases, for example Uab ═ Ua-Ub.

As previously mentioned

When the switch Sa is 1, ua (t) Udc; when the switch Sb is 1, ub (t) is Udc; when the switch Sc is 1, uc (t) Udc.

The three-phase voltage gives the synthesized voltage vector rotation angular speed to be omega-2 pi f, and the time required by one rotation is T-1/f; if the carrier frequency is fs, the frequency ratio is R ═ fs/f. Thus, the voltage rotation plane etc. is cut into R small increments, i.e. the angle of each increment of the voltage vector is set to: γ is 2 π/R.

Now, assuming that a space vector Uref needs to be output, we first take out the I-th sector separately and then represent it with two voltage space vectors adjacent to it.

In a two-phase stationary reference frame (α, β), let U_refAnd U₄The angle between is θ, which can be obtained by the sine theorem:

because of | U₄|＝|U₆|＝2U_dcAnd/3, therefore, the state retention time of each vector can be obtained as follows:

where m is the SVPWM modulation factor (modulation ratio),

and the time allocated for the zero voltage vector is:

T₇＝T₀＝(T_S-T₄-T₆)/2

having obtained the time for Uref synthesized with U4, U6, U7, and U0, it follows how to generate the actual pulse width modulated waveform. In the SVPWM modulation scheme, the selection of the zero vector is the most flexible, and the zero vector is properly selected, so that the switching frequency can be reduced to the maximum extent, the switching action at the moment when the load current is large can be avoided as much as possible, and the switching loss can be reduced to the maximum extent. Therefore, we aim to reduce the number of switching times and choose the distribution principle of the basic vector action sequence as follows: the switching state of only one of the phases is changed at each switching state transition. And the zero vectors are equally distributed in time to make the generated PWM symmetrical, thereby effectively reducing harmonic components of the PWM. It can be seen that when U4(100) is switched to U0(000), only the upper and lower pairs of switches of phase a need to be changed, and when U4(100) is switched to U7(111), the upper and lower pairs of switches of phase B, C need to be changed, which doubles the switching loss. Therefore, to change the magnitudes of the voltage vectors U4(100), U2(010), and U1(001) needs to match the zero voltage vector U0(000), and to change the magnitudes of the voltage vectors U6(110), U3(011), and U5(100) needs to match the zero voltage vector U7 (111). Thus, by arranging different switching sequences in different intervals, symmetrical output waveforms can be obtained, with the switching sequences for other sectors shown in Table 2-2.

TABLE 2-2U_REFThe position and switch switching sequence of the switch

U_REFAt the position of	Switch switching
		Zone I (theta is more than or equal to 0 degree and less than or equal to 60 degree)	...0-4-6-7-7-6-4-0...
Zone II (theta is more than or equal to 60 degrees and less than or equal to 120 degrees)	...0-2-6-7-7-6-2-0...
		Zone III (theta is more than or equal to 120 degrees and less than or equal to 180 degrees)	...0-2-3-7-7-3-2-0...
IV zone (theta is more than or equal to 180 degrees and less than or equal to 240 degrees)	...0-1-3-7-7-3-1-0...
		V zone (theta is more than or equal to 240 degrees and less than or equal to 300 degrees)	...0-1-5-7-7-5-1-0...
VI zone (theta is more than or equal to 300 degrees and less than or equal to 360 degrees)	...0-4-5-7-7-5-4-0...

Therefore, Uref can be shown by the sequence and time length of U4, U6, U7 and U0.

Taking sector I as an example, the generated three-phase wave modulation waveform is in a carrier period time Ts, the sequence of voltage vectors is U0, U4, U6, U7, U6, U4 and U0, and the three-phase waveform of each voltage vector corresponds to the representation symbol of the switch in table 2-2. Then, the angle of the next carrier period Ts and Uref is increased by gamma, and new values of T0, T4, T6 and T7 can be recalculated by using the formula (2-33) to obtain a new synthesized three-phase waveform; thus, each carrier period TS will synthesize a new vector, and as θ increases, Uref will enter the I, II, III, IV, V, VI regions in sequence. After a period of voltage vector rotation, R resultant vectors are generated. SVPWM performs a calculation once per carrier period.

Through the derivation and analysis of the SVPWM rule, it can be known that to implement the real-time modulation of the SVPWM signal, the interval position where the reference voltage vector Uref is located needs to be known first, and then the reference voltage vector is synthesized by using the two adjacent voltage vectors of the located sector and a proper zero vector.

The control system needs to output a vector voltage signal Uref which rotates anticlockwise in space at a certain angular frequency omega, when the vector voltage signal Uref rotates to a certain 60-degree sector of a vector diagram, the system calculates a basic voltage space vector required by the interval, and drives the power switching element to act according to the state corresponding to the vector. When the control vector rotates 360 degrees in space, the inverter can output sine wave voltage of one period.

The modulation actions generated by the space vector wave-transmitting method can be represented by 8 voltage vectors. And the bus current and the phase current have a determined relationship under each voltage vector.

During the action period of each basic voltage vector, the switching state of the basic voltage vector is kept unchanged, the current loop is fixed, and at the moment, the bus current and the phase current of the motor winding have a certain corresponding relation. The state that the upper switch tube of each phase of bridge arm is conducted, the lower switch tube is conducted and is turned off is defined as '1', the state that the lower switch tube is conducted, the upper switch tube is conducted and is turned off is defined as '0', so that a current loop is obtained when the switch state is (100), current flows into the phase A winding at the moment, current flows out of the phase B, C winding, and the bus current is consistent with the phase A current; similarly, when the switch state is (110), the current flows into the current loop from the A, B phase winding and flows out from the C phase winding, namely, the bus current is consistent with the C phase current.

The corresponding relation between the bus current and the phase current shown in the table 1 is obtained by analyzing the current loop under the eight basic voltage vector switch states. Wherein the direction of current flow into the winding is defined as positive and the direction of current flow out of the winding is defined as negative.

TABLE 1

According to the table, the bus current can be calculated in real time according to the switching state, the average value of the bus current is obtained after low-pass filtering (the response of the direct current bus current estimation has no clear index, and the direct current bus current estimation is only used by a whole vehicle controller at present and does not need to be carried out quickly), the action time of each voltage vector in one switching period can be calculated, and the same effect can be achieved.

Within this switching period there are four switching vectors: u0, U4, U6, U7. The magnitude of the bus current for each switching vector is known from table 1. The magnitude of the bus current can then be calculated throughout this switching cycle by integration. The calculation formula is as follows:

I_dc＝I_dc0+I_dc4+I_dc6+I_dc7 (1)

I_c＝-(I_a+I_b) (5)

wherein: i is_dcIs the bus current; i is_dcnThe bus current is the bus current when the vector n is switched; t is a carrier period;

I_a、I_b、I_cABC three-phase current; t is_pwmHalf of the carrier period;

CMPa, CMPb and CMPc are ABC three-phase square wave comparison values

From (1) (2) (3) (4) (5) it can be deduced:

I_dc＝-(CMPa*I_a+CMPb*I_b+CMPc*I_c)/T_pwm (6)

and deducing other combinations of various switching vectors to obtain a bus current calculation expression shown in the formula (6). I is_dcObtaining an estimated value I before compensation through moderate low-pass filtering_{dc_est}。

Aiming at estimation errors possibly brought by dead zone effects, the following two compensation modes are adopted:

the compensation method 1: for the case that no dead zone compensation is started in the program, on the basis of the above estimation method, a compensation coefficient K is multiplied, where K is (dead zone ratio + modulation coefficient)/modulation coefficient, where the dead zone ratio is determined by bench test, positive in the case of power-on and negative in the case of feedback. The switching between motoring and regenerative conditions is determined by the sign of I _ (dc _ est).

The compensation method 2 comprises the following steps: for the case of open dead-time compensation in the program, the bus current calculation is performed using the comparison value without dead-time compensation.

In addition, to ensure that the comparison value corresponds to the current value, the comparison value that is being sent out in the current cycle needs to be tried.

Therefore, for the compensation method 1, the sampling time of CMPa, CMPb, and CMPc needs to be before the calculation of the comparison value in the period is completed. For the compensation method 2, the values of CMPa, CMPb and CMPc are required to be placed before the compensation algorithm of the dead zone of the upper period.

And comprehensively considering the two points, the value taking moments of CMPa, CMPb and CMPc need to be put after the comparison value of the square wave in the upper period is calculated and before the dead zone compensation algorithm.

Example 1:

and on the material object rack, estimating the bus current according to the technical scheme description method, and taking the bus current measured by the power analyzer as a reference. The motor voltage platform is 380V, the rated power is 30KW, the rated rotating speed is 3000rpm, and the rated torque is 95.5N.

During the experiment, the rotating speed is 500rpm, 1000rpm and 3000 rpm; the torque is selected from 0%, 10%, 40% and 70% of rated torque. And testing the accuracy of the estimated bus current at each working condition point. The test results are shown in table 2:

TABLE 2

It can be seen that under various working conditions, a more accurate bus current can be estimated by the method. The relative error can be controlled within 3 percent.

The embodiment of the invention also provides driving equipment, which comprises the controller in any embodiment.

The driving device provided by the embodiment of the invention comprises the control device of the fan in the embodiment, so that the driving device has the same technical characteristics as the control device of the fan in the embodiment, the same technical problems can be solved, and the same technical effects can be achieved.

It should be understood that the specific order or hierarchy of steps in the processes disclosed is an example of exemplary approaches. Based upon design preferences, it is understood that the specific order or hierarchy of steps in the processes may be rearranged without departing from the scope of the present disclosure. The accompanying method claims present elements of the various steps in a sample order, and are not intended to be limited to the specific order or hierarchy presented.

The foregoing description of the embodiments and specific examples of the invention have been presented for purposes of illustration and description; it is not intended to be the only form in which the embodiments of the invention may be practiced or utilized. The embodiments are intended to cover the features of the various embodiments as well as the method steps and sequences for constructing and operating the embodiments. However, other embodiments may be utilized to achieve the same or equivalent functions and step sequences.

In the foregoing detailed description, various features are grouped together in a single embodiment for the purpose of streamlining the disclosure. This method of disclosure is not to be interpreted as reflecting an intention that the claimed embodiments of the subject matter require more features than are expressly recited in each claim. Rather, as the following claims reflect, invention lies in less than all features of a single disclosed embodiment. Thus, the following claims are hereby expressly incorporated into the detailed description, with each claim standing on its own as a separate preferred embodiment of the invention.

The previous description of the disclosed embodiments is provided to enable any person skilled in the art to make or use the present invention. To those skilled in the art; various modifications to these embodiments will be readily apparent, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the disclosure. Thus, the present disclosure is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

What has been described above includes examples of one or more embodiments. It is, of course, not possible to describe every conceivable combination of components or methodologies for purposes of describing the aforementioned embodiments, but one of ordinary skill in the art may recognize that many further combinations and permutations of various embodiments are possible. Accordingly, the embodiments described herein are intended to embrace all such alterations, modifications and variations that fall within the scope of the appended claims. Furthermore, to the extent that the term "includes" is used in either the detailed description or the claims, such term is intended to be inclusive in a manner similar to the term "comprising" as "comprising" is interpreted when employed as a transitional word in a claim. Furthermore, any use of the term "or" in the specification of the claims is intended to mean a "non-exclusive or".

Those of skill in the art will further appreciate that the various illustrative logical blocks, units, and steps described in connection with the embodiments disclosed herein may be implemented as electronic hardware, computer software, or combinations of both. To clearly illustrate the interchangeability of hardware and software, various illustrative components, elements, and steps have been described above generally in terms of their functionality. Whether such functionality is implemented as hardware or software depends upon the particular application and design requirements of the overall system. Skilled artisans may implement the described functionality in varying ways for each particular application, but such implementation decisions should not be interpreted as causing a departure from the scope of the present embodiments.

The various illustrative logical blocks, or elements, described in connection with the embodiments disclosed herein may be implemented or performed with a general purpose processor, a digital signal processor, an Application Specific Integrated Circuit (ASIC), a field programmable gate array or other programmable logic device, discrete gate or transistor logic, discrete hardware components, or any combination thereof designed to perform the functions described herein. A general-purpose processor may be a microprocessor, but in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine. A processor may also be implemented as a combination of computing devices, e.g., a digital signal processor and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a digital signal processor core, or any other similar configuration.

The foregoing is considered as illustrative of the preferred embodiments of the invention and the technical principles employed. It will be understood by those skilled in the art that the present invention is not limited to the particular embodiments described herein, but is capable of various obvious modifications, rearrangements, combinations and substitutions as will now become apparent to those skilled in the art without departing from the scope of the invention. Therefore, although the present invention has been described in greater detail by the above embodiments, the present invention is not limited to the above embodiments, and may include other equivalent embodiments without departing from the spirit of the present invention, and the scope of the present invention is determined by the scope of the appended claims.

Claims

1. A permanent magnet synchronous motor controller with a bus current estimation function is characterized in that: comprises that

The angle acquisition module is used for acquiring the position angle of the motor rotor, sending the position angle to the coordinate transformation module, carrying out differential processing on the position angle to obtain a feedback rotating speed, and sending the feedback rotating speed to the rotating speed loop module;

the current sampling module is used for collecting three-phase current of the motor and sending the three-phase current to the coordinate transformation module;

the coordinate transformation module is used for carrying out coordinate transformation on the three-phase current according to the position angle to obtain feedback current in a rotating coordinate system and sending the feedback current to the current loop module; the SVPWM module is used for converting the coordinate of the given voltage to obtain the static voltage under a two-phase static coordinate system and sending the static voltage to the SVPWM module

The rotating speed loop module is used for determining a given torque according to a set rotating speed and a feedback rotating speed, obtaining a given current after table lookup and sending the given current to the current loop module;

the current loop module is used for determining a given voltage according to the feedback current and the given current, sending the given voltage to the coordinate transformation module and carrying out inverse transformation on the coordinate transformation module;

the SVPWM module is used for calculating the duty ratio of the three-phase square wave according to the static voltage and sending the duty ratio to the inverter and the bus current calculation module;

2. The permanent magnet synchronous motor controller with bus current estimation function according to claim 1, wherein: calculating the bus current by the following formula

I_dc＝-(CMPa*I_a+CMPb*I_b+CMPc*I_c)/T_pwm

3. The permanent magnet synchronous motor controller with bus current estimation function according to claim 2, wherein: and the value taking time of the CMPa, the CMPb and the CMPc is between the time after the square wave comparison value of the previous period and the time before the dead zone compensation algorithm of the previous period.

4. The permanent magnet synchronous motor controller with bus current estimation function according to claim 1, wherein: and after the bus current calculation module calculates the bus current, the bus current is filtered to obtain an estimated bus current, and then the estimated bus current is compensated to obtain a final compensated bus current.

5. The permanent magnet synchronous motor controller with bus current estimation function according to claim 4, wherein: compensating for bus current in the following manner

I′_dc＝I_dc*K

Wherein，I'_dcTo compensate for bus current; i is_dcIs the bus current; k is a compensation coefficient.

6. The permanent magnet synchronous motor controller with bus current estimation function according to claim 5, wherein: the K is determined by the following formula:

k is (dead zone ratio + modulation factor)/modulation factor.

7. The permanent magnet synchronous motor controller with bus current estimation function according to claim 1, wherein: the angle sampling module is an angle encoder.

8. The permanent magnet synchronous motor controller with bus current estimation function according to claim 1, wherein: the current sampling module is a Hall current sensor.

9. The permanent magnet synchronous motor controller with bus current estimation function according to claim 1, wherein: the coordinate transformation module comprises

10. A steering device, characterized in that it comprises a permanent magnet synchronous motor controller according to any of the preceding claims 1-9.