CN113824664B - Demodulation method of TCM-CPM signal under multipath channel - Google Patents

Demodulation method of TCM-CPM signal under multipath channel Download PDF

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CN113824664B
CN113824664B CN202111285561.7A CN202111285561A CN113824664B CN 113824664 B CN113824664 B CN 113824664B CN 202111285561 A CN202111285561 A CN 202111285561A CN 113824664 B CN113824664 B CN 113824664B
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state node
tcm
frequency offset
code element
demodulator
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CN113824664A (en
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孙靖翔
孙锦华
张春晖
刘玉涛
许名岳
赵伊倩
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Xidian University
CETC 54 Research Institute
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/10—Frequency-modulated carrier systems, i.e. using frequency-shift keying
    • H04L27/14—Demodulator circuits; Receiver circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/10—Frequency-modulated carrier systems, i.e. using frequency-shift keying
    • H04L27/14—Demodulator circuits; Receiver circuits
    • H04L27/142—Compensating direct current components occurring during the demodulation and which are caused by mistuning

Abstract

The invention provides a demodulation method of a TCM-CPM signal under a multipath channel, which comprises the following steps: the TCM-CPM modulator modulates the binary sequence, the frequency offset estimator performs frequency offset compensation on the modulated signal, the demodulator performs first Viterbi demodulation on the frequency offset compensation signal, and then the demodulator performs second Viterbi demodulation on the frequency offset compensation signal. In the process of calculating the branch measurement by the second Viterbi demodulation, the demodulated code element, the current demodulated code element and the modulation signal corresponding to the first demodulation result are used as the reference signal to carry out convolution with the channel gain value, so that the influence of multipath interference on the demodulation result can be inhibited to the maximum extent, and the bit error rate is effectively reduced.

Description

Demodulation method of TCM-CPM signal under multipath channel
Technical Field
The invention belongs to the technical field of wireless communication, and further relates to a demodulation method of a TCM-CPM signal under a multipath channel, which can be used for demodulating the TCM-CPM signal under a multipath environment.
Background
TCM-CPM modulation is of great interest due to its excellent power and bandwidth efficiency. By introducing inherent inter-symbol interference (ISI) into the signal, the spectral shape of the TCM-CPM signal can be easily adjusted to meet system requirements.
Continuous phase modulation CPM originated in the late 70's of the last century and was less affected by nonlinear distortion due to its constant envelope characteristics. In addition, the method has the excellent characteristics of compact frequency spectrum, less high-frequency components, large sidelobe suppression and the like, and has good development prospect. However, in today's complex wireless communication environment, the CPM modulation technique alone cannot achieve better error performance, and the channel coding technique must be adopted. For the channel with limited power, the channel coding can be adopted to obtain coding gain for the communication system to a certain extent, and improve the power utilization rate, but the cost is to increase the transmission bandwidth or reduce the useful information rate. Due to the contradiction between the effectiveness and the reliability of the system, the traditional code modulation method is not careful in the channel with limited power and frequency band.
In the 80 s of the 20 th century, Trellis-coded modulation (TCM) technology was proposed for the first time by g.ungerboeck et al, where TCM designed modulation and coding as a whole, and on one hand, it adopted error correction coding, increased the number of signal sets, and did not increase transmission bandwidth, but gained coding gain. TCM technology is a revolution in channel coding technology, which for the first time solves the problem of reduced spectrum utilization caused by channel coding. Therefore, combining TCM technology with multilevel CPM technology combines the advantages of both technologies, and will further improve the performance of the communication system.
In a multipath channel, the delay spread caused by multipath fading will result in more intersymbol interference and will result in an increased error rate of demodulation. In the demodulation of the TCM-CPM signal under the multipath channel, the key point is how to eliminate the multipath interference and obtain better error code performance. In order to eliminate multipath interference and reduce the error rate, an equalizer may be added in front of the demodulator to eliminate multipath interference, but this will inevitably lead to a complicated structure of the receiving end and increase the design difficulty of the receiving end.
Zhangbo is published in 2017, and in the study of TCM-CPM modulation and demodulation technology, a coherent and incoherent mode TCM-CPM signal demodulation method with low complexity is disclosed, in coherent demodulation, a phase distance path exclusion-based algorithm is adopted to reduce the correlation operation amount of a receiving end, in incoherent demodulation, a simplified state incoherent sequence detection algorithm based on decision feedback is adopted, the number of decoded states can be reduced through reasonable state combination, and the complexity of a receiver is reduced under the condition of small performance loss.
Disclosure of Invention
The present invention aims to provide a demodulation method of TCM-CPM signals in a multipath channel for solving the technical problem of high error rate in the prior art.
In order to achieve the purpose, the technical scheme adopted by the invention comprises the following steps:
(1) the TCM-CPM modulator modulates the binary sequence:
(1a) a TCM-CPM modulator at a sending end adds a locally stored M-bit pilot frequency sequence in front of a binary sequence generated by an information source to form an N-bit sequence X to be modulated, wherein M is more than or equal to 1 and less than N, and N is more than or equal to 2;
(1b) a TCM-CPM modulator at a sending end carries out TCM-CPM digital modulation on a to-be-modulated sequence X to obtain a discrete TCM-CPM modulation signal S ═ Sk|1≤k≤NNsAnd S is passed through NcA multipath channel is transmitted to the receiving end, where skDenotes the kth sample point, N denotes the number of symbols contained in S, NsIndicating the number of samples contained in each symbol, the first MN in SsOne sample point is a pilot signal, and the rest (N-M) NsEach sample point being a data signal, Nc≥2；
(2) And the frequency offset estimator performs frequency offset compensation on the modulation signal:
the frequency deviation estimator at the receiving end receives S through NcA multipath interference signal Y formed after a multipath channel is formed, a rotating average periodogram RPA method is adopted, frequency offset estimation is carried out on Y through a pilot signal in Y, frequency offset compensation is carried out on Y through a frequency offset value delta f obtained by frequency offset estimation, and then a frequency offset compensation signal R of Y is sent to a demodulator, wherein Y is { Y ═ after frequency offset compensation is carried out on Y through a frequency offset value delta f obtained by frequency offset estimationk|1≤k≤NNs}，R＝{rk|1≤k≤NNs}，ykDenotes skSamples after multipath interference, rkDenotes ykSampling points after frequency offset compensation;
(3) the demodulator performs a first viterbi demodulation on the frequency offset compensation signal R:
(3a) initialization parameters of a demodulator at a receiving end: the initialization symbol time n is 1, the initialization demodulator trellis diagram comprises V state nodes, and the fixed initial state node is
Figure BDA0003332817920000021
The path metric value of each state node is 0, and a reference demodulation code element is initialized
Figure BDA0003332817920000022
(3b) The demodulator at the receiving end calculates P branch metrics entering each state node at the nth symbol time, wherein the slave state node
Figure BDA0003332817920000031
Transition to a State node
Figure BDA0003332817920000032
Branch metric of
Figure BDA0003332817920000033
The calculation formula of (2) is as follows:
Figure BDA0003332817920000034
wherein P is more than or equal to 2, cnIs a state node at the time of the (n-1) th symbol
Figure BDA0003332817920000035
State node transition to nth symbol time
Figure BDA0003332817920000036
The corresponding demodulated symbols are then transmitted to the receiver,
Figure BDA0003332817920000037
the v' th state node of the demodulator trellis at the n-1 th symbol time,
Figure BDA0003332817920000038
n is more than or equal to 1 and less than or equal to N, V is more than or equal to 1, V' is less than or equal to V, r is the V-th state node of the demodulator grid diagram at the nth code element momentkIs the m-th sample of the N-th symbol, k ═ N-1Ns+m，1≤m≤Ns，
Figure BDA0003332817920000039
Is StempOne of the plurality of sampling points is selected,
Figure BDA00033328179200000310
representing the channel gain estimate and the reference signal SrefThe convolution of (a) with (b) is performed,
Figure BDA00033328179200000311
is an estimate of the channel gain, jmaxIs ρ of maximum modulus valuejCorresponding subscript, SrefRepresents { Ctemp,cnA corresponding TCM-CPM modulation signal;
(3c) a demodulator at a receiving end respectively accumulates P branch metrics entering each state node at the nth code element moment into the path metric of a transition state node corresponding to the (n-1) th code element moment to obtain P updated path metric values of each state node at the nth code element moment;
(3d) the demodulator at the receiving end takes the path corresponding to the maximum value in the P updated path metric values of each state node at the nth code element moment as a survival path, takes the path with the maximum path metric value in the survival paths of all the state nodes as a temporary maximum likelihood path, backtracks the temporary maximum likelihood path, and obtains a demodulation code element which is the reference demodulation code element Ctemp＝{ci|1≤i≤n}；
(3e) Judging whether N is true or not, if so, selecting the path metric value with the maximum value from the survivor paths of the Nth code element time as the maximum likelihood path, and backtracking the maximum likelihood path to obtain the first Viterbi demodulation result Cref＝{c'nIf not, making N equal to N +1, and executing the step (3 b);
(4) the demodulator performs second Viterbi demodulation on the frequency offset compensation signal R:
(4a) the demodulator at the receiving end carries out initialization: the initialization symbol time n is 1, the initialization demodulator trellis diagram comprises V state nodes, and the fixed initial state node is
Figure BDA00033328179200000312
The path metric value of each state node is 0, and a reference demodulation code element is initialized
Figure BDA00033328179200000313
(4b) The demodulator at the receiving end calculates P branch metrics entering each state node at the nth symbol time, wherein the slave state node
Figure BDA00033328179200000314
Transition to a State node
Figure BDA00033328179200000315
Branch metric of
Figure BDA00033328179200000316
The calculation formula of (2) is as follows:
Figure BDA0003332817920000041
wherein the content of the first and second substances,
Figure BDA0003332817920000042
is S'tempOne of the plurality of sampling points is selected,
Figure BDA0003332817920000043
representing the channel gain estimation value and the reference signal S'refThe convolution of (a) with (b) is performed,
Figure BDA0003332817920000044
is an estimate of the channel gain, jmaxIs ρ of maximum modulus valuejCorresponding subscript, S'refRepresents { Ctemp,cn,C′ref} corresponding TCM-CPM modulation signals, C'refAs a first demodulation result CrefThe symbol following the nth symbol;
(4c) a demodulator at a receiving end respectively accumulates P branch metrics entering each state node at the nth code element moment into the path metric of a transition state node corresponding to the (n-1) th code element moment to obtain P updated path metric values of each state node at the nth code element moment;
(4d) the demodulator at the receiving end takes the path corresponding to the maximum value in the P updated path metric values of each state node at the nth code element moment as a survival path, takes the path with the maximum path metric value in the survival paths of all the state nodes as a temporary maximum likelihood path, backtracks the temporary maximum likelihood path, and obtains a demodulation code element which is the reference demodulation code element Ctemp＝{ci|1≤i≤n}；
(4e) Judging whether N is true or not, if so, selecting the path metric value with the maximum value from the survivor paths of the Nth code element time as the maximum likelihood path, backtracking the maximum likelihood path, and obtaining the second Viterbi demodulation result C which is { C ═ CnI 1 is not less than N and not more than N, i.e. the demodulation result of the TCM-CPM signal under the multipath channel, otherwise N is equal to N +1, and execute the step(4b)；
Compared with the prior art, the invention has the following advantages:
the invention firstly carries out the first Viterbi demodulation on the frequency deviation compensation signal R through the demodulator, and takes the result of the first Viterbi demodulation as a reference signal, carries out the second Viterbi demodulation on the frequency deviation compensation signal R, and in the process of calculating the branch measurement by the second Viterbi demodulation, takes the demodulated code element, the current demodulated code element and the modulation signal corresponding to the first demodulation result as the reference signal to carry out convolution with the channel gain value, thereby being capable of furthest inhibiting the influence of multipath interference on the demodulation result.
Drawings
FIG. 1 is a flow chart of an implementation of the present invention.
FIG. 2 is a comparison graph of error rate curves for TCM-CPM signal demodulation with channel parameters of [0.1,0.98,0.66] according to the present invention and the prior art.
FIG. 3 is a comparison graph of error rate curves for TCM-CPM signal demodulation with channel parameters of [0.2,0.98,0.66] according to the present invention and the prior art.
FIG. 4 is a comparison graph of error rate curves for TCM-CPM signal demodulation with channel parameters of [0.3,0.98,0.66] according to the present invention and the prior art.
Detailed Description
The invention is described in further detail below with reference to the figures and specific examples.
Referring to fig. 1, the present invention includes the steps of:
the embodiment includes two parts, namely a transmitting end and a receiving end. The sending end comprises an information source and a TCM-CPM modulator, and the receiving end comprises a frequency deviation estimator and a demodulator.
Step 1) the TCM-CPM modulator modulates the binary sequence:
(1a) the TCM-CPM modulator at the transmitting end adds a locally stored M-bit pilot sequence to the front of a binary sequence generated by the source to form an N-bit sequence X to be modulated, where M is 53 and N is 1059 in this embodiment.
(1b) TCM-CPM modulation of transmitting endThe device carries out TCM-CPM digital modulation on a sequence X to be modulated to obtain a discrete TCM-CPM modulation signal S ═ Sk|1≤k≤NNsAnd passes S through NcA multipath channel is transmitted to the receiving end, where skDenotes the kth sample point, N denotes the number of symbols contained in S, NsIndicating the number of samples contained in each symbol, the first MN in SsOne sample point is a pilot signal, and the rest (N-M) NsEach sample point being a data signal, Nc＝3,Ns＝8；
The kth sample point skThe TCM-CPM digital modulation formula is as follows:
Figure BDA0003332817920000051
Figure BDA0003332817920000052
Figure BDA0003332817920000053
Figure BDA0003332817920000054
where T is the symbol period, E is the symbol energy, fcIs the carrier frequency, TsAt intervals of sample points, Ts＝T/Ns，α＝(αnN is more than or equal to 1 and less than or equal to N) is a J system information symbol sequence mapped by TCM coding, alphaiIs an element of alpha, phi (kT)sα) is a phase function, θnDenotes the accumulated phase of the nth symbol, k ═ nNs+m,m∈{1,2,...,NsH denotes the modulation index, L denotes the phase constraint length, q (kT)s) Is a phase pulse.
Step 2), the frequency offset estimator performs frequency offset compensation on the modulation signal:
the frequency deviation estimator at the receiving end receives S through NcA multipath interference signal Y formed after a multipath channel is formed, a rotating average periodogram RPA method is adopted, frequency offset estimation is carried out on Y through a pilot signal in Y, frequency offset compensation is carried out on Y through a frequency offset value delta f obtained by frequency offset estimation, and then a frequency offset compensation signal R of Y is sent to a demodulator, wherein Y is { Y ═ after frequency offset compensation is carried out on Y through a frequency offset value delta f obtained by frequency offset estimationk|1≤k≤NNs}，R＝{rk|1≤k≤NNs}，ykDenotes skSamples after multipath interference, rkDenotes ykSampling points after frequency offset compensation;
wherein, the kth sampling point y after multipath interferencekThe formula of the frequency offset compensation is as follows:
Figure BDA0003332817920000061
where e is a natural constant and j represents an imaginary unit.
The purpose of frequency offset compensation is mainly to eliminate doppler shift caused by mutual movement of a transmitting end and a receiving end, and frequency offset estimation is necessary because the existence of doppler shift can cause the increase of demodulation error rate and the deterioration of demodulation performance.
Step 3), the demodulator performs first Viterbi demodulation on the frequency offset compensation signal R:
(3a) initialization parameters of a demodulator at a receiving end: the initialization symbol time n is 1, the initialization demodulator trellis diagram comprises V state nodes, and the fixed initial state node is
Figure BDA0003332817920000062
The path metric value of each state node is 0, and a reference demodulation code element is initialized
Figure BDA0003332817920000063
(3b) The demodulator at the receiving end calculates P branch metrics entering each state node at the nth symbol time, wherein the slave state node
Figure BDA0003332817920000064
Transition to a State node
Figure BDA0003332817920000065
Branch metric of
Figure BDA0003332817920000066
The calculation method comprises the following steps:
Figure BDA0003332817920000067
wherein P is more than or equal to 2, cnIs a state node at the time of the (n-1) th symbol
Figure BDA0003332817920000068
State node transition to nth symbol time
Figure BDA0003332817920000069
The corresponding demodulated symbols are then transmitted to the receiver,
Figure BDA00033328179200000610
the v' th state node of the demodulator trellis at the n-1 th symbol time,
Figure BDA00033328179200000611
n is more than or equal to 1 and less than or equal to N, V is more than or equal to 1, V' is less than or equal to V, r is the V-th state node of the demodulator grid diagram at the nth code element momentkIs the m-th sample of the N-th symbol, k ═ N-1Ns+m，1≤m≤Ns，
Figure BDA00033328179200000612
Is StempOne of the plurality of sampling points is selected,
Figure BDA00033328179200000613
representing the channel gain estimate and the reference signal SrefThe convolution of (a) with (b) is performed,
Figure BDA0003332817920000071
is an estimate of the channel gain, jmaxIs a modulus valueMaximum rhojCorresponding subscript, SrefRepresents { Ctemp,cnA corresponding TCM-CPM modulation signal;
in the process of calculating branch metric, the estimated value of channel gain and reference signal S are usedrefConvolution is performed so that the effect of the symbols before the nth symbol instant is eliminated, but due to the reference signal SrefThere is no code element behind the nth code element time, so the influence of the following code element time can not be completely eliminated, the multipath interference can not be completely eliminated, the calculation of the branch measurement is not the most accurate, thereby the increase of the error rate is caused, and the demodulation performance is influenced;
(3c) a demodulator at a receiving end respectively accumulates P branch metrics entering each state node at the nth code element time to the path metric of a transition-out state node corresponding to the (n-1) th code element time through the calculated P branch metrics, so as to obtain P updated path metric values of each state node at the nth code element time;
(3d) the demodulator at the receiving end takes the path corresponding to the maximum value in the P updated path metric values of each state node at the nth code element moment as a survival path, takes the path with the maximum path metric value in the survival paths of all the state nodes as a temporary maximum likelihood path, backtracks the temporary maximum likelihood path, and obtains a demodulation code element which is the reference demodulation code element Ctemp＝{ci|1≤i≤n}；
(3e) Judging whether N is true or not, if so, selecting the path metric value with the maximum value from the survivor paths of the Nth code element time as the maximum likelihood path, and backtracking the maximum likelihood path to obtain the first Viterbi demodulation result Cref＝{c'nIf not, making N equal to N +1, and executing the step (3 b);
step 4), the demodulator performs second Viterbi demodulation on the frequency offset compensation signal R:
(4a) the demodulator at the receiving end carries out initialization: the initialization symbol time n is 1, the initialization demodulator trellis diagram comprises V state nodes, and the fixed initial state node is
Figure BDA0003332817920000072
The path metric value of each state node is 0, and a reference demodulation code element is initialized
Figure BDA0003332817920000073
Because the multi-path interference cannot be completely eliminated in the first demodulation process, the result of the first demodulation is not necessarily the most accurate, and in order to reduce the error rate, the result of the first demodulation is used as a reference signal so as to eliminate the multi-path interference;
(4b) the demodulator at the receiving end calculates P branch metrics entering each state node at the nth symbol time, wherein the slave state node
Figure BDA0003332817920000074
Transition to a State node
Figure BDA0003332817920000075
Branch metric of
Figure BDA0003332817920000076
The calculation method comprises the following steps:
Figure BDA0003332817920000077
wherein the content of the first and second substances,
Figure BDA0003332817920000081
is S'tempOne of the plurality of sampling points is selected,
Figure BDA0003332817920000082
representing the channel gain estimation value and the reference signal S'refThe convolution of (a) with (b) is performed,
Figure BDA0003332817920000083
is an estimate of the channel gain, jmaxIs ρ of maximum modulus valuejCorresponding subscript, S'refRepresents { Ctemp,cn,C′refThe corresponding TCM-CPM modulation signal，C′refAs a first demodulation result CrefThe symbol following the nth symbol;
reference signal S 'in the calculation of branch metrics'refIs composed of three parts, respectively { Ctemp,cn,C′refConvolution is carried out on the estimated value of the channel gain, so that the influence of the front and rear code elements can be completely eliminated, the multipath interference is eliminated, and the error rate is reduced;
(4c) a demodulator at a receiving end respectively accumulates P branch metrics entering each state node at the nth code element moment into the path metric of a transition state node corresponding to the (n-1) th code element moment to obtain P updated path metric values of each state node at the nth code element moment;
(4d) a demodulator at a receiving end takes a path corresponding to the maximum value in the P updated path metric values of each state node at the nth code element moment as a survivor path, takes the path with the maximum path metric value in the survivor paths of all the state nodes as a temporary maximum likelihood path, backtracks the temporary maximum likelihood path, and obtains a demodulation code element which is a reference demodulation code element Ctemp＝{ci|1≤i≤n}；
(4e) Judging whether N is true or not, if so, selecting the path metric value with the maximum value from the survivor paths of the Nth code element time as the maximum likelihood path, backtracking the maximum likelihood path, and obtaining the second Viterbi demodulation result C which is { C ═ CnAnd l 1 is less than or equal to N and less than or equal to N, namely, a demodulation result of the TCM-CPM signal in the multipath channel, otherwise, N is equal to N +1, and step (4b) is performed.
The technical effects of the invention are further explained by simulation experiments as follows:
1. simulation conditions and contents:
the simulation uses MATLAB R2017a simulation software, channel parameters are respectively [0.1,0.98,0.66], [0.2,0.98,0.66] and [0.3,0.98,0.66], and only the parameter of the first path is changed because the first path is corresponding to the influence of the code element at the time after the current code element after convolution. The number of simulations was 25000.
The bit error rate comparison simulation of the TCM-CPM signal demodulation is performed when the channel parameters are [0.1,0.98,0.66], [0.2,0.98,0.66], [0.3,0.98,0.66] for the present invention and the conventional demodulation method of TCM-CPM signals, and the results are shown in fig. 2, fig. 3, and fig. 4.
2. And (3) simulation result analysis:
referring to fig. 2, the signal-to-noise ratio is plotted on the abscissa in dB and the bit error rate is plotted on the ordinate. As can be seen from FIG. 2, the bit error rate of the present invention is relatively close to that of the prior art demodulation when the SNR is low, and the bit error rate advantage of the present invention is gradually reflected when the SNR is gradually increased, and the bit error rate is 10-5The level performance is improved by about 0.3 dB. As can be seen from fig. 3, the bit error rate of the present invention is 10-5The level performance is improved by about 0.8 dB. As can be seen from fig. 3, the bit error rate of the present invention is 10-5The level performance is improved by about 1.2 dB. Comparing fig. 2, fig. 3 and fig. 4, it can be seen that, when the channel parameter before the main path is larger, i.e. the parameter of the first path in the simulation process is larger, the error rate reduction of the present invention is larger compared with the prior art.

Claims (3)

1. A demodulation method of TCM-CPM signals under a multipath channel is characterized by comprising the following steps:
(1) the TCM-CPM modulator modulates the binary sequence:
(1a) a TCM-CPM modulator at a sending end adds a locally stored M-bit pilot frequency sequence in front of a binary sequence generated by an information source to form an N-bit sequence X to be modulated, wherein M is more than or equal to 1 and less than N, and N is more than or equal to 2;
(1b) a TCM-CPM modulator at a sending end carries out TCM-CPM digital modulation on a to-be-modulated sequence X to obtain a discrete TCM-CPM modulation signal S ═ Sk|1≤k≤NNsAnd S is passed through NcA multipath channel is transmitted to the receiving end, where skDenotes the kth sample point, N denotes the number of symbols contained in S, NsIndicating the number of samples contained in each symbol, the first MN in SsOne sample point is a pilot signal, and the rest (N-M) NsEach sample point being a data signal, Nc≥2；
(2) And the frequency offset estimator performs frequency offset compensation on the modulation signal:
the frequency deviation estimator at the receiving end receives S through NcA multipath interference signal Y formed after a multipath channel is formed, a rotating average periodogram RPA method is adopted, frequency offset estimation is carried out on Y through a pilot signal in Y, frequency offset compensation is carried out on Y through a frequency offset value delta f obtained by frequency offset estimation, and then a frequency offset compensation signal R of Y is sent to a demodulator, wherein Y is { Y ═ after frequency offset compensation is carried out on Y through a frequency offset value delta f obtained by frequency offset estimationk|1≤k≤NNs}，R＝{rk|1≤k≤NNs}，ykDenotes skSamples after multipath interference, rkDenotes ykSampling points after frequency offset compensation;
(3) the demodulator performs a first viterbi demodulation on the frequency offset compensation signal R:
(3a) initialization parameters of a demodulator at a receiving end: the initialization symbol time n is 1, the initialization demodulator trellis diagram comprises V state nodes, and the fixed initial state node is
Figure FDA0003639672500000011
The path metric value of each state node is 0, and a reference demodulation code element is initialized
Figure FDA0003639672500000012
(3b) The demodulator at the receiving end calculates P branch metrics entering each state node at the nth symbol time, wherein the slave state node
Figure FDA0003639672500000013
Transition to State node
Figure FDA0003639672500000014
Branch metric of
Figure FDA0003639672500000015
The calculation formula of (c) is:
Figure FDA0003639672500000016
wherein P is more than or equal to 2, cnIs a state node at the time of the (n-1) th symbol
Figure FDA0003639672500000017
State node transition to nth symbol time
Figure FDA0003639672500000021
The corresponding demodulated symbols are then transmitted to the receiver,
Figure FDA0003639672500000022
for the v' state node of the demodulator trellis at the n-1 symbol time,
Figure FDA0003639672500000023
n is more than or equal to 1 and less than or equal to N, V is more than or equal to 1, V' is less than or equal to V, r is the V-th state node of the demodulator grid diagram at the nth code element momentkIs the m-th sample of the N-th symbol, k ═ N-1Ns+m，1≤m≤Ns，
Figure FDA0003639672500000024
Is StempOne of the plurality of sampling points is selected,
Figure FDA0003639672500000025
representing the channel gain estimate and the reference signal SrefThe convolution of (a) with (b) is performed,
Figure FDA0003639672500000026
is an estimate of the channel gain, jmaxIs greatest in modulus
Figure FDA0003639672500000027
Corresponding subscript, SrefRepresents { Ctemp,cnA corresponding TCM-CPM modulation signal;
(3c) a demodulator at a receiving end respectively accumulates P branch metrics entering each state node at the nth code element moment into the path metric of a transition state node corresponding to the (n-1) th code element moment to obtain P updated path metric values of each state node at the nth code element moment;
(3d) the demodulator at the receiving end takes the path corresponding to the maximum value in the P updated path metric values of each state node at the nth code element moment as a survival path, takes the path with the maximum path metric value in the survival paths of all the state nodes as a temporary maximum likelihood path, backtracks the temporary maximum likelihood path, and obtains a demodulation code element which is the reference demodulation code element Ctemp＝{ci|1≤i≤n}；
(3e) Judging whether N is true or not, if so, selecting the path metric value with the maximum value from the survivor paths of the Nth code element time as the maximum likelihood path, and backtracking the maximum likelihood path to obtain the first Viterbi demodulation result Cref＝{c'nIf not, making N equal to N +1, and executing the step (3 b);
(4) the demodulator performs a second viterbi demodulation on the frequency offset compensation signal R:
(4a) the demodulator at the receiving end carries out initialization: the initialization symbol time n is 1, the initialization demodulator trellis diagram comprises V state nodes, and the fixed initial state node is
Figure FDA0003639672500000028
The path metric value of each state node is 0, and a reference demodulation code element is initialized
Figure FDA0003639672500000029
(4b) The demodulator at the receiving end calculates P branch metrics entering each state node at the nth symbol time, wherein the slave state node
Figure FDA00036396725000000210
Transition to a State node
Figure FDA00036396725000000211
Branch metric of
Figure FDA00036396725000000212
The calculation formula of (2) is as follows:
Figure FDA00036396725000000213
wherein, s "k+jmax-1∈S’tempIs S'tempOne of the plurality of sampling points is selected,
Figure FDA00036396725000000214
representing the channel gain estimation value and the reference signal S'refThe convolution of (a) with (b) is performed,
Figure FDA00036396725000000215
is an estimate of the channel gain, jmaxIs greatest in modulus value
Figure FDA00036396725000000216
Corresponding subscript, S'refRepresents { Ctemp,cn,C'ref} corresponding TCM-CPM modulation signals, C'refAs a first demodulation result CrefThe symbol following the nth symbol;
(4c) a demodulator at a receiving end respectively accumulates P branch metrics entering each state node at the nth code element moment into the path metric of a transition state node corresponding to the (n-1) th code element moment to obtain P updated path metric values of each state node at the nth code element moment;
(4d) the demodulator at the receiving end takes the path corresponding to the maximum value in the P updated path metric values of each state node at the nth code element moment as a survival path, takes the path with the maximum path metric value in the survival paths of all the state nodes as a temporary maximum likelihood path, backtracks the temporary maximum likelihood path, and obtains a demodulation code element which is the reference demodulation code element Ctemp＝{ci|1≤i≤n}；
(4e) Judging whether N is equal to N or notIf yes, selecting the path metric value with the maximum value from the survivor paths of the Nth code element time as the maximum likelihood path, and backtracking the maximum likelihood path to obtain the second time Viterbi demodulation result C ═ CnAnd l 1 is less than or equal to N and less than or equal to N, namely, a demodulation result of the TCM-CPM signal in the multipath channel, otherwise, N is equal to N +1, and step (4b) is performed.
2. The method as claimed in claim 1, wherein the TCM-CPM modulator at the transmitting end performs TCM-CPM digital modulation on the sequence X to be modulated by the TCM-CPM modulator at the transmitting end, wherein the kth sample s iskThe TCM-CPM digital modulation formula is as follows:
Figure FDA0003639672500000031
Figure FDA0003639672500000032
Figure FDA0003639672500000033
Figure FDA0003639672500000034
where T is the symbol period, E is the symbol energy, fcIs the carrier frequency, TsAt intervals of sample points, Ts＝T/Ns，α＝(αnN is more than or equal to 1 and less than or equal to N) is a J system information symbol sequence after TCM coding mapping, alphaiIs an element of alpha, phi (kT)sα) is a phase function, θnDenotes the accumulated phase of the nth symbol, k ═ nNs+m,m∈{1,2,...,NsH denotes the modulation index, L denotes the phase constraint length, q (kT)s) Is a phase pulse.
3. The method as claimed in claim 2, wherein the step (2) performs frequency offset compensation on Y using the frequency offset value Δ f obtained by frequency offset estimation, wherein the kth sample Y after multipath interference is obtainedkThe formula of the frequency offset compensation is as follows:
Figure FDA0003639672500000041
where e is a natural constant and j represents an imaginary unit.
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Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1901509A1 (en) * 2006-09-14 2008-03-19 BRITISH TELECOMMUNICATIONS public limited company M-GFSK receiver with a Viterbi-decoder having reduced trellis states
CN105429729A (en) * 2015-11-03 2016-03-23 西安烽火电子科技有限责任公司 Ultrashort wave CPTCM (Continuous Phase Trellis Coded Modulation) coherent demodulation method
CN105763500A (en) * 2014-12-20 2016-07-13 西安飞东电子科技有限责任公司 Frequency deviation, time delay and phase deviation combined synchronization method of continuous phase modulation signals

Family Cites Families (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7620881B2 (en) * 2005-03-09 2009-11-17 Harris Corporation System and method for communicating data using iterative equalizing and decoding and recursive inner code

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1901509A1 (en) * 2006-09-14 2008-03-19 BRITISH TELECOMMUNICATIONS public limited company M-GFSK receiver with a Viterbi-decoder having reduced trellis states
CN105763500A (en) * 2014-12-20 2016-07-13 西安飞东电子科技有限责任公司 Frequency deviation, time delay and phase deviation combined synchronization method of continuous phase modulation signals
CN105429729A (en) * 2015-11-03 2016-03-23 西安烽火电子科技有限责任公司 Ultrashort wave CPTCM (Continuous Phase Trellis Coded Modulation) coherent demodulation method

Non-Patent Citations (4)

* Cited by examiner, † Cited by third party
Title
Concatenated trellis coding and full response CPM for wireless DS/SSMA communications;Fengfan Yang等;《Gateway to 21st Century Communications Village. VTC 1999-Fall. IEEE VTS 50th Vehicular Technology Conference》;20020806;第5卷;第2576-2580页 *
TCM-CPM调制解调技术的研究;张荣波;《中国优秀硕士学位论文全文数据库信息科技辑》;20180415(第4期);第I136-244页 *
低信噪比下跳频突发传输系统关键技术研究;张立增;《中国优秀硕士学位论文全文数据库信息科技辑》;20210515(第5期);第I136-438页 *
低复杂度的跳频连续相位调制传输技术研究;孙锦华;《中国博士学位论文全文数据库信息科技辑》;20090815(第8期);第I135-43页 *

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