CN113644855B

CN113644855B - High-frequency converter

Info

Publication number: CN113644855B
Application number: CN202110812309.0A
Authority: CN
Inventors: 钟瀚中; 谭享波
Original assignee: Shenzhen Viking Drive Co ltd
Current assignee: Shenzhen Viking Drive Co ltd
Priority date: 2021-06-07
Filing date: 2021-07-17
Publication date: 2024-04-02
Anticipated expiration: 2041-07-17
Also published as: CN113644855A

Abstract

The invention discloses a high-frequency converter, which is oriented to high-performance application occasions of permanent magnet motors such as high-speed grinding machines, fuel cell compressors, PCB drilling, automobile main driving and the like, and high precision or high reliability; the control targets are high speed, high precision, low loss and low vibration; the type of motor which can be driven is a three-phase direct current brushless, permanent magnet synchronous and alternating current asynchronous motor, a magnetic field directional control algorithm is adopted, PG is arranged, and the position, speed or torque control can be realized; the invention has no position sensor, no PG and can realize speed control, and the invention has originality; obvious advantages and wide application.

Description

High-frequency converter

Technical Field

The present invention relates to a high frequency converter.

Background

The high-frequency converter and the main shaft are core functional components of equipment such as a medium-high end numerical control grinding machine, a compressor, a fan, a PCB (printed Circuit Board) drilling, a molecular pump and the like, a motor/main shaft driving system with high rotating speed, high efficiency, high reliability and low vibration is one of key technologies for ensuring performance indexes such as precision, efficiency, service life and the like of equipment processing, the output frequency of the frequency converter on the market is generally lower than 600Hz (the high-frequency converter provided by the invention is a frequency converter with the output frequency higher than 600 Hz), the rotating speed of a corresponding pair of motors/main shafts is lower than 3.6 ten thousand revolutions per minute, and if the rotating speed is required to be higher, the motor/main shaft driving system can be realized by adding a speed increasing box in a traditional mechanical transmission link, and the defects of high cost of purchasing a machine, high running cost, high maintenance cost, large volume and the like exist.

In order to realize high-performance control on a motor/main shaft with high rotating speed, high efficiency, high reliability, low vibration and the like, a matched driving system in the prior scheme is required to adopt a high-performance vector frequency converter, namely a mechanical position sensor is used for collecting rotor position and rotating speed data, a high-performance MCU is realized by a magnetic field directional control (FOC) algorithm, but the installation of the position sensor leads to the defects of increased system complexity, reduced reliability, high-speed fault rate, high use cost and the like, and the invention adds a non-inductive control algorithm on the basis of the prior inductive control for a motor high-speed operation occasion (more than 600 Hz), and the basic principle is that: the permanent magnet motor is regarded as a sensor, the three-phase input voltage of the motor is regarded as excitation of the sensor, the three-phase feedback current of the motor is regarded as sensor response, the rotor position is accurately calculated on line through an algorithm, so that the magnetic field orientation control is realized, the speed high-precision control of the permanent magnet motor is realized under the condition that a mechanical position sensor is not used, the production, assembly and use costs of the motor are greatly reduced, and meanwhile, the popularization technical threshold of the permanent magnet motor is reduced.

In a motor driving system, the PWM carrier frequency of a frequency converter must be higher than 30 times of the output frequency of the frequency converter, otherwise, indexes such as vibration, efficiency, heating and the like of the motor can rise sharply, the operation speed of an MCU in FOC control must be synchronous with the PWM carrier frequency, namely, the MCU must complete an SVC algorithm once in a PWM period, for example, the frequency converter outputs 2500Hz, the PWM carrier frequency is 80kHz, the corresponding time interval is 1.25us, more than 30us is required for one SVC algorithm according to the operation capacity of the general MCU in the existing market, and the requirement of the high-frequency driver on the operation capacity of a controller cannot be met, which is one of the important reasons that the output frequency of the general variable frequency driver in the current market is limited to be below 500-600 Hz.

Disclosure of Invention

The invention aims to solve the technical problem of providing a high-frequency converter which is easy to implement and can obviously improve the rotating speed of a motor.

The technical proposal of the invention is as follows:

a high-frequency converter comprises a controller, a signal acquisition circuit and a high-frequency inverter bridge; the signal acquisition circuit is connected with the controller; the controller is connected with the high-frequency inverter bridge through an isolation circuit;

the controller is provided with a core module; the core module comprises an MCU minimum system and a peripheral circuit; the method can be realized through an IP card or by a GPU; the controller is also provided with a soft core control module; the soft core control module comprises a finite state machine FSM, an ADC synchronous acquisition unit (ADC interface), a Clarke conversion unit, a Park conversion unit, a counter electromotive force observer (ABobsrv), a speed filter (GetVel), a speed loop controller (vPi), a current loop controller (dPi/qPi), a Park inverse transformation unit (invPark), a Clarke inverse transformation unit (invClarke) and a central symmetry vector PWM modulator (SVM); the controller sends out driving pulse through the centrosymmetric vector PWM modulator to drive the inverter to work.

Finite state machine FSM: in a control period, the FSM core firstly controls the ADC synchronous acquisition unit to acquire feedback data required by the SVC algorithm, then controls the SVC algorithm peripheral to execute the SVC algorithm in a chip according to the sequence from top to bottom, and finally controls the driving of the off-chip high-frequency inverter bridge by the output control pulse of the SVM core, and repeatedly executes the steps in the next period; the analog quantity that ADC synchronous acquisition unit gathered includes: three-phase current i _a/b/c Bus voltage V _DC Temperature Temp of motor _MOTOR Temperature Temp of frequency converter base _VFD And the analog input of the upper computer is 0-10V/4-20mA; the analog quantity input of the upper computer is 0-10V/4-20mA, and is used for receiving control signals of the upper computer (industrial personal computer, PLC and the like);

the Clarke transformation unit is used for executing the equivalent Clarke transformation of the amplitude to obtain i _α 、i _β

Clarke transform, positive transform is current, inverse transform is voltage; wherein:

i _A 、i _B and i _C Respectively three-phase currents, i.e. corresponding to i _a/b/c ；i _α 、i _β Two-phase current of an alpha-beta coordinate system of the motor; the Park conversion unit is used for converting the rotor position theta of the previous control period _r Performing coordinate axis rotation operation to obtain an alternating-axis and direct-axis current component i _q 、i _d ；

A back electromotive force observer (abossrv) for obtaining back electromotive force;

the transfer function of the back emf observer is:wherein->Is the output of the back emf observer; g (z) is a digital model of a motor single-phase winding, the input quantity of the digital model is the difference between input phase voltage and motor back electromotive force, and the output quantity is phase current: />L, R is inductance and resistance of the motor stator phase winding, and Tp is SVPWM period; d (z) is the viewer controller: />K _p ＝2ξω ₀ L-R，K _I ＝ω ₀ LT _s ，ξ、ω ₀ The damping ratio and the undamped oscillation frequency of the counter electromotive force observer are respectively; e (z) is the z-transform of the phase back EMF; kp, KI are the proportional and integral coefficients Kp, KI is the prior art, and its values are configured according to the motor winding parameters

The speed filter (GetVel) is used to obtain the speed dθ (n) at the nth sampling point, and the calculation formula is: to (3) the point;

dθ (n) =θ (n) - θ (n-1), θ (n) and θ (n-1) are rotor positions at times n and n-1, respectively; a speed loop controller (vPi) for setting a given value Ref as a speed command, a feedback value Fb as a feedback speed, and an output Out as a q-axis command current i in a vPi speed loop _q ；i _q Is dependent on the rated current of the motor; current loop controller (dPi/qPi): dPi current loop controller, given value Ref of 0, feedback value Fb of feedback current i _d Output is V _d The method comprises the steps of carrying out a first treatment on the surface of the qPi current loop controller, given value Ref is speed loop output i _q The feedback value Fb is the feedback current i _q Output is V _q The method comprises the steps of carrying out a first treatment on the surface of the The amplitude of the two current loop controller outputs Vd, vq must satisfy:vPi, dPi, qPi are identical IP cores, the functions to be implemented are: a Pi controller resistant to integral saturation;

an Park inverse transform unit (invPark) for Park inverse transform, i.e. transforming the d-q axis to the alpha-beta axis; the Park inverse transformation is the prior art, and the specific formulas areV output by dPi, qPi controller _d 、v _q Transforming from a rotor coordinate system to a stator coordinate system;

inverse clarke transform (invClarke) is used for inverse clarke transform, equivalently converting the alpha-beta axis voltage vector into an a-b-c three-phase coordinate system;vdc is the dc bus voltage and Tp is the SVPWM period.

A centrosymmetric vector PWM modulator (SVM) is state of the art, see links:

https://wenku.baidu.com/view/9b51ae394531b90d6c85ec3a87c24028915f85 22.htmlSVPWM is a state of the art.

Calculation logic: the process of generating va, vb and vc according to the output result vd, vq of the controller is thatObtain v _α ,v _β

Also, according to va+vb+vc=0, va, vb, vc are obtained

And obtaining the SVPWM saddle-shaped PWM pulse width Tu, tv and Tw by an inversion matrix mode.

The high-frequency converter also comprises a Modbus-RTU protocol IP core for high-speed bus control, and the Modbus-RTU protocol IP core is in butt joint with a touch display screen, a PC or a PLC.

The high-frequency converter also comprises a permanent magnet motor SVC IP core and a permanent magnet motor non-inductive vector control (SVC) for determining the rotor position of the permanent magnet motor without a position sensor.

The controller is communicated with the upper computer through a serial port. For example, the UART1 interface of the controller communicates with the PC through the RS232 protocol.

A drive control method of high frequency converter, based on the aforesaid high frequency converter; the control method is a non-inductive vector control method;

the method designs an FPGA-based Soc (System on chip) as a main control of the high-frequency converter, and an FOC algorithm (FOC-Field Oriented Control, also called vector control) and a SVC (Sensorless Vector Control) algorithm of a motor control algorithm are realized by adopting an IP soft core.

The processes of starting, running, stopping and parameter self-adaptive adjustment of the motor operation are independently completed by the soft core control modules, the bus is not occupied, no interruption is caused, and the technical route can enable the Soc system to process the SVC algorithm at the speed of 120kHz at maximum in real time.

The beneficial effects are that:

the invention provides a high-frequency converter which is oriented to high-performance application occasions of permanent magnet motors such as high-speed grinding machines, fuel cell compressors, PCB drilling, automobile main driving and the like, and has high precision or high reliability; the control targets are high speed, high precision, low loss and low vibration; the experimental result can prove that the high-frequency converter can realize high speed, high precision, low loss and low vibration, the types of the motor which can be driven are three-phase direct current brushless, permanent magnet synchronous and alternating current asynchronous motors, a magnetic field directional control algorithm is adopted, PG is arranged, and the position, speed or torque control can be realized; no position sensor and no PG are provided, and speed control can be realized.

The core of the invention is to design a high-speed controller based on IP soft core

The invention provides a structural design method of an integrally formed product, which is used for isolating, radiating and shielding.

The high-frequency converter can be used for three-phase permanent magnet synchronous motors, direct current brushless motors and alternating current asynchronous motors, and adopts the FOC algorithm.

The high-frequency converter adopts non-inductive vector control, the permanent magnet motor is regarded as a sensor, the input voltage of the motor is regarded as excitation of the sensor, the feedback current of the motor is regarded as sensor response, the rotor position is accurately calculated on line through an algorithm, so that the magnetic field orientation control is realized, and the speed high-precision control of the permanent magnet motor is realized under the condition that a mechanical position sensor is not used.

The high-frequency converter adopts Soc (System on chip) based on FPGA as main control of the high-frequency converter, and the FOC algorithm and the SVC (Sensorless Vector Control) algorithm of the motor control algorithm are realized by adopting IP soft cores.

The invention provides a high-frequency direct-drive scheme, the highest output frequency can reach 2500Hz and 8000kHz respectively when a frequency converter drives a permanent magnet motor and an asynchronous motor, the rotating speeds of a corresponding pair of motors are 15 ten thousand and 48 ten thousand revolutions per minute respectively.

The invention develops a complete Hardware Description Language (HDL) -based algorithm SVC (Sensorless Vector Control), a hardware platform is based on a very large scale integrated circuit, each functional unit of a control flow is converted into an independent chip by a modularized design method, and an IP soft core is used for realizing SVC algorithm control, so that the bottleneck problem of MCU computing capacity is solved.

Drawings

FIG. 1 is a block diagram of the overall structure of a high frequency converter;

FIG. 2 (a) is a schematic diagram of the analog control of the high frequency converter;

FIG. 2 (b) is a schematic diagram of digital quantity control of the high frequency converter;

FIG. 2 (c) is a schematic diagram of a high frequency converter bus control;

FIG. 2 (d) is a schematic diagram of a high frequency transducer servo control;

FIG. 3 is a schematic diagram of a Soc-based high frequency converter system;

FIG. 4 is a centrosymmetric vector PWM timing diagram;

FIG. 5 is a block diagram of an anti-integral saturation PI controller;

FIG. 6 is a single phase winding circuit of the stator;

FIG. 7 is a signal flow diagram of a single phase winding of a stator;

fig. 8 is a back emf viewer;

fig. 9 is a back electromotive force detection equivalent flowchart;

FIG. 10 is a block diagram of a general master control board system for a high frequency converter;

FIG. 11 is a block diagram of a high frequency drive plate structure;

FIG. 12 is a three-phase current measured waveform;

FIG. 13 is a commanded pulse width measured waveform;

FIG. 14 is a three-phase voltage waveform;

fig. 15 is a back emf viewer alpha axis input and output synchronization waveforms;

fig. 16 shows the back emf and the phase detection result in the α - β coordinate system;

FIG. 17 is an alpha-axis reverse motor and current waveform;

FIG. 18 is a waveform diagram of the input and output of the speed controller during the acceleration phase;

FIG. 19 is a control accuracy and speed controller output during high speed steady state operation;

fig. 20 is a torque waveform display at the acceleration/deceleration stage of the high-frequency converter.

Description of the variables in the figures and in the text of the description:

u (t): motor phase voltages; u(s), U (z): laplace transform, z transform of phase voltages;

i (t): motor phase current; s, z is Laplace operator, z transform operator; i(s), I (z): laplace transform, z transform of phase current; e (t): a motor phase back electromotive force; e(s), E (z): laplacian transform, z transform of phase back electromotive force; l, R equivalent inductance and resistance of the motor stator; g (z): an ideal digital model of a single-phase winding, wherein phase voltage and phase counter electromotive force are input, phase current is output;a G (z) approximation to an idealized model; t (T) _p PWM carrier wave period; t (T) _u/v/w U, v, w phase PWM instruction pulse width; t (T) _DB PWM dead time; u (U) _T 、V _T 、W _T U, v, w phase inverter bridge upper arm switch signals, controller chip pins; u (U) _B 、V _B 、W _B U, v, w phase inversion bridge lower arm switch signals, controller chip pins; FOC: field Oriented Control (magnetic field orientation control); SVC: sensorless Vector control (dead vector control); z, a digital system delay algorithm; kp is a controller proportion parameter; ki, integrating parameters by a controller; kc, controller anti-saturation coefficient; ref, a controller instruction reference value; fb: the controller inputs a feedback value; out: controller inputDischarging; v _α 、v _β Two-phase voltages of an alpha-beta coordinate system; v _a 、v _b 、v _c A, b, c three-phase voltage; i.e _α 、i _β Two-phase current of an alpha-beta coordinate system; i.e _a 、i _b 、i _c A, b, c three-phase current; i.e _d 、i _q D-q coordinate system two-phase current; v _d 、v _q D-q coordinate system two-phase voltage; e, e _α 、e _β Alpha-beta coordinate system two-phase back electromotive force; θ _r : rotor position.

Detailed Description

The invention will be described in further detail with reference to the accompanying drawings and specific examples:

1. the invention provides a high-frequency converter for high-speed motor technology driving, which replaces a general DSP/ARM with an FPGA, designs a control method based on Soc (System on chip) as a main control of the high-frequency converter, develops a complete algorithm based on HDL (hardware description language) to realize SVC (Sensorless Vector Control), uses an IP soft-check to realize a motor control algorithm, improves the operation speed of a torque ring from 20kHz to 100kHz, correspondingly improves the highest output frequency of the frequency converter from 600Hz to 2500Hz, and can realize high-precision, high-efficiency and low-vibration speed control of a direct-current brushless motor and a magnetic synchronous motor.

2. The invention adopts a magnetic field orientation control algorithm to realize high-performance control of a permanent magnet motor, and the 'orientation' aims at solving the problem of accurate detection of the rotor position at first. Second kind: a digital model of the motor is built without a position sensor, the permanent magnet motor is regarded as a sensor, the input voltage of the motor is regarded as excitation of the sensor, the feedback current of the motor is regarded as sensor response, and the position of the rotor is accurately calculated on line through an advanced algorithm; the method has the advantages of high precision, high reliability and low cost, but the motor cannot work in a torque control mode and a position control mode.

3. Aiming at the problems of heat dissipation, electromagnetic interference, strong and weak point isolation and compatibility among product families of the high-frequency converter, the invention provides a structural design method for an integrally formed product, which is used for isolating, heat dissipation, shielding and integral forming.

Example 1:

as shown in FIG. 1, the high-speed motor is mostly a nonstandard product, the high-frequency converter adopts direct-current power supply input, the voltage range which can be input is set to be 24-400 Vdc, and the bus voltage in a wide range is used for widening the adaptability and the application field of the frequency converter to motors with various specifications. The invention belongs to a core functional component in the industrial field, which is completely compatible with the prior industrial personal computers, PLC, nonstandard systems and other upper computers, the functions of interfaces with the upper computers in the system of figure 1 are shown in a table 1, the RS232 interfaces in the table are isolated from strong electricity, other interfaces are commonly grounded with the upper computers, and are completely isolated from a drive control unit circuit, and the isolation voltage level is 2.1kV. The programmable communication interface can be set by a touch screen or a panel operator, can be configured into 3 RS485, one RS485 and one RS422 or 3 LVDS inputs/outputs, the communication protocol can be adjusted according to an upper computer, an integration scheme with an upper computer system is shown in fig. 3, an analog quantity is given as a speed instruction input in fig. 2 (a), a digital quantity is given as a speed input in fig. 2 (b), a motor is controlled in a bus mode in fig. 2 (c), and the motor servo control scheme is given in fig. 2 (d), so that the high-frequency converter is completely compatible with the current upper computer control through the interface. UART1 interfaces RS232 protocol and PC, UART0 uses Modbus-RTU protocol and touch screen.

Table 1: control interface and function for driver products

The data acquisition unit, the IO unit and the programmable communication interface unit of the high-frequency converter of the figure 1 are all designed with isolation interface circuits with the withstand voltage value of more than 2.1kV, and each functional module has high response speed, high reliability and strong anti-interference capability; the high-frequency inverter is used for synchronously and high-speed running of the data acquisition unit, the high-load operation unit and the high-frequency inverter.

The invention is oriented to the field of high-speed and high-performance driving control of permanent magnet motors, adopts a single-chip FPGA as a master controller, invents a high-performance processor special for a high-frequency converter based on a Soc soft core, the Soc system architecture of the high-frequency converter is shown in figure 3, and the sources of IP in the Soc system are divided into two types:

the free IP core provided by the third party comprises a soft core processor in the FPGA block in fig. 3, a phase-locked loop PLL on the left side of the main control board, an SPI interface, an SDRAM interface, a digital input/output PIO and an asynchronous serial port UARTx (x=0-4) IP core, so that the FPGA minimum system and the universal periphery are formed, and the module can be replaced by a GPU (comprising a DSP/ARM/single chip microcomputer) under the condition of low cost.

Independently developing IP cores, wherein the shaded part on the right side in the FPGA block diagram in fig. 3 comprises Modbus-RTU protocol IP cores for high-speed bus control and serial IP cores for permanent magnet motor non-inductive vector control (SVC), wherein one of small batch and high-frequency calculation functional units in the SVC algorithm flow is converted into a special Soc peripheral, the Soc system is only responsible for the initialization of the peripheral, the process data reading, the FSM control of a motor running state machine, and the processes of starting, running, stopping, parameter self-adaption and the like of the motor running are independently completed by the peripheral, so that the bus is not occupied and the SVC algorithm is not interrupted.

As shown in fig. 3, the FOC control dedicated peripheral in the Soc system includes a finite state machine FSM, an ADC synchronous acquisition ADC interface, clarke transformation, park transformation, a back emf observation period ABobsrv, a speed filter GetVel, a speed loop controller vPi, a current loop controller dPi/qPi, a Park inverse transformation invPark, clarke inverse transformation invClarke, a central symmetry vector PWM modulator SVM, and the like.

The control of the controller to the motor is realized by reading and writing a finite state machine FSM, the FSM soft core is a controlled finite state machine, and in a control period, the FSM core firstly controls the off-chip ADC to collect feedback data required by the SVC algorithm at a high speed according to the time point shown in fig. 4, and then controls the peripheral of the SVC algorithm in the series on the right side of fig. 3 to be from top to bottomSVC algorithm is executed in the next sequential chip, and finally the SVM core controls FGPA pin U _T/B 、V _T/B 、W _T/B The off-chip high frequency inverter bridge is driven according to the timing sequence of fig. 4, and the steps are repeatedly executed in the next cycle.

The first step: analog quantity synchronous detection and coordinate transformation

In order to prevent interference of power devices, the ADC interface soft core controls peripheral multi-channel synchronous ADC chip, and the sampling time is precisely determined to be at the midpoint of PWM, as shown in FIG. 4, the acquired analog quantity comprises: three-phase current i _a/b/c Bus voltage V _DC Temperature Temp of motor _MOTOR Temperature Temp of frequency converter base _VFD Inputting 0-10V/4-20mA into the upper computer analog quantity, and after sampling, executing the equivalent Clarke conversion of the amplitude to obtain i _α 、i _β Voltage u _α 、u _β

The Park soft core is based on the rotor position theta of the last control period _r Rotating the coordinate axis to obtain an alternating current component i and a direct current component i _q 、i _d 。

And a second step of: rotor position acquisition technique

The direct detection mode comprises the following steps: as shown in fig. 1, the rotor position is directly acquired by a mechanical position sensor PG arranged on the motor, and the PG input signal is connected with a programmable communication interface of a general main control board.

Indirect detection mode: the rotor position is calculated through an input voltage and feedback current design observer of the motor, the permanent magnet motor is regarded as a sensor, the input voltage of the motor is regarded as excitation of the sensor, the feedback current of the motor is regarded as sensor response, and the rotor position is accurately calculated on line through an algorithm so as to realize magnetic field orientation control, and under the condition that a mechanical position sensor is not used, the speed high-precision control of the permanent magnet motor is realized. The implementation method of the technical scheme is as follows:

the stator single-phase winding circuit is shown in fig. 6, fig. 7 is a flow chart of stator single-phase winding current signals, the L, R circuit G(s) has low-pass characteristic to current, the discretization step in fig. 2 can be advanced, and a digital model is built according to detected parametersIn parallel with the ideal model G (z), a back electromotive force detection flowchart shown in fig. 8 is obtained.

The upper leg measured phase current I (z) in FIG. 8 can be expressed as

I(z)＝(U(z)-E(z))G(z) (3)

The lower leg in fig. 8 estimates currentCan be expressed as

The output of the viewer in FIG. 8

In FIG. 7, parameters of the single-phase winding of the motor can be precisely determined, and the bandwidth of the current sampling frequency is far higher than G(s) in the high-frequency converter, assuming the single-phase winding digital model of FIG. 9Is a precise approximation of the actual model G (z) of the motor in FIG. 7, i.e.

Bringing the formula (6) into the formula (5)

Writing the expression (7) into the transmission form shown in the expression (8)

(8) System description observer equivalent model referring to fig. 9, it can be seen that the back emf is independent of the excitation voltage, and D (z) is set according to the G (z) model parameters and the high frequency converter output frequency such that |d (z) G (z) |>>1, thenReverse electric detection is realized. Note that the system of fig. 9 shown in equation (8) is not directly implemented, and the ABobsv soft core performs the algorithm shown in fig. 8, and as can be seen by analysis of equation (3-8), the output result of the soft core is equivalent to performing the system of fig. 9.

During the operation of the motor, the observation shown in FIG. 8 is realized under an alpha-beta coordinate system, and the counter electromotive force e during the operation of the motor can be obtained _α 、e _β

The Newton dichotomy is adopted, and the position of the rotor is calculated through coordinate axis rotation operation by the getVel soft core

And then theta is set _r Input (11) filter, delta is filter coefficient, and 0<δ<1

And finally, calculating the output of the filter (11) again, and inputting the output to the vPi soft core as the feedback speed of the motor.

And a third step of: speed, flux linkage and torque control

The speed, flux linkage and torque control all adopt anti-integral saturation PI control IP cores, the control algorithm is shown in figure 5, the IP soft cores are exemplified as three peripheral devices vPi, dPi, qPi in the SVC control algorithm, and the three peripheral devices correspond to the speed loop, the d-axis current loop and the q-axis current loop controllers respectively.

In the vPi speed loop, ref is the speed command, fb is the feedback speed, and Out is the q-axis command current i _q ；i _q Is dependent on the rated current of the motor;

dPi the current loop controller Ref is 0 and fb is the feedback current i in equation (3) _d Output is V _d ；

qPi the current loop Ref is the speed loop output i _q Fb is the feedback current i in formula (3) _q Output is V _q ；

The amplitude of the two current loop controller outputs Vd, vq must satisfy:

fourth step: voltage vector space synthesis outputs v from dPi and qPi controllers _d 、v _q Transformation from rotor to stator co-ordinate system, θ _r The result of the calculation from the second step (10)

According to the principle of power equivalence, the 2 phases are restored to 3-phase voltage

According to the principle of voltage vector synthesis, the three-phase voltages can be expressed as

In FIG. 3, the inClarke cell eliminates v according to equation (11-12) _a 、v _b 、v _c Calculating the PWM instruction pulse width T of u, v and w phases _U 、T _V 、T _W The SVM function unit is based on pulse width instruction T _U 、T _V 、T _W According to the time sequence shown in FIG. 4, the FPGA pin U is controlled _T/B 、V _T/B 、W _T/B And outputting a central symmetry vector PWM, and driving the high-frequency inversion unit after isolation.

3. Isolation, heat dissipation, shielding and forming integrated design scheme for high-frequency converter

The industrialization scheme of the high-frequency converter product needs to solve the following problems:

(1) The heat dissipation problem, the switching loss and the on-state loss of the power device of the high-frequency converter;

(2) Shielding problem, high frequency and heavy current electromagnetic interference output by the frequency converter; the plane and the space between the strong current and the weak current are insulated;

(3) Standardization, small batch and multiple varieties of clients in the industrialization process;

considering the problems comprehensively, the invention provides an integrated design scheme for designing isolation, heat dissipation, shielding and molding, which comprises the following implementation method:

the design scheme of the high-frequency converter is used for carrying out function segmentation on a frequency converter system according to the factors of strong and weak current separation, heat dissipation, interference resistance and the like, and the frequency converter system is divided into a general main control board and a driving board, wherein the general main control board has the main functions of: and receiving instructions of a panel operator, a touch screen, a PC, a job site or PLC and other upper computers, feeding back data, controlling a motor through a drive board, collecting motor operation process data in real time, and executing an SVC algorithm. As shown in FIG. 5, the general master control board is composed of a high-performance MCU, a multi-channel ADC, an isolated digital quantity IO, an isolated analog quantity input, a programmable communication interface, an isolated DC/DC and a UART interface. The main functions of the driving plate are as follows: and controlling the motor, and outputting three-phase current, bus voltage, power semiconductor device temperature and motor temperature signals to the main control board. As shown in fig. 6, the driving board is composed of a surge voltage/current protection circuit, an isolated high-frequency three-phase inverter unit, an isolated bus voltage detection circuit, an isolated frequency converter/motor temperature detection circuit, an isolated three-phase current detection circuit, and an isolated switching power supply.

The general main control board of the high-frequency converter and the driving board are arranged on two sides of the base in rows, the general main control board realizes the control analog quantity data of the reading driving board and the control of the high-frequency inversion unit through the control wiring harness, and the structure has the following advantages that

Heat dissipation technical scheme

(1) The heat source power semiconductor on the driving plate is adhered to the high heat conduction and high magnetic conduction base through a heat conduction material, the thickness W of the heat dissipation base and the power of the heat dissipation fan are selected according to the power of the driving plate, and the heat dissipation base and the heat dissipation fan are cooled through the heat dissipation fins; (2) The base is fixed on a metal main board in the user case, and the heat of the base is transferred to the metal case; (3) The PCB of the driving plate adopts a thick copper plate, copper is covered on a large area at the installation position of the power semiconductor device, and a window through hole is additionally formed, so that the self heat dissipation capacity of the driving plate is improved.

Shielding technical scheme

(1) The high-frequency converter system is divided into a general main control board and a driving board, so that strong and weak electric isolation is realized; (2) The high-permeability grounding base separates the driving board from the general main control board, so that electromagnetic interference to the main control board and controlling the linear speed is generated by high voltage and high current of the driving board; (3) The universal main control board and the driving board PCB are divided into different ground planes by digital ground, analog ground, power ground and machine shell, so that the problem of interference between signals is effectively solved; (4) the drive plate is grounded to the chassis via the mounting base.

Product standardization technical scheme

The power range of the high-frequency converter driving plate is between 0.2 and 30kW, and the product production adopts the following standardized technical scheme:

(1) Installation specification standardization

The series frequency converter adopts a unified universal control board; the series of frequency converter driving boards adopt PCBs with uniform specification, the difference among the driving boards is represented by the difference of three components of a power semiconductor, a current sensor and an electrolytic capacitor, but the mounting positions of the three components on the PCBs and the element packaging are completely consistent within the power range; in the integrated molding technology, as shown in fig. 7, the length and width of the base are completely consistent in the power range, only the thickness W of the base is changed according to the heat dissipation requirement of the high-frequency converter, the power W also increases in response to the larger power W, and the base is the mounting carrier of the universal control board, the driving board and the housing.

(2) Flexible design of system software

According to the power level and the rotating speed range of the motor, a corresponding driving plate is selected, the driving plate and the universal plate are installed in the mode of fig. 7, the universal plate is set by operating a touch screen, a panel operator, a PLC, an industrial personal computer or a personal computer, a high-frequency converter is matched with driving product families of different motors and loads, and the main settable parameters are as follows

Table 1: hardware configuration of driver products

According to the design method, products in a full power range can be realized by using 2 PCBs, standardized production is realized, the product applicability is improved, meanwhile, the cost of the products is reduced, a software system with standard and flexibility is uniformly installed, the products can be ensured to be rapidly adapted to the dynamic change of the motor market, different products with low cost and high performance can be produced in a very short development period, the problems of small batch and multiple varieties of customers in the industrialization process are solved, and the survivability and the competitiveness of the products are improved.

To verify the experimental effect of the present invention, a pair of 280W three-phase high-speed DC stages were driven by the high-frequency converter of the present inventionBrushless motor, motor shaft rigid coupling impeller, algorithm for examining operation of fan under high-speed and heavy-load condition and speed accuracy, vibration and efficiency, PWM carrier frequency is set as T _p Time of dead time t=60 kHz _db =300 ns, busbar voltage V _dc The motor phase inductance 23uH with the phase resistance of 0.174 Ω is 32V, the SVC is adopted, the control parameters are shown in fig. 13, the motor rotation speed is 60000rpm, three-channel data are recorded at high speed by adopting the RAM in the FPGA chip, the experimental sampling frequency shown in fig. 20 is 3750Hz, and the other experimental sampling frequencies are 6000Hz.

The precise detection of current and voltage is the key of precisely identifying the rotor position by the noninductive vector control, the current sensor is a precise linear hall sensor, the noise range is within +/-0.17A under the bandwidth of 120kHz at normal temperature, the three-phase current waveform of the motor of the direct-current brushless motor under 60000rpm and the heavy-load running condition is shown in figure 12, the current curve is smooth, the isolation and shielding technical measures effectively eliminate electromagnetic interference, the switching noise shown in the ADC interface soft-core control ADC figure 4 is sampled at a small moment, dv/dt and di/dt noise are restrained, the detection signal noise source is mainly the linear hall itself, the signal waveform approximates to sine wave, and the problems of large torque pulsation, large harmonic current and limited power when the traditional 3-hall and six-step control of the direct-current brushless motor are effectively solved.

The PWM command pulse width T shown in FIG. 13 _U 、T _V 、T _W The soft core of inClarke is calculated according to the output of the current loop dPi and qPi controller (11-12), thus T _U 、T _V 、T _W The fluctuation of the waveform depends on the disturbance of the load, the load of the high-speed fan is constant in the steady-state operation stage, and the curve of the graph of fig. 13 is smooth, so that the low-vibration technical performance of the high-frequency converter under the high-speed operation condition is indirectly shown.

The back electromotive force of the direct current brushless motor is ladder-shaped, the traditional control method is that three-phase windings are conducted two by two, namely 120 DEG power is applied, and a jump magnetic field taking 60 DEG as step length is generated; in the present invention, T is _U 、T _V 、T _W The three-phase voltage waveform of the motor shown in fig. 14 can be obtained by converting the three-phase voltage waveform into voltage according to the method (12), the waveform approximates to sine wave, and the waveform is realThe current 180 degree electrifying mode generates a continuous rotating magnetic field, which ensures the high precision, high efficiency and low vibration of the permanent magnet motor control in principle.

FIG. 15 shows the input and output results of the observer of FIG. 8, with the ABobsv soft cores calculating e in real time driven by sync pulses _α 、e _β The viewer input v is shown _α 、i _α Output e _α In the present invention, the rotor position theta is calculated by detecting the alpha-beta axis back electromotive force component and then by (10) _r Therefore, the present invention only focuses on the back electromotive force e in parameter setting _α 、e _β The relative proportions of the fundamental components, irrespective of e _α 、e _β Absolute accuracy of (c), thus e in the figure _α The trapezoidal back emf of the experimental dc brushless motor is not presented, but approximates a sine, i.e., the fundamental component of the trapezoidal back emf.

The getVel soft core then pair e _α 、e _β Performing adaptive filtering to obtain e _αf 、e _βf And calculate θ _r The three synchronous waveforms are shown in figure 16, θ _r Good linearity and small waveform distortion.

In the field oriented control, the current vector leads the rotor position by 90 ° to achieve maximum torque, and as can be seen from equation (9), the back emf phase also leads the rotor position value by 90 °, i.e., both are ideally co-directional. The current vector phase is precisely measurable, and the alpha-axis current i is given from top to bottom in fig. 17 _α Back electromotive force e _α And a torque angle (phase difference between two waveforms), e _α The value calculation principle is shown in fig. 8, and the alpha axis component output by the ABobsv soft core in the Soc system can be seen in the figure. The back electromotive force vector phase approximates to the current vector phase, the torque angle fluctuates within a range of +/-2 degrees, the control effect is close to PG control, and experimental results show that: under the condition of no position sensor, the rotor position acquisition technical scheme has higher detection precision.

The speed precision experiments are divided into two groups, the sampling frequency is 3750Hz, 2000 points are recorded, the experiment result in the acceleration stage is shown in figure 18, and the instruction rotating speed and the feedback speed are respectively from top to bottomDegree and q-axis command current i _qr (Ref as a qPi soft core), it can be seen that the feedback speed can quickly and accurately follow the commanded speed, and the commanded current varies with the following error. The analysis of the steady-state operation experiment result of the motor 60000rpm is shown in fig. 19, and the speed error calculation mode is as follows

Fig. 19 shows a speed error curve, and the speed accuracy of the high-frequency converter can reach + -2 per mill when the high-frequency converter controls the load of the experimental fan. Experimental results show that the high-frequency converter keeps higher speed control precision and efficiency in the whole speed regulation range.

In order to record the acceleration and deceleration process of the motor in the full speed range from the static rotation speed to 60000rpm, the Soc system transmits q-axis current data to the touch screen every 0.1s through a UART0 serial port, and the waveform (converted into torque on the touch screen) of the q-axis current data is displayed on the touch screen in real time. The experimental steps are as follows: the method is characterized in that the motor is accelerated to 60000rpm, then is decelerated and stopped, and then is started to 60000rpm, the method completely records the acceleration and deceleration waveform of the motor, the recording result is shown in figure 20, a left descending curve is a deceleration process, a right ascending curve is an acceleration process, an open loop control is adopted below an open/closed loop critical frequency in a starting stage, q-axis current cannot react torque at the stage, after the critical frequency is exceeded, the rotor position is locked, and then the control mode is switched from the open loop to a magnetic field directional control mode in the deceleration stage, the q-axis current curve is in a quadratic function shape, because the experimental motor is a fan load, the torque is proportional to the square of the speed, the torque is proved to track the load in the full speed range of the torque of the high-frequency converter, the high efficiency and the accuracy of a control algorithm are reflected, the torque curve is smooth, the torque pulsation is small, and the low vibration control in the full speed range is realized.

A permanent magnet motor integrated driving and detecting system comprises a driver (i.e. a frequency converter) and a data detecting module; the driver is used for driving the permanent magnet motor to rotate; the data detection module is a synchronous data detection module of an IP soft core based on the FPGA; the data collected by the data detection module comprises:

(1) Ch0 speed set point w _r The method comprises the steps of carrying out a first treatment on the surface of the (2) Ch1 speed feedback value w _f The method comprises the steps of carrying out a first treatment on the surface of the (3) CH2 q-axis command current I _q The method comprises the steps of carrying out a first treatment on the surface of the (4) CH3 q-axis feedback current I _qf The method comprises the steps of carrying out a first treatment on the surface of the (5) CH4 q axis voltage command V _q The method comprises the steps of carrying out a first treatment on the surface of the (6) CH5 d-axis command current i _d The method comprises the steps of carrying out a first treatment on the surface of the (7) Feedback current i of CH6 d axis _df The method comprises the steps of carrying out a first treatment on the surface of the (8) Ch7 d axis voltage command V _d ；(9)CH8 V _α (10)CH9 V _β The method comprises the steps of carrying out a first treatment on the surface of the (11) cha phase SVPWM pulse width PWMA; (12) CHB phase SVPWM pulse width PWMB; (13) CHC phase SVPWM pulse width PWMC; (14) CHD i _a ；(15)CHE i _β ；(16)CHF i _b ；(17)CHG i _c The method comprises the steps of carrying out a first treatment on the surface of the (18) CHH phase voltage amplitude U _m The method comprises the steps of carrying out a first treatment on the surface of the (19) CHI phase voltage phase(20) CHJ phase current amplitude I _m The method comprises the steps of carrying out a first treatment on the surface of the (21) CHK phase current phase->(22) CHL counter electromotive force amplitude E _m The method comprises the steps of carrying out a first treatment on the surface of the (23) CHM back emf phase->(24) CHN torque angle, i.e.)>(25) CHO power angle, i.e.)>(26) CHP stall coefficient; (27) CHKa phase back emf waveform E _α The method comprises the steps of carrying out a first treatment on the surface of the (28) CHR b-phase back EMF waveform E _β The method comprises the steps of carrying out a first treatment on the surface of the (29) CHS a current estimate i _α ^* The method comprises the steps of carrying out a first treatment on the surface of the (30) CHT rotor position θ (encoder measurement when sensor is present, constant 0 when no sensor is present); (31) Actual measured speed w of CHU rotor ₀ (encoder measurement when there is a sensor, constant 0 when there is no sensor); (32) CHV encoder line number Rev (encoder measurement with sensor, constant 65535 without sensor)

In the channel, PWMA/B/C is given value, ia/B/C is measured value, and w and θ are calculated values in a non-inductive vector control algorithm; when the PG control algorithm exists, the measurement value is obtained, and all other parameters are calculated values.

Data that must be collected or calculated: the amount that the noninductive vector control must acquire is ia/b/c; rotor position θ must be acquired on this basis with PG (rotor position detector) control.

Calculating motor operation dynamic parameters based on the collected data; the motor operation dynamic parameters include: load torque, winding equivalent resistance/inductance, motor active/reactive power, driver active/reactive power, torque parameters, motor speed, driver temperature, motor temperature, back emf, power angle, load angle.

The data output by the data detection module is stored in a local memory or output to a touch screen for display or output to an upper computer.

The motor static parameter identification module is also included;

the working process of the motor static parameter identification module is as follows: the tester sends out a static parameter identification command (through an upper computer or a touch screen or a keyboard), the controller injects three-phase rotating high-frequency voltage under the condition of motor stalling according to the static parameter identification command, after the current data is stable, excitation voltage and feedback current data are stored in an SRAM (static random access memory) in the controller, then the stored synchronous data are read into an SOC (system on chip) system, and the SOC system calculates the AC and DC shaft inductances, winding resistance, salient pole coefficients and the initial position of a rotor of the motor offline according to a permanent magnet motor data model under a permanent magnet motor alpha-beta coordinate system.

The calculation process of the static parameters of the motor is as follows:

the mathematical model of the permanent magnet motor under the alpha-beta coordinate system is shown as the formula (1)

In which L ₁ ,L ₂ Respectively is an AC/DC axis inductance L _d ,L _q The sum and difference averages, the relationship may be expressed as:

measurement method, under the condition of motor locked-rotor state, injection amplitude is U _i Angular velocity of omega _i High frequency voltage, the excitation voltage can be expressed as

/>

Angular frequency omega of excitation signal in the above _i The inductance of the winding is far greater than the resistance R of the winding, and (1) the first term is negligible; in the measuring process, the motor is blocked, and the last term of (1) is 0; neglecting higher harmonic components; the response current of the motor can be expressed as, by approximation

Vector is combined withDot product vector->Is available in the form of

Vector is combined withCross vector +.>Is available in the form of

(6) Wherein the first term is constant and the second term has a frequency of 2Ω _i Note that (4) only considers fundamental wave and not harmonic wave, and according to the principle of current transformation technology, the inverter circuit inevitably has integral harmonic kΩ of output frequency _i In order to improve the detection precision, an M-order digital trap is designed, and the sampling frequency of a system is set to be f _pwm The excitation frequency of the variable frequency output is f _i Then M can be selected as

M＝l×f _pwm /f _i (l＝1,2,3,…) (7)

Setting the excitation signal frequency f _i Can be divided by f _pwm Ensuring that (7) M is an integer, the transfer function H (z) of the digital filter is shown as (8), the amplitude response of the M-order trap is shown as (7), and the frequency kΩ is shown in the figure _i (k=1, 2, …, 9) is mapped to the zero point of a digital filter, i.e. the z-plane unit circle, which filter can completely eliminate frequencies of kΩ _i (k=1, 2, …) fundamental waves, harmonics of the respective orders.

The filter has all-pass characteristics for baseband signals, has trap characteristics for baseband and subharmonic signals of the output frequency of the frequency converter,for a given value->In the process of high-frequency injection, clicking a 'Sample' button in fig. 3 to store all variables in fig. 3 into an SRAM according to a time sequence shown as 5, reading SRAM voltage and current data by a Soc soft core processor, calculating left data operation results according to a formula (5-6), and obtaining a formula (6) right direct current component value by passing the data through a filter in fig. 8, and then synthesizing (5-6) to calculate L ₁ 、L ₂ Further obtain the equivalent resistance R and inductance L of the motor winding _d/q Coefficient of salient pole L ₂ /L ₁ 。

The permanent magnet motor integrated driving and detecting method is characterized in that the permanent magnet motor integrated driving and detecting system is adopted;

driving the permanent magnet motor through a driver;

and acquiring and calculating actual data by adopting a data detection module.

The high-speed and synchronous data acquisition IP core is shown as a figure, the unit is a soft core functional module generated by hardware description language instantiation, the right side is an IP core input, clock and reset_n are clock and reset inputs of an acquisition system, w_r high/low level represents read/write SRAM, rdAddr is a write address pointer counter, synClk is a write synchronous signal, and the signal is determined by human-computer interface setting and can be synchronous with a position ring, a speed ring or an acceleration ring; and sequentially collecting data of all 32 channels in each position loop, speed loop or acceleration loop period. The left side is IP core output, the smpF lag is the flag bit of the SRAM data, the address is connected with the peripheral SRAM input address, the SRAM data is connected with the peripheral SRAM data port, the port is bidirectional IO, the dataout is the SRAM output corresponding to the address input rdAddress, and the CE_n, OE_n, UB_n, LB_n and WE_n are memory read-write control signals

When the Soc system receives the sampling instruction sent by the upper computer, the SRAM write Enable signal Enable is set, and after the ADC conversion is completed, the high-speed acquisition unit generates CE_n, OE_n, UB_n, LB_n and WE_n command waveforms according to the time sequence, CHx (x=0, 1,2 and …) is the channel address, and the data selected by the channel address is recorded in the memoryThe falling edge is written into SRAM by CHx partition Page address, +.>Rising edge, CHx value is increased by 1, execution 2 ^K After the number of times, all variables are written into the corresponding partition, page address Page is added with 1, the next sampling is waited to finish repeating the time sequence until SRAM is full, then the smpF flag bit is set, one time of sampling command, and the data quantity of continuous sampling is that32 channels x 2 ^K Word/channel x 16 bit/word 2 ^K+9 bit. K=13, i.e. one acquisition command, the ip data acquisition process will acquire 4Mbit of data.

The above embodiments are merely for illustrating the computing concept and features of the present invention, and are intended to enable those skilled in the art to understand the present invention and implement it accordingly, and the scope of the present invention is not limited to the above embodiments. Therefore, all equivalent changes or modifications according to the principles and design ideas of the present invention are within the scope of the present invention.

Claims

1. The high-frequency converter is characterized by comprising a controller, a signal acquisition circuit and a high-frequency inverter bridge; the signal acquisition circuit is connected with the controller; the controller is connected with the high-frequency inverter bridge through an isolation circuit;

the controller is provided with a core module; the core module comprises an MCU minimum system and a peripheral circuit;

the controller is also provided with a soft core control module; the soft core control module comprises a finite state machine FSM, an ADC synchronous acquisition unit, a Clarke transformation unit, a Park transformation unit, a back electromotive force observer, a speed filter, a speed loop controller, a current loop controller, a Park inverse transformation unit, a Clarke inverse transformation and a central symmetry vector PWM modulator;

the controller sends out driving pulses through the centrosymmetric vector PWM modulator to drive the inverter to work;

finite state machine FSM: in a control period, the FSM core firstly controls the ADC synchronous acquisition unit to acquire feedback data required by the SVC algorithm, then controls the SVC algorithm peripheral to execute the SVC algorithm in a chip according to the sequence from top to bottom, and finally controls the driving of the off-chip high-frequency inverter bridge by the output control pulse of the SVM core, and repeatedly executes the steps in the next period;

the analog quantity that ADC synchronous acquisition unit gathered includes: three-phase current i _A 、i _B And i _C DC bus voltage V _DC Temperature Temp of motor _MOTOR Temperature Temp of frequency converter base _VFD And the analog quantity input of the upper computer is 0-10V or 4-20mA;

the Clarke transformation unit is used for executing the equivalent Clarke transformation of the amplitude to obtain i _α 、i _β ；

Wherein:

i _α 、i _β two-phase current of an alpha-beta coordinate system of the motor;

the Park conversion unit is used for converting the rotor position theta of the previous control period _r Performing coordinate axis rotation operation to obtain an alternating-axis and direct-axis current component i _q 、i _d ；

The back electromotive force observer is used for obtaining back electromotive force;

the transfer function of the back emf observer is:wherein->Is the output of the back emf observer; g (z) is a digital model of a motor single-phase winding, the input quantity of the digital model is the difference between input phase voltage and motor back electromotive force, and the output quantity is phase current: />

L, R is inductance, resistance, T of motor stator phase winding _p For the SVPWM period, D (z) is the observer controller:

K _p ＝2ξω ₀ L-R

K _i ＝ω ₀ LT _s

ξ、ω ₀ damping ratio and undamped oscillation frequency of the back electromotive force observer are respectively;

e (z) is the z-transform of the phase back EMF; k (K) _p ，K _i Is a proportional coefficient and an integral coefficient;

the speed filter is used for obtaining the speed dθ (n) at the nth sampling point moment, and the calculation formula is as follows:

dθ (n) =θ (n) - θ (n-1), θ (n) and θ (n-1) are rotor positions at times n and n-1, respectively;

a speed loop controller, wherein in the speed loop of the speed loop controller, a given value Ref is a speed command, a feedback value Fb is a feedback speed, and an output Out is a q-axis command current i _q ^* ；i _q ^* Is dependent on the rated current of the motor;

the current loop controller comprises a dPi current loop controller and a qPi current loop controller;

dPi current loop controller, given value Ref of 0, feedback value Fb of feedback current i _d ' output is V _d ；

qPi current loop controller, given value Ref is speed loop output i _q ^* The feedback value Fb is the feedback current i _q ' output is V _q ；

V _d 、V _q Two-phase voltages of a d-q coordinate system;

V _d 、V _q the amplitude of (2) must satisfy:

the Clarke inverse transformation is used for Clarke inverse transformation, and the alpha-beta axis voltage vector is equivalently converted into an a-b-c three-phase coordinate system;

v _a 、v _b 、v _c a, b and c are respectively three-phase voltages;

V _DC is the voltage of the direct current bus, and the voltage of the direct current bus is the voltage of the direct current bus,tp is the SVPWM period.

2. The high frequency converter of claim 1, further comprising a Modbus-RTU protocol IP core for high speed bus control.

3. The high frequency inverter of claim 1, further comprising an IP core of the permanent magnet motor SVC for determining permanent magnet motor rotor position without a position sensor.

4. A high frequency converter according to any of claims 1-3, wherein the controller communicates with the host computer via a serial port.

5. The high-frequency converter according to claim 4, wherein the control method of the high-frequency converter is a noninductive vector control method;

soc based on FPGA is designed to be used as a main control of the high-frequency converter, and the FOC algorithm and the SVC algorithm of the motor control algorithm are realized by adopting IP soft cores.