CN113595415A - AC/DC resonant converter - Google Patents

AC/DC resonant converter Download PDF

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Publication number
CN113595415A
CN113595415A CN202110662362.7A CN202110662362A CN113595415A CN 113595415 A CN113595415 A CN 113595415A CN 202110662362 A CN202110662362 A CN 202110662362A CN 113595415 A CN113595415 A CN 113595415A
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China
Prior art keywords
output
series
voltage
resonant
diode
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CN202110662362.7A
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Chinese (zh)
Inventor
袁源兰
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Shenzhen Songsheng Innovation Technology Co ltd
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Individual
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4241—Arrangements for improving power factor of AC input using a resonant converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33592—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)
  • Rectifiers (AREA)

Abstract

The invention provides an AC/DC resonance converter, a power conversion circuit is composed of basic switch units and a resonance tank, a controller adaptively controls a power conversion circuit according to different AC input voltages or different DC output voltages, the power conversion circuit is switched into one or more basic switch units to be stacked in series and parallel to work, the power conversion circuit works in a high-frequency full-voltage control mode when AC input low voltage or DC output high voltage is input, and works in a low-frequency voltage division control mode when AC input high voltage or DC output low voltage is output, the two level control modes share the same resonance device to simplify the design of the resonance tank, a multiplier function unit is arranged in the controller, and AC input current tracks the frequency and the phase of the input voltage in real time to realize the power factor correction function. The resonant converter directly converts alternating current input instantaneous voltage into direct current output voltage, so that a conventional PFC converter and a direct current bus capacitor thereof can be saved, the conversion efficiency is improved, and the cost of components is reduced.

Description

AC/DC resonant converter
Technical Field
The invention relates to the technical field of switching power supplies, in particular to an AC/DC resonant converter.
Background
The switching power supply realizes alternating current-direct current (AC/DC) power conversion by utilizing high-frequency work of a power switching tube, has the characteristics of small volume, light weight, high efficiency and the like, is widely applied to the field of industrial and civil electronic equipment, such as chargers, power adapters, LED drivers, industrial and control power supplies and the like, and has the function of Power Factor Correction (PFC) when the switching power supply with a certain output power is regulated by related standards. The single-stage conversion structure in the AC/DC power converter is simple and has lower cost, but the requirements of high Power Factor (PF) and low ripple cannot be met, and meanwhile, the power tube has the risk of overhigh voltage or current stress, so the two-stage conversion structure is widely used in the industry: the front-stage PFC converter is used for adjusting a power factor and realizing input and output energy balance; and the rear-stage DC/DC converter is used for adjusting the output voltage and reducing the output ripple voltage or current. A typical composition structure of the AC/DC converter is shown in fig. 1, where Cb is a DC bus capacitor to balance the input and output power or energy difference and to be used as a filter; co is an output filter capacitor; RL is the dc output load.
The two-stage AC/DC converter circuit topology is shown in FIG. 2, the AC input part comprises an input EMI filter and a rectifying circuit, wherein the EMI filter comprises a filter inductor Lf and a filter capacitor Cf, the input rectifying circuit diodes D1-D4, and the high-frequency filter capacitor Ci. The pre-stage PFC converter uses a Boost converter (Boost) composed of a Boost inductor Lpfc, a Boost diode Dpfc, a power switching tube Qpfc and a body diode D thereofQpfcAnd (4) forming. The bypass diode Db pre-charges the DC bus capacitor Cb when the cold machine is started, and provides a current path for the Cb to absorb when the input end is struck by lightning so as to avoid the influence of transient large current impact on the normal work of the Boost converter. The rear-stage DC/DC converter uses LLC half-bridge resonant converter, mainly composed of first series resonant inductor Lr, parallel resonant inductor Lp, main transformer T1, secondary side rectifier diodes D5, D6, power switch tubes Q1, Q2 and body diode D thereofQ1、DQ2And (4) forming. The two controllers respectively control the boost PFC and LLC half-bridge resonant converter to work, and generally adopt PWM, frequency conversion or mixed chips.
The two-stage AC/DC converter can achieve high power factor and can reduce output ripples, but the number of components in a power conversion circuit and a controller is large, so that the material cost is high. Meanwhile, the total output can be obtained only after the alternating current input is subjected to two-stage power conversion, more power loss is inevitably generated, and the conversion efficiency is the product of the conversion efficiency of the PFC and the DC/DC, so that the overall conversion efficiency is lower. When the boost PFC converter works at a lower alternating-current input voltage, the conduction loss of the boost PFC converter is increased due to the increase of the input current, so that the conversion efficiency of a low-voltage section in a wide input voltage range of 85-276 Vac is lower. Although the LLC half-bridge resonant converter has low turn-on loss, when operating in a wide output voltage range, even if the dc bus voltage remains constant, the switching frequency will also vary greatly and deviate from its resonant frequency point, resulting in increased turn-off and turn-on losses, and therefore is not suitable for wide output voltage situations. Therefore, there is a need in the industry for new AC/DC converters that achieve both simpler conversion structures and reduced component costs, and that achieve high power factors while being capable of operating over a wide range of input and output voltages.
Disclosure of Invention
In order to overcome the defects of the prior art, the invention aims to provide the AC/DC resonant converter, which directly converts the alternating current input instantaneous voltage into the direct current output voltage, can save the conventional PFC converter and the direct current bus capacitor thereof, improves the conversion efficiency and reduces the cost of components; the power tube is soft switch work under different operating conditions and can reduce turn-off loss, and low frequency partial pressure control mode has lower equivalent switching frequency simultaneously to greatly reduce its switching loss, further improve conversion efficiency, high integration controller design can further reduce the switching power supply volume in addition.
The invention provides an AC/DC resonant converter, which comprises an output rectifying and filtering circuit, a power conversion circuit and a controller, wherein the power conversion circuit is formed by a plurality of basic switch units and a resonant tank, the power conversion circuit is connected with an input rectifying circuit of a switching power supply, the output rectifying and filtering circuit is connected with the power conversion circuit, the controller is connected with the output rectifying and filtering circuit, the power conversion circuit and the input rectifying circuit, the basic switch units comprise high-frequency filter capacitors, a plurality of power switch tubes and body diodes of the power switch tubes, the power switch tubes are connected in series, the high-frequency filter capacitors are connected in series with the power switch tube series circuit, the basic switch units form a series and parallel stacked mode, and the controller comprises a high-voltage starting and internal power supply unit, an enable and protection unit, The device comprises an output voltage and current sampling and feedback unit, a light load processing mode unit, a voltage-controlled oscillator and pulse modulator, a primary side current sampling and conditioning unit, a multiplier and main control logic unit, a grid signal enabling and driving unit and an alternating current input voltage sampling and processing unit, wherein the voltage-controlled oscillator and pulse modulator are connected with the grid signal enabling and driving unit, the enabling and protecting unit, the multiplier and the main control logic unit, and the primary side current sampling and conditioning unit, the output voltage and current sampling, the light load processing mode unit and feedback unit and the alternating current input voltage sampling and processing unit are connected with the multiplier and the main control logic unit.
Further, the power conversion circuit is formed by two basic switch units which are stacked in series, the resonant tank comprises a first series resonant inductor, a parallel resonant inductor and a series resonant capacitor, the output rectifying circuit adopts a transformer center tap diode full-wave rectifying structure form and comprises a first rectifying diode, a second rectifying diode, a main transformer and an output filter capacitor, the anodes of the first rectifying diode and the second rectifying diode are respectively connected to two ends of a secondary winding of the main transformer, the cathodes of the first rectifying diode and the second rectifying diode are connected with the anode of the output filter capacitor, the cathode of the output filter capacitor is connected to the center tap of the main transformer, and two ends of an output load are respectively connected to the anode and the cathode of the output filter capacitor; the output midpoint of the first basic switch unit is connected to one end of a first series resonance inductor, the output midpoint of the second basic switch unit is connected to one end of a series resonance capacitor, the other ends of the first series resonance inductor and the series resonance capacitor are respectively connected to two ends of a parallel resonance inductor, and two ends of the parallel resonance inductor are connected with two ends of a primary winding of a main transformer in parallel; the controller comprises an output resistor, a voltage error amplifier, a primary and secondary side isolation optocouplers, a current source, a multiplier, an input resistor, a current error amplifier, a voltage-controlled oscillator, a pulse modulator, a grid signal enable and drive unit and a grid amplifier, wherein the output resistor is connected with the anode of the output filter capacitor, the sampled direct current output voltage of the output resistor is connected with the negative end of the voltage error amplifier in parallel, the positive end of the voltage error amplifier is connected with a voltage reference signal, the output end of the voltage error amplifier is connected to the cathode of a light-emitting diode in the primary and secondary side isolation optocouplers, the anode of the light-emitting diode is connected to the anode of the output filter capacitor through a resistor, the emitter of a triode in the primary and secondary side isolation optocouplers is connected with a primary ground, the collector is connected with the current source in the controller, is connected to the grid signal enable and drive unit, and is connected to one end of the multiplier for input, the input resistor is connected with the output end of the input rectifying circuit, the sampled and rectified voltage is connected to the other end of the multiplier to be input, the output end of the multiplier is connected to the positive end of the current error amplifier to serve as a current reference signal, the primary input current sampling signal is connected to the negative end of the current error amplifier, the output end of the current error amplifier is connected to the voltage-controlled oscillator to generate a sawtooth wave oscillation signal, the sawtooth wave oscillation signal enters the pulse modulator, and a pulse driving signal is generated and passes through the grid amplifier to drive the power switch tube of the basic switch unit.
The direct current power supply circuit further comprises a capacitance energy storage branch circuit, wherein the capacitance energy storage branch circuit is connected with the output positive electrode and the output negative electrode of the input rectification circuit in parallel, and the capacitance energy storage branch circuit is formed by connecting a first power switch tube, a body diode of the first power switch tube and a direct current capacitor in series;
the second-stage LC filter circuit is composed of a filter inductor and a second filter capacitor, the second filter capacitor is connected with the output filter capacitor in parallel, and the filter inductor Ld is connected between the anode of the output filter capacitor and the anode of the second filter capacitor;
the cascade step-down DC/DC converter is composed of a second power switch tube, a body diode of the second power switch tube, a main inductor, a freewheeling diode and a third filter capacitor, the freewheeling diode and the output filter capacitor are connected in parallel, the drain electrode of the second power switch tube is connected with the third filter capacitor and the negative electrode of the freewheeling diode, and the main inductor is connected between the negative electrode of the freewheeling diode and the positive electrode of the output filter capacitor.
Further, the output rectification circuit comprises a main converter and an auxiliary converter, and is provided with a corresponding main controller and an auxiliary controller, the main converter is provided with two auxiliary windings which are respectively a main winding and an auxiliary winding, the main winding of the main converter forms a main output after being rectified and filtered by the first rectifying diode, the second rectifying diode and the output filter capacitor, the auxiliary winding of the auxiliary converter forms a direct current bus voltage after being rectified and filtered by the third rectifying diode, the fourth rectifying diode and the third filter capacitor, and forms an auxiliary output after being rectified and filtered by the second power switch tube, the body diode of the second power switch tube, the main inductor, the fly-wheel diode and the third filter capacitor, and the main output and the auxiliary output are connected in series to obtain a total output voltage which is provided for an output load.
Further, the power conversion circuit uses a diode-clamped AC/DC three-level LLC half-bridge resonant converter, and further includes a first diode and a second diode, a series midpoint of the high-frequency filter capacitor of the first basic switching unit and the high-frequency filter capacitor of the second basic switching unit is connected to an anode of the first diode, a cathode of the second diode, and the series resonant capacitor, a cathode of the first diode is connected to a series midpoint of the power switch tube of the first basic switching unit, an anode of the second diode is connected to a series midpoint of the power switch tube of the second basic switching unit, and a series midpoint of the power switch tube of the first basic switching unit and the power switch tube of the second basic switching unit is connected to the first series resonant inductor.
Further, the power conversion circuit uses a diode-clamped AC/DC three-level LLC full-bridge resonant converter, the power conversion circuit further comprises a third diode, a fourth diode, a third basic switch unit and a fourth basic switch unit, the series midpoint of the high-frequency filter capacitor of the first basic switch unit and the high-frequency filter capacitor of the second basic switch unit is connected with the anode of the third diode and the cathode of the fourth diode, the cathode of the third diode is connected with the series midpoint of the power switch tube of the third basic switch unit, the anode of the fourth diode is connected with the series midpoint of the power switch tube of the fourth basic switch unit, the series midpoint of the power switch tube of the first basic switch unit and the power switch tube of the second basic switch unit is connected with the series resonant capacitor, the power switch tube of the third basic switch unit and the power switch tube of the fourth basic switch unit are connected with the first series resonance inductor in series connection with the midpoint.
Furthermore, the number of the AC/DC resonant converters is two, and the inputs of the first set of LLC half-bridge resonant converter and the second set of LLC half-bridge resonant converter are connected in series or adopt a phase-staggered 180-degree control mode;
the number of the AC/DC resonant converters is three, and EMI filter inductors and filter capacitors of the A phase, the B phase and the C phase of three-phase alternating current are respectively connected with the AC/DC resonant converters through an input rectification circuit.
Further, the power conversion circuit is formed by stacking four basic switch units in series, a power switch tube series branch of a third basic switch unit is connected between power switch tubes of the first basic switch unit, a power switch tube series branch of a fourth basic switch unit is connected between power switch tubes of the second basic switch unit, the output midpoint of the third basic switch unit is connected to one end of the first series resonant inductor, and the output midpoint of the fourth basic switch unit is connected to one end of the series resonant capacitor; the other ends of the first series resonance inductor and the series resonance capacitor are respectively connected to two ends of a parallel resonance inductor, and the parallel resonance inductor is connected with two ends of a primary winding of the main transformer in parallel;
the power conversion circuit is formed by stacking a basic switch unit in series, the resonant tank comprises a first series resonant inductor, a parallel resonant inductor and/or a parallel resonant capacitor and a series resonant capacitor, the middle point of the output of the basic switch unit is connected to the series resonant capacitor, the series resonant capacitor is connected with the first series resonant inductor, a power switch tube branch of the basic switch unit is connected with the parallel resonant inductor and/or the parallel resonant capacitor, and the parallel resonant inductor and/or the parallel resonant capacitor are connected with two ends of a primary winding of the main transformer in parallel;
the resonant tank further comprises a second series resonant inductor, and the second series resonant inductor is connected between the first series resonant inductor and the main transformer;
EMI filter inductors and filter capacitors of the three-phase alternating current phase A, the phase B and the phase C are connected with the basic switch unit through an input rectification circuit;
the power conversion circuit is formed by parallelly stacking two basic switch units, two power switch tube series branches are parallelly connected and parallelly connected with a high-frequency filter capacitor, the output midpoint of one power switch tube series branch is connected with a first series resonance inductor, and the output midpoint of the other power switch tube series branch is connected with a series resonance capacitor.
Further, the number of the resonant tanks is two, a first series resonant inductor of a first resonant tank is connected with an output midpoint of a series branch circuit of a power switch tube of a first basic switch unit, a series resonant capacitor of the first resonant tank is connected with the output midpoint of the series branch circuit of the power switch tube of the first basic switch unit, a first series resonant inductor of a second resonant tank is connected with the output midpoint of the series branch circuit of the power switch tube of a second basic switch unit, and a series resonant capacitor of the second resonant tank is connected with the series branch circuit of the power switch tube of the second basic switch unit; the high-frequency filter circuit also comprises a voltage section selection switch which is connected between the input rectifying circuit and the series midpoint of the high-frequency filter capacitor of the first basic switch unit and the high-frequency filter capacitor of the second basic switch unit.
Further, the output rectifying circuit comprises a first rectifying diode, a second rectifying diode, an output filter capacitor, a third rectifying diode and a fourth rectifying diode, wherein the first rectifying diode, the second rectifying diode, the third rectifying diode and the fourth rectifying diode form a rectifying bridge, the input end of the rectifying bridge is connected with the parallel resonant inductor, and the output end of the rectifying bridge is connected with the output filter capacitor;
the anodes of the first rectifier diode and the second rectifier diode are respectively connected to the two ends of the parallel resonant inductor, the cathodes of the first rectifier diode and the second rectifier diode are connected with the anode of the output filter capacitor, the cathode of the output filter capacitor is connected to a center tap of the main transformer, and the two ends of an output load are respectively connected to the anode and the cathode of the output filter capacitor; the output midpoint of the first basic switch unit is connected to one end of a first series resonance inductor, the output midpoint of the second basic switch unit is connected to one end of a series resonance capacitor, the other ends of the first series resonance inductor and the series resonance capacitor are respectively connected to two ends of a parallel resonance inductor, and two ends of the parallel resonance inductor are connected with two ends of a primary winding of a main transformer in parallel;
the output rectifying circuit comprises an autotransformer, a first rectifying diode, a second rectifying diode, an output filter capacitor and a fourth filter capacitor, the first series resonant inductor and the series resonant capacitor are connected with the primary side of the autotransformer, the first rectifying diode and the output filter capacitor are connected to two ends of the secondary side of the autotransformer in parallel, the fourth filter capacitor is connected between the secondary side of the autotransformer and the negative electrode of the first rectifying diode, and the positive electrode of the second rectifying diode is connected between the negative electrode of the first rectifying diode and the positive electrode of the output filter capacitor.
Compared with the prior art, the invention has the beneficial effects that:
the invention provides an AC/DC resonant converter, which is free of a conventional PFC converter and a direct-current bus capacitor thereof, improves the conversion efficiency of the converter and reduces the cost of components; under the working conditions of alternating current input low voltage or direct current output high voltage, and alternating current input high voltage or direct current output low voltage, the power tubes are soft switches to work, so that the turn-off loss can be reduced; the low equivalent switching frequency is achieved when the alternating current is input to high voltage or the direct current is output to low voltage, so that the switching loss of the high equivalent switching frequency is greatly reduced, and the conversion efficiency is further improved; the high integration controller design can further reduce the size of the switching power supply; the method is suitable for single-phase or/and three-phase alternating current input systems.
The foregoing description is only an overview of the technical solutions of the present invention, and in order to make the technical solutions of the present invention more clearly understood and to implement them in accordance with the contents of the description, the following detailed description is given with reference to the preferred embodiments of the present invention and the accompanying drawings. The detailed description of the present invention is given in detail by the following examples and the accompanying drawings.
Drawings
The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this application, illustrate embodiment(s) of the invention and together with the description serve to explain the invention without limiting the invention. In the drawings:
FIG. 1 is a schematic diagram of an exemplary structure of an AC/DC converter according to the background art of the present invention;
FIG. 2 is a schematic diagram of a circuit topology of a two-stage AC/DC converter according to the background art of the present invention;
FIG. 3 is a schematic diagram of an AC/DC resonant converter of the present invention;
FIG. 4 is a schematic diagram of a basic switch cell stack of the present invention;
FIG. 5 is a schematic diagram of the output rectifying filter circuit according to the present invention;
FIG. 6 is a schematic diagram of the main functional units inside the controller according to the present invention;
fig. 7 is a schematic diagram of an AC/DC LLC half-bridge resonant converter with automatic switching of the secondary-side feedback AC input voltage range and its controller according to a first embodiment of the present invention;
FIG. 8 is a schematic diagram of the main operating waveforms of the AC input low-voltage section according to the first embodiment of the present invention;
FIG. 9 is a schematic diagram of an equivalent circuit of the switching mode of the AC input low-voltage section t 0-t 1 according to the first embodiment of the present invention;
FIG. 10 is a schematic diagram of an equivalent circuit of the switching mode of the AC input low-voltage section t 1-t 2 according to the first embodiment of the present invention;
FIG. 11 is a schematic diagram of an equivalent circuit of the switching mode of the AC input low-voltage section t 2-t 3 according to the first embodiment of the present invention;
FIG. 12 is a schematic diagram of an equivalent circuit of the switching mode of the AC input low-voltage section t 3-t 4 according to the first embodiment of the present invention;
fig. 13 is a schematic diagram of the main operating waveforms of the ac input high voltage section according to the first embodiment of the present invention;
FIG. 14 is a schematic diagram of an equivalent circuit of the switching mode of the AC input high-voltage section t 0-t 1 according to the first embodiment of the present invention;
FIG. 15 is a schematic diagram of an equivalent circuit of the switching mode of the AC input high-voltage section t 1-t 2 according to the first embodiment of the present invention;
FIG. 16 is a schematic diagram of an equivalent circuit of the switching mode of the AC input high-voltage section t 2-t 3 according to the first embodiment of the present invention;
FIG. 17 is a schematic diagram of an equivalent circuit of the switching mode of the AC input high-voltage section t 3-t 4 according to the first embodiment of the present invention;
FIG. 18 is a schematic diagram of an equivalent circuit of the switching mode of the AC input high-voltage section t 4-t 5 according to the first embodiment of the present invention;
FIG. 19 is a schematic diagram of an equivalent circuit of the switching mode of the AC input high-voltage section t 5-t 6 according to the first embodiment of the invention;
FIG. 20 is a schematic diagram of an equivalent circuit of the switching mode of the AC input high-voltage section t 6-t 7 according to the first embodiment of the present invention;
FIG. 21 is a schematic diagram of an equivalent circuit of the switching mode of the AC input high-voltage section t 7-t 8 according to the first embodiment of the present invention;
fig. 22 is a schematic diagram of PFC gain requirement, LLC voltage gain, and instantaneous output power according to a first embodiment of the present invention;
FIG. 23 is a diagram of an AC/DC LLC half-bridge resonant converter with an AC input voltage range auto-switching circuit for lightning protection and hold-up time extension according to a second embodiment of the invention;
FIG. 24 is a diagram of an AC/DC LLC half-bridge resonant converter with an output filter and an automatic switching function for the AC input voltage range according to a third embodiment of the invention;
fig. 25 is an AC/DC LLC half-bridge resonant converter with automatic switching of the AC input voltage range of the output cascaded buck DC/DC converter according to the fourth embodiment of the present invention;
fig. 26 is an AC/DC LLC half-bridge resonant converter with automatic switching of the AC input voltage range of the output series step-down DC/DC converter according to the fifth embodiment of the present invention;
fig. 27 is a diode-clamped AC/DC three-level LLC half-bridge resonant converter of a sixth embodiment of the present invention;
fig. 28 is a diode-clamped AC/DC three-level LLC full-bridge resonant converter according to a seventh embodiment of the present invention;
fig. 29 is an AC/DC LLC half-bridge resonant converter with two-phase input connected in series/parallel with an AC input voltage range for automatic switching according to an eighth embodiment of the present invention;
fig. 30 shows an AC/DC LLC half-bridge resonant converter with three-phase AC input voltage range automatic switching in parallel output according to the ninth embodiment of the present invention;
fig. 31 is a diagram of an AC/DC LLC half-bridge resonant converter with automatic switching over a wider AC input voltage range according to a tenth embodiment of the present invention;
FIG. 32 is an AC/DC LLC half-bridge resonant converter of an eleventh embodiment of the invention;
FIG. 33 is an AC/DC LCC half bridge resonant converter of a twelfth embodiment of the present invention;
FIG. 34 shows an AC/DC CLCL half-bridge resonant converter according to a thirteenth embodiment of the invention;
FIG. 35 is an AC/DC LCLC half-bridge resonant converter in accordance with a fourteenth embodiment of the present invention;
FIG. 36 is a schematic diagram of a fifteen embodiment three-phase AC/DC LLC half-bridge resonant converter;
fig. 37 is an AC/DC LLC full-bridge resonant converter according to a sixteenth embodiment of the present invention;
FIG. 38 is a diagram illustrating a switching mode equivalent circuit of a low-voltage section with ac inputs t 0-t 1 according to a sixteenth embodiment of the present invention;
FIG. 39 is a diagram illustrating a switching mode equivalent circuit of a low-voltage section with ac inputs t 1-t 2 according to a sixteenth embodiment of the present invention;
FIG. 40 is a diagram illustrating a switching mode equivalent circuit of a low-voltage section with ac inputs t 2-t 3 according to a sixteenth embodiment of the present invention;
FIG. 41 is a diagram illustrating a switching mode equivalent circuit of a low-voltage section with ac inputs t 3-t 4 according to a sixteenth embodiment of the present invention;
FIG. 42 is a diagram illustrating a switching mode equivalent circuit of a sixteen-way AC input high-voltage section from t0 to t1 according to an embodiment of the present invention;
FIG. 43 is a switch mode equivalent circuit of the high-voltage section with ac input at t 1-t 2 according to a sixteenth embodiment of the present invention;
FIG. 44 is a diagram illustrating a switching mode equivalent circuit of a sixteen-way AC input high-voltage section from t2 to t3 according to an embodiment of the present invention;
FIG. 45 is a diagram illustrating a switching mode equivalent circuit of a sixteen-way AC input high-voltage section from t3 to t4 according to an embodiment of the present invention;
fig. 46 is an AC/DC LLC double half-bridge resonant converter with automatic switching of AC input voltage range according to a seventeenth embodiment of the present invention;
fig. 47 shows an AC input voltage range auto-switching non-isolated AC/DC LLC half-bridge resonant converter in accordance with an eighteenth embodiment of the present invention;
fig. 48 shows a nineteenth embodiment of the present invention of a non-isolated high gain AC/DC LLC half-bridge resonant converter with automatic switching of the AC input voltage range.
Detailed Description
The present invention will be further described with reference to the accompanying drawings and the detailed description, and it should be noted that any combination of the embodiments or technical features described below can be used to form a new embodiment without conflict.
The development trend of switching power supply technology is high conversion efficiency, high power density and wide voltage range. The resonant converter has the inherent characteristics of Zero Voltage Switching (ZVS) and Zero Current Switching (ZCS), and can reduce the switching loss to achieve higher conversion efficiency, so that the switching frequency can be higher to reduce the volume of a passive device to improve the power density. Resonant converters generally adopt frequency conversion control, when input/output voltage or load changes, the switching frequency needs to be correspondingly changed to adjust voltage gain, and higher switching frequency causes larger turn-off loss. The wider the range of the ac input voltage or the dc output voltage, the more the switching frequency will change and deviate from its resonant frequency, and the larger the input current will cause larger conduction losses at lower input voltages. The more the switching frequency deviates from the resonant frequency, the greater the reactive circulating current loss becomes, the more the conversion efficiency of the resonant converter is reduced, and the more difficult it is to optimally design the resonant device and the main transformer thereof.
Radio frequency Power Amplifiers (PAs) in communication devices widely use frequency multiplication, which is a DC/AC inverter circuit whose output signal frequency is a harmonic component of the input signal frequency. If the input signal frequency is f0, the output signal frequency component is N x f0 due to the frequency-selective filtering function of the resonant tank, where N is a natural number. Therefore, other useless frequency components can be effectively filtered, and the limitation of parasitic factors can be broken through to further improve the signal frequency. Many of the switching power supply technologies come from communication technologies, such as soft switching technologies from Class D/E (Class-D/E) power amplifiers, Pulse Width Modulation (PWM) technologies from carrier modulation and demodulation, and so on. The frequency multiplication technique can also be extended to switching power supplies or AC/DC converters, but its purpose is not to achieve higher switching frequencies, but to adjust the input and output voltage gains, so that the switching frequency can be varied less to adapt to work with a wider AC input voltage or DC output voltage range, and to achieve Power Factor Correction (PFC) functions. The rated voltage of the power device can be lower, so that higher conversion efficiency can be conveniently achieved, and the conversion efficiency can be maintained to be basically unchanged when the power device works in a wider voltage range, so that the requirements of different voltage grades of a global power grid are met.
The switching power supply is mainly composed of an EMI filter, an input rectification circuit, and an AC/DC resonant converter, as shown in fig. 3. The EMI filter comprises a filter inductor Lf and a filter capacitor Cf, and the input rectifying circuit comprises rectifying diodes D1-D4. The AC/DC resonant converter comprises an output rectifying and filtering circuit, a power conversion circuit consisting of a basic switch unit and a resonant tank and a controller thereof. The power conversion circuit is connected with an input rectification circuit of the switching power supply, the output rectification filter circuit is connected with the power conversion circuit, and the controller is connected with the output rectification filter circuit, the power conversion circuit and the input rectification circuit. Resonant tanks are of various types, such as LLC, LCC, CLCL, LCLC, etc., and the main transformer in an isolated converter can also be considered part of the resonant tank. The AC input voltage Vin is filtered by a filter inductor Lf and a filter capacitor Cf in the EMI filter, rectified voltage | Vin | obtained by D1-D4 rectification is supplied to a power conversion circuit, a controller samples DC output voltage and AC input current Iin, appropriate driving signals are output to a basic switch unit of the power conversion circuit after logic processing, high-frequency switch work is carried out, power transmission in an isolation or non-isolation mode is carried out, and stable DC is finally suppliedVoltage or current to output load RL. The invention mainly relates to a series-parallel stacked structure of basic switch units, which adopts two level control modes of high-frequency full voltage and low-frequency voltage division, and a power conversion circuit directly converts alternating current input instantaneous voltage into direct current output voltage, so that a conventional PFC converter and a direct current bus capacitor thereof can be saved, the conversion efficiency is improved, and the cost of components is reduced. The controller is internally provided with a multiplier function unit, and the input current tracks the frequency and the phase of the alternating current input voltage in real time so as to realize the function of Power Factor Correction (PFC).
A basic switch unit comprises a high-frequency filter capacitor Ci, two power switch tubes Q1, Q2 and a body diode D of two power switch tubes Q1, Q2Q1、DQ2The basic switching cells may be constructed in series and parallel stacks as shown in fig. 4. It should be noted that the number of the basic switch units connected in series or in parallel is not limited, so that a more complicated power conversion circuit can be configured. Different types of fully-controlled power switching devices such as metal oxide field effect transistors (MOSFET), Insulated Gate Bipolar Transistors (IGBT) and power triodes (BJT) can be used in the basic switching unit, and the gate drive circuit or the gate drive circuit of the power switching devices needs to be adjusted correspondingly when the different types of power switching devices are used.
The structural form of the output rectifying and filtering circuit is shown in fig. 5, T1 is a main transformer in isolation mode power transmission, (1) is a transformer center tap diode full-wave rectification, and is generally used in the occasion of outputting low voltage and large current; (2) the diode full-bridge rectifier is generally used for outputting high-voltage and low-current occasions; (3) the diode voltage-multiplying rectification is generally used for outputting high voltage; (4) the full-wave synchronous rectification is carried out on the center tap of the transformer, and is generally used for low-voltage and high-current occasions with higher attention to conversion efficiency. Different types of power switching tubes can be used in the synchronous rectification circuit, such as MOSFET, IGBT and BJT. The embodiment of the invention includes but is not limited to the structural form of the output rectifying and filtering circuit, and mainly takes full-wave rectification of a transformer center tap diode as an example.
The controller includes a plurality of conventional functional units, as shown in fig. 6, and mainly includes a high-voltage starting and internal power supply unit, an enabling and protecting unit, an output voltage and current sampling and feedback unit, a light load processing mode unit, a voltage-controlled oscillator and pulse modulator, a primary side current sampling and conditioning unit, a main control logic unit, a gate driving unit, and the like. After the high-voltage starting is finished, the internal power supply starts to work, so that the two can share one functional unit. The enabling and protecting unit realizes multiple safety protection functions of overhigh input or/and output voltage, undervoltage, overcurrent, overtemperature and the like. The main control logic unit is a core control unit, and can flexibly select different control modes under different working conditions to realize a frequency modulation mode (PFM), a pulse width modulation mode (PWM) or a mixed mode. The output voltage and current sampling and feedback unit is also important, and the accuracy and the error range of the output voltage and current sampling and feedback unit are directly influenced. The light load processing mode unit mainly controls light load working performance, such as working in an intermittent mode, limiting the switching frequency range of the light load processing mode unit, controlling light load working noise, reducing light load power loss and the like. A Voltage Controlled Oscillator (VCO) is a frequency conversion PFM core unit, the oscillation frequency of the VCO is controlled through the amplitude of a voltage error signal, and the resonant working performance is greatly influenced by the frequency precision. After the VCO generates a sawtooth wave signal, the drive signal is obtained after pulse signal processing in the pulse modulator, and is provided for the grid drive unit, and the internal dead zone generating circuit influences not only the soft switching characteristic but also the working reliability of the switching power supply. The current controller chip is controlled in a current mode more, so that a gain curve of the resonant converter is prevented from entering a capacitance area, the working reliability of the switching power supply is further improved, and a primary current sampling and conditioning unit is arranged in the controller. The grid driving unit is mainly used for driving a power switch tube MOSFET and is required to provide instantaneous large-current driving output.
The primary side current sampling and conditioning unit not only samples a primary side current signal, but also is more importantly provided with a current error amplifier, and two input signals of the current error amplifier are respectively from the output of the multiplier and the primary side current sampling signal. The multiplier is a core unit for realizing PFC function, two input signals of the multiplier are respectively from input rectified voltage sampling and an original output signal of a conventional output voltage error amplifier, and the multiplier is provided with a signal filtering circuit. The alternating current input voltage sampling and processing unit samples and detects input rectified voltage, the output voltage sampling unit samples and detects direct current output voltage, and the output voltage sampling and detecting unit respectively compares the direct current output voltage with internal reference signals and then sends out logic instructions to the main control logic unit so as to identify judgment conditions. The pulse modulator adaptively switches the serial-parallel stacked structure of the basic switch unit under the control of the main control logic unit, and the power switch tube has more driving signals and also needs to realize the corresponding logic time sequence requirement. The grid signal selection and enabling unit receives the pulse driving signal from the pulse modulator, and provides instantaneous large-current driving output after internal selection and enabling logic processing. The controller can be built by using discrete electronic components, and can also be designed and used by using special integrated circuits, such as an analog control chip, a singlechip (MCU) programmed by software or a programmable logic device (FPGA/CPLD) and the like. The power conversion circuit and the rectifying unit can adopt a discrete device mode or a respective integrated and mixed integrated mode, and can also be uniformly integrated into the controller to form a large-scale mixed integrated circuit, and the design of the high-integration controller can reduce the size of the switching power supply.
The AC/DC resonant converter uses the frequency multiplication technology, and the output of the high-frequency square wave voltage by the resonant tank is mainly realized through a multi-level conversion circuit. Conventional multi-level inverters generate a plurality of different levels to reduce output Total Harmonic Distortion (THD), whereas AC/DC resonant converters always operate in a two-level mode over a wide AC input voltage range. In an embodiment, as shown in fig. 7, an AC/DC LLC half-bridge resonant converter and its controller are automatically switched for an AC input voltage range, and a Secondary-Side Regulation or feedback mode (SSR) is adopted, and optionally a Primary-Side Regulation or feedback mode (PSR) is adopted. The power conversion circuit is formed by stacking two basic switch units in series, wherein a high-frequency filter capacitor Ci1, power switch tubes Q1, Q2 and a body diode D thereofQ1、DQ2The first basic switch unit is formed, the two tubes are connected in series to form a bridge arm, and the output midpoint of the bridge arm is connected to one end of a series resonance inductor Lr; high-frequency filter capacitor Ci2, power switch tubes Q3, Q4 and body diode D thereofQ3、DQ4Forming a second basic switch unit, connecting two tubes in series to form another bridge arm, and outputtingThe middle point is connected to one end of a series resonance capacitor Cr; the other ends of the Lr and the Cr are respectively connected to two ends of a parallel resonant inductor Lp, and two ends of the Lp are connected with two ends of a primary winding of a main transformer T1 in parallel. The output rectifying circuit adopts a transformer center tap diode full-wave rectification structure, the anodes of rectifying diodes D5 and D6 are respectively connected to two ends of a T1 secondary winding, the cathodes of the rectifying diodes D5 and D6 are connected with the anode of an output filter capacitor Co, the cathode of Co is connected to a T1 center tap, and an output load RLTwo ends are respectively connected to the anode and the cathode of the output filter capacitor Co. The series resonance inductor Lr and the parallel resonance inductor Lp can use independent inductors, and can also respectively use leakage inductance and excitation inductance of the main transformer T1, so that the magnetic integration design is realized. The controller mainly comprises a plurality of internal functional units, and also comprises an output voltage Vo, an output current Io sampling and feedback circuit, an original secondary side isolation optocoupler and other peripheral control circuits. The resistors Ro1 and Ro2 sample the DC output voltage and are connected to the negative terminal of the voltage error amplifier U1, and the positive terminal of U1 is connected with a voltage reference signal Vref. The output end of the U1 Is connected to the cathode of a light emitting diode in the primary and secondary side isolation optocoupler U2, the anode of the light emitting diode Is connected to the anode of Co through a resistor, the emitter of an internal triode in the U1 Is connected to the primary side ground, and the collector of the triode Is connected with a current source Is in the controller, Is connected to other functional units and Is also connected to one end input of a multiplier U3. The resistors Ri1 and Ri2 sample the rectified voltage and then are connected to the other end of the multiplier U3 for input, i.e., the ac input voltage is indirectly detected. The output end of the U3 is connected to the positive end of a current error amplifier U4 as a current reference signal, a primary side input current sampling signal Iin is connected to the negative end of U4, namely, the AC input current is indirectly detected, and a current sampling device can optionally use a current sensor, a current transformer or a resistor, and can also sample the voltage at two ends of Cr. The output end of the U4 is connected to a Voltage Controlled Oscillator (VCO) to generate sawtooth wave oscillation signals, and the sawtooth wave oscillation signals enter other functional units such as a pulse modulator, so that pulse driving signals are generated and pass through gate amplifiers U6-U7 to drive Q1-Q4.
An alternating input voltage Vin is filtered by Lf and Cf, and rectified by D1-D4 to obtain a rectified voltage | Vin | which is respectively connected to one ends of Ci1 and Ci2 and input ends of two series-connected stacked switch units, namely to a drain of Q1 and a source of Q4, the other ends of Ci1 and Ci2 are connected together and connected to a source of Q2 and a drain of Q3, and Q1 and Q3 are connected to the drain of Q2 and the drain of Q3The source is connected to the drains of Q2 and Q4 respectively. The controller outputs driving signals and provides the driving signals to Q1-Q4 through gate-level driving amplifying circuits U6-U9, high-frequency square waves are provided to primary windings of resonant tanks Lr, Cr and T1 after switching power conversion, a secondary winding of T1 provides direct-current output voltage Vo to a load R after passing through output rectifying circuits D5 and D6 and Co filteringL. U1 samples the DC output voltage, adjusts the DC output voltage and realizes voltage stabilization through the proportional integral or derivative (PID) compensation design of the corresponding voltage outer ring. The rectified voltage | vin | waveform is a sine-changing steamed bread wave, the instantaneous value of which is a continuously changing alternating current waveform and is provided to the power conversion circuit, and the direct conversion technology of alternating current input instantaneous voltage and direct current output voltage is also called as the direct conversion technology because the previous PFC converter in the traditional two-stage AC/DC converter can be saved. The U4 samples the primary side input current Iin, and due to the multiplier U3, the waveform of the alternating current input current can infinitely trend to the rectified voltage or the alternating current input voltage at the input end of the U3 through the corresponding current inner loop proportional integral or differential (PID) compensation design, so that the input current tracks the alternating current input frequency and phase in real time to realize the function of Power Factor Correction (PFC). It should be noted that all voltage and current sampling signals in the controller generally adopt a voltage dividing resistor detection form, signals at the input end of the multiplier also need to adopt corresponding filtering measures, and the voltage error amplifier and the current error amplifier both need to adopt PID compensation, and can adopt different compensation forms such as second-order or multi-order compensation and PI or PID.
In order to adapt to a wider alternating current input voltage, the three-level conversion circuit adopts two-level control modes, and the resonant converter can work in a low-voltage section or a high-voltage section according to different input voltages. The working principle of the ac input low-voltage section is shown in fig. 8-12, in this control mode, the high-frequency filter capacitors Ci1 and Ci2 are commonly connected in series for filtering, fig. 8 is a main working waveform, and the main working waveform includes a midpoint voltage V34 of the gate driving signals of the power switching tubes Q1 and Q4, the gate driving signals Q2 and Q3, the gate driving signals Q1/Q2 and Q3/Q4, and a resonant current ir flowing through the series resonant inductor Lr from top to bottom. Fig. 9 to 12 are equivalent circuits of the switching mode, and the specific working process is as follows:
(t 0-t 1) the power switches Q1 and Q4 have achieved zero voltage turn-onWhen the resonant tank is switched on (ZVS), Q2 and Q3 are continuously switched off, an input rectified voltage | vin | is applied to a parallel resonant inductor Lp and a primary side winding of a main transformer T1 (namely, an LLC resonant circuit is formed) after passing through a resonant tank series resonant inductor Lr and a series resonant capacitor Cr, the amplitude of a midpoint voltage V34 is | vin |, a resonant current ir is approximate to a forward sine wave, a secondary side winding of the resonant circuit is rectified by a diode D5, filtered by an output filter capacitor Co and then provided for a load RL。
During the period from t1 to t2, Q1 and Q4 start to turn off, Q2 and Q3 continue to turn off, and the magnitudes of drain and source voltages of Q1 and Q4 are equal to | vin |, which is also called the first dead zone. Lp current passes through primary windings of Lr, Cr and T1 and a body diode DQ2And DQ3Freewheeling, where the midpoint voltage V34 begins to resonate down to zero and ir begins to fall, where no energy is provided to the output on the primary side and Co provides energy to the load RL. Note that the gate drive signals must be applied to Q2 and Q3 before the Lp current is ramped down to zero to achieve ZVS turn-on.
During the period from T2 to T3, Q1 and Q4 are continuously turned off, Q2 and Q3 realize ZVS on, the voltage at two ends of Cr is applied to Lp and T1 primary windings through Lr in opposite phases (namely, the same LLC resonant circuit is formed), the midpoint voltage V34 is zero, the resonant current ir is approximate reverse sine wave, the secondary winding is rectified through D6, filtered by Co and then provided for a load RL。
During the period from t3 to t4, Q1 and Q4 continue to turn off, Q2 and Q3 start to turn off, the magnitudes of drain and source voltages of Q2 and Q3 are equal to | vin |, and this period is the second dead zone. Lp current passes through Ci1, Ci2, Lr, Cr, T1 primary winding and body diode DQ1And DQ4Freewheeling, the midpoint voltage V34 begins to rise from zero at resonance and ir begins to fall in reverse, no energy is provided on the primary side to the output, and Co provides energy to the load RL. Note that the gate drive signals must be applied to Q1 and Q4 before the Lp current reverses to zero to achieve its ZVS turn-on.
From the above, it can be seen that the duty cycles of the four power switching tubes Q1-Q4 are all 0.5, and the illustration adopts frequency conversion control, and optionally uses phase-shift PWM control. In fact, the resonant frequency under the duty ratio setting and the variable frequency control condition is the same as the switching frequency of the switching tube.
The working principle of the alternating-current input high-voltage section is shown in fig. 13-21, the high-frequency filter capacitors Ci1 and Ci2 in the control mode are respectively independent, fig. 13 is a main working waveform, and the main working waveform comprises gate driving signals Vgs of power switching tubes Q1-Q4, midpoint voltage V34 of Q1/Q2 and Q3/Q4 and resonant current ir flowing through a series resonant inductor Lr from top to bottom. Fig. 14 to 21 are equivalent circuits of the switching mode, and the specific working process is as follows:
during the period (T0-T1), the power switch tube Q1 has already realized ZVS and turned on, Q3 continues to be turned on, Q2 and Q4 continue to be turned off, half of the rectified voltage | vin |/2, namely the voltage at two ends of the high-frequency filter capacitor Ci1 passes through the resonant tank series resonant inductor Lr and the series resonant capacitor Cr and then is applied to the parallel resonant inductor Lp and the primary winding of the main transformer T1 (namely to form a first LLC resonant circuit), the amplitude of the midpoint voltage V34 is | vin |/2, the resonant current ir is an approximate forward sine wave, the secondary winding is rectified by the diode D5 and is provided to the load R after being filtered by the output filter capacitor CoL。
During the period from t1 to t2, Q1 starts to turn off, Q3 continues to turn on, Q2 and Q4 continue to turn off, and the amplitude of the drain and source voltages of Q1 is equal to | vin |/2, which is also called as a first dead zone. Lp current passes through primary windings of Lr, Cr and T1 and a body diode DQ2Freewheeling, where the midpoint voltage V34 begins to resonate down to zero and ir begins to fall, where no energy is provided to the output on the primary side and Co provides energy to the load RL. Note that the gate drive signal must be applied to Q2 before the Lp current is ramped down to zero to achieve its ZVS turn-on.
(T2-T3) the Q1 and Q4 are turned off continuously, the Q2 realizes ZVS on and Q3 is turned on continuously, the voltage at two ends of Cr is applied to primary windings of Lp and T1 in opposite phases through Lr, the midpoint voltage V34 is zero, the resonant current ir is approximate reverse sine wave, the secondary winding is rectified through D6, filtered by Co and provided for the load RL。
During the period from t3 to t4, Q1 and Q4 are turned off continuously, Q2 is turned on continuously, Q3 is turned off, the amplitude of the drain and source voltages of Q3 is equal to | vin |/2, and the period is the second dead zone. Lp current flows through Ci2, Lr,Cr, T1 primary winding, Q2 and body diode DQ4Freewheeling, the midpoint voltage V34 begins to rise from zero at resonance and ir begins to fall in reverse, no energy is provided on the primary side to the output, and Co provides energy to the load RL. Note that the gate drive signal must be applied to Q4 before the Lp current reverses to zero to achieve its ZVS turn-on.
During the period (T4-T5), Q2 is continuously conducted, Q4 achieves ZVS opening, Q1 and Q3 are continuously turned off, half of rectified voltage | vin/2, namely voltage at two ends of a high-frequency filter capacitor Ci2 passes through Lr and Cr and then is applied to primary windings of Lp and T1 (namely, a second LLC resonant circuit is formed), the amplitude of the midpoint voltage V34 is | vin/2, the resonant current ir is approximate to a forward sine wave, a secondary winding of the resonant current is rectified by a diode D5, filtered by an output filter capacitor Co and then provided for a load RL。
During the period from t5 to t6, Q2 continues to be turned on, Q4 starts to be turned off, Q1 and Q3 continue to be turned off, the amplitude of the drain and source voltages of Q4 is equal to | vin |/2, and the period is also called as a third dead zone. Lp current passes through primary windings of Lr, Cr and T1 and a body diode DQ3Freewheeling, where the midpoint voltage V34 begins to resonate down to zero and ir begins to fall, where no energy is provided to the output on the primary side and Co provides energy to the load RL. Note that the gate drive signal must be applied to Q3 before the Lp current is ramped down to zero to achieve its ZVS turn-on.
During the period from T6 to T7, Q1 and Q4 are continuously turned off, Q2 is continuously turned on, Q3 is turned on with ZVS, the voltage at two ends of Cr is applied to primary windings of Lp and T1 in opposite phases through Lr, the midpoint voltage V34 is zero, the resonant current ir is approximate reverse sine wave, the secondary winding is rectified through D6, filtered by Co and provided for a load RL。
During the period from t7 to t8, Q1 and Q4 are turned off continuously, Q2 is turned off and Q3 is turned on continuously, the amplitude of the drain voltage and the source voltage of Q2 is equal to | vin |/2, and the period is a fourth dead zone. Lp current passes through Ci1, Lr, Cr, T1 primary winding, Q3 and body diode DQ1Freewheeling, the midpoint voltage V34 begins to rise from zero at resonance and ir begins to fall in reverse, no energy is provided on the primary side to the output, and Co provides energy to the load RL. Need to pay attention toThat is, the gate drive signal must be applied to Q1 before the Lp current reverses to zero to achieve its ZVS turn-on.
From the above, it can be seen that the duty ratio of two outer tubes Q1, Q4 in the four power switching tubes is 0.25, and the duty ratio of two inner tubes Q2, Q3 is 0.75, and the illustration adopts frequency conversion control, and optionally uses phase shift PWM control. In fact, the resonant frequency under this duty cycle setting is twice the switching frequency of the power switch. Alternatively, the duty ratio of the two outer tubes Q1 and Q4 can be set to be 0.75, and the duty ratio of the two inner tubes Q2 and Q3 can be set to be 0.25, so that the midpoint voltage will be changed in a high-frequency square wave between | vin |, | vin |/2, and the midpoint voltage can also be used in the ac input low-voltage section working condition, and the resonant frequency is also twice the switching frequency of the power switch tube under the duty ratio setting and the variable frequency control condition.
When the alternating-current input voltage changes, the controller can adaptively control the power conversion circuit, flexibly switch the power conversion circuit into one or more basic switch units to be in series-parallel connection and stacked work, and adopt two-level control modes: high-frequency full voltage and low-frequency partial voltage. When alternating current is input into a low-voltage section, two of the four power switching tubes are connected in series, and the four power switching tubes can be regarded as single tubes essentially, the circuit structure is similar to a traditional LLC half-bridge resonant converter, the switching frequency is the same as the switching frequency, and meanwhile, the midpoint voltage is | vin |, so that the high-frequency full-voltage mode is called; when alternating current is input into a high-voltage section, four power switching tubes respectively and independently work, the circuit structure is similar to that two traditional LLC half-bridge resonant converters are stacked in series, if the resonant frequency of the resonant converters is set to be the same as that of the low-voltage section, the switching frequency of the power switching tubes can be halved, and meanwhile, the midpoint voltage is | vin |/2, so that the low-frequency full-voltage mode is called. When the power conversion circuit works in a high-frequency full-voltage mode, the input rectified voltage is, for example, | vin |, and the midpoint voltage V34 output by the power conversion circuit is 0 and | vin | -high-frequency square wave. If the input rectified voltage is changed into 2| vin |, the resonant converter is changed into a low-frequency voltage division mode, voltages at two ends of Ci1 and Ci2 are | vin |, the output midpoint voltage of the power conversion circuit is also 0 and | vin | -high-frequency square waves, and the equivalent switching frequency of a MOSFET (metal-oxide-semiconductor field effect transistor) of the power switch tube is half of the resonant frequency. The two modes use the same resonant circuit, and can share the same set of series resonant inductor Lr, parallel resonant inductor Lp and series resonant capacitor Cr, so that the design of a resonant tank can be simplified and the cost of the device can be reduced by sharing the same set of resonant device.
It can be seen from the above that, under the condition of using the same resonant circuit, when the alternating-current input voltage is in a low-voltage section, the resonant converter works in a high-frequency full-voltage mode, when the power switch tube is turned off, the voltage of the drain and the source is | vin |, and the switching frequency is fs; when the AC input voltage is in a high-voltage section, the resonant converter works in a low-frequency voltage division mode, the voltage of the drain electrode and the source electrode of the resonant converter is reduced to | vin |/2 when the power switch tube is turned off, and the switching frequency of the resonant converter is reduced to fs/2. When the switching frequency fs is less than the resonant frequency fr, ZVS (zero voltage switch) switching-on or ZCS (zero current switch) switching-off can be realized by the power switch tube and the secondary rectifier diode in the two control modes, and the switching loss of the resonant converter can be reduced, so that the whole conversion efficiency of the low-high voltage section in the whole range is improved. When the high-frequency full-voltage control mode power switch tube is turned off, the voltage of the drain and the source is | vin |, but the alternating-current input voltage is lower, so the actual voltage stress of the power switch tube is lower. Even if the ac input voltage is high, the voltage of the drain and the source of the power switch is | vin |/2 when the power switch is turned off, so the voltage stress of the power switch is still low. The power tubes are soft switches under different working conditions, so that the turn-off loss can be reduced, and meanwhile, the low-frequency voltage division control mode has lower equivalent switching frequency, so that the switching loss is greatly reduced, and the conversion efficiency is further improved. Due to the two control modes, the midpoint voltage and the resonant current in the resonant tank are basically unchanged, so that the power losses of the resonant tank, the main transformer and the secondary rectifier diode are approximately unchanged, and meanwhile, two power switch tubes are conducted on the resonant current path or one power switch tube is conducted with one body diode, namely, the power losses of the four power switch tubes are basically the same, so that the overall conversion efficiency is approximately unchanged when the alternating-current input voltage is changed by two times.
The characteristic is particularly suitable for a wide alternating current input voltage range, for example, under the condition that the input voltage range is 85-276 Vac, the alternating current input voltage can be divided into a low voltage section and a high voltage section, and the determination of the critical point voltage is related to the rated voltage selection of the power switch tube. When the low voltage section is 85-140 Vac, high frequency is adoptedA full pressure control mode; and in the high-voltage section 140-276 Vac, a low-frequency voltage division control mode is adopted. The critical point of the alternating-current input voltage is selected to be 140V, so that a power switch tube MOSFET with the rated voltage of 200V can be used, and the lower-voltage-resistant power switch tube has better electrical performance and lower device cost. Therefore, the resonant tank needs to handle a voltage gain variation range of
Figure BDA0003115855650000101
To
Figure BDA0003115855650000102
Namely, the variation range of the AC input voltage is 2 times, while the variation range of the voltage gain of the traditional LLC resonant converter is
Figure BDA0003115855650000103
To
Figure BDA0003115855650000104
Namely, the variation range of the AC input voltage is 3.25 times, so that the equivalent variation range of the AC input voltage can be narrowed to the original range which is close to 0.6 time. The global power grid has different voltage levels, and the common voltage levels are 120Vrms and 230Vrms (230/2-115), so that the optimal operating point of the resonant frequency can be designed.
A front-stage PFC converter and a rear-stage LLC half-bridge resonant converter in a switching power supply are cascaded to form a two-stage power topology, the PFC converter outputs direct-current bus voltage and serves as LLC half-bridge resonant converter input, and direct-current voltage is finally output, so that the latter is a DC/DC converter. In practice, the ac input voltage sine wave may be regarded as a multi-segment waveform, and each segment corresponds to a dc voltage operating point, i.e. the ac input voltage sine wave may be subdivided into segments of dc voltage. By adopting frequency multiplication technology and two control modes and switching the working modes of the power conversion circuit, the basic switch unit and the peripheral resonant circuit thereof form an LLC half-bridge resonant converter, and the LLC half-bridge resonant converter can directly convert alternating current input instantaneous voltage into direct current output voltage by utilizing the inherent gain characteristic and meet various voltage gains required by a PFC function. The ac input voltage is a sinusoidal waveform, and its instantaneous value is:
Figure BDA0003115855650000105
wherein VinRMSThe effective value of the AC input voltage is theta, and the phase angle of the sine wave is theta. If the dc output current is Io, the dc output power is:
Po＝Vo·Io (2)
under the condition of realizing the function of power factor correction, namely the alternating current input current is also in a sine wave shape, the instantaneous value of the output power is as follows:
Figure BDA0003115855650000106
wherein IinRMSThe effective value of the alternating input current. To simplify the calculation, assuming that the conversion efficiency η is 1, one can obtain:
Pout＝Vo·Io＝η·VinRMS·IinRMS (4)
the instantaneous output power is therefore:
pout(θ)＝2·sin(θ)2·Po (5)
therefore, when the phase angle of the alternating-current input voltage changes, the instantaneous output power of the LLC half-bridge resonant converter changes in proportion, and the current peak values of the power switch tube and the primary and secondary windings of the main transformer are correspondingly doubled. Because the input rectified voltage-
Figure BDA0003115855650000111
Figure BDA0003115855650000112
The gain requirement of the PFC is firstly obtained in order to realize PFC and obtain stable DC output voltage, and the gain requirement of the PFC is as follows:
Figure BDA0003115855650000113
it follows that the maximum PFC voltage gain occurs at the maximum output voltage and the minimum input voltage. For the LLC half-bridge resonant converter, Lr is the series resonant inductor, Lp is the parallel resonant inductor, and Cr is the series resonant capacitor, so the voltage gain is:
Figure BDA0003115855650000114
where k is the inductance ratio, i.e., k ═ Lp/Lr, and Q is the resonance quality factor. Resonant converters have their inherent property of voltage gain reduction at heavy loads and vice versa. In order to realize the PFC function, the voltage gain of the LLC half-bridge resonant converter can be set as follows: the gain is minimum when the frequency converter works near the series resonance frequency fr and maximum when the frequency converter works near the parallel resonance frequency fp, and the gain requirements are met under all conditions, such as different direct current output voltages, different alternating current input voltage phase angles and the like. The fundamental wave analysis can obtain:
Figure BDA0003115855650000115
wherein the minimum value of Q occurs at the maximum output voltage, n is the primary and secondary side turn ratio of the main transformer, and for the LLC half-bridge resonant converter:
Figure BDA0003115855650000116
where the nmax occurs at the maximum input voltage and the minimum output voltage. To achieve primary side power switching tube soft switching or Zero Voltage Switching (ZVS), the LLC resonant converter must operate in the inductive region, i.e., the switching frequency must be higher than the resonant tank parallel resonant frequency fp:
Figure BDA0003115855650000117
in addition, to realize secondary side rectifier diode soft switching or Zero Current Switching (ZCS), the LLC resonant converter must have a switching frequency lower than the resonant tank series resonant frequency fr:
Figure BDA0003115855650000118
the switching frequency fs range of the AC/DC resonant converter is (fp < fs < fr) or (fs > fr), and cannot be present under any condition (fs < fp). The primary power switching tube ZVS and the secondary rectifier diode ZCS can be realized when the switching frequency range is (fp < fs < fr), and the primary power switching tube ZVS can only be realized when the switching frequency range is (fs > fr), but the secondary rectifier diode is hard switch, thereby increasing the switching loss and influencing the conversion efficiency.
When the sine wave phase angle theta of the AC input voltage is changed between (0-pi), the instantaneous output power p can be obtainedoutThe curves of (θ), PFC gain requirement G (θ) and LLC half-bridge voltage gain M (θ) are shown in fig. 23. In order to realize the PFC function, the preconditions must be satisfied throughout the sine wave input interval: m (theta) is not less than G (theta). As can be seen from the figure, when the phase angle of the sine wave is 90 degrees, the required voltage gain is lowest, and the instantaneous output power is maximum; the voltage gain is highest and the instantaneous output power is minimum at the zero crossing. The characteristic completely matches with a voltage gain curve of a traditional LLC half-bridge converter, namely that the voltage gain is increased when the load is light and the voltage gain is decreased when the load is heavy. Through the design process, the LLC half-bridge resonant converter can meet the voltage gain required by the PFC function.
In summary, the controller has a multiplier function unit therein, and the input current tracks the ac input frequency and phase in real time to realize a Power Factor Correction (PFC) function. The basic switch unit and the peripheral resonant circuit thereof can form an LLC half-bridge resonant converter, can directly convert the AC input instantaneous voltage into DC output voltage by utilizing the inherent gain characteristic, and can meet the voltage gain required by the PFC function, thereby saving the conventional PFC converter and the DC bus capacitor thereof, improving the conversion efficiency thereof and reducing the component cost.
It should be noted that, although the above operating principle only addresses two segments of low and high ac input voltages, the ac input voltage may be subdivided into more voltage segments, and still be similar to the basic principle of full high-frequency and low-frequency voltage division. Through proper design of resonant tank devices, the LLC half-bridge resonant converter can completely meet the voltage gain required by realizing a PFC function. In addition, the above description is directed to a wide ac input voltage range, and in fact, the basic principles can be applied to a wide dc output voltage range. For example, when high voltage needs to be output, the power conversion circuit works in a high-frequency full-voltage control mode; when the low voltage needs to be output, the power conversion circuit works in a low-frequency voltage division control mode, the basic principle is similar, and the description is not repeated here. The controller can be built by using discrete electronic components, and can also be designed into a special integrated circuit, such as an analog control chip, a singlechip (MCU) programmed by software or a programmable logic device (FPGA/CPLD). The power conversion circuit and the rectifying unit can adopt a discrete device mode or a respective integrated and mixed integrated mode, and can also be uniformly integrated into the controller to form a large-scale mixed integrated circuit, and the design of the high-integration controller can reduce the size of the switching power supply.
When the AC/DC LLC half-bridge resonant converter is used in special situations such as industrial power supplies and communication power supplies, it is generally required to have a higher-level lightning protection requirement, and it is also required to have a power-off holding time requirement, and the embodiment with a lightning protection and holding time extension circuit is the second embodiment, as shown in fig. 24. In fig. 7, the output positive and negative poles of the input rectifying circuit are connected in parallel with a capacitor energy storage branch, which is formed by connecting a power switch tube S, a body diode Ds thereof and a direct current capacitor Cb in series. Optionally, a resistor may be connected in parallel to the both ends of Cb, so that energy may be discharged in an abnormal situation. When the alternating current input end is struck by lightning, the transient lightning impulse stores energy on the direct current capacitor Cb through the body diode Ds, and the lightning surge is absorbed by utilizing the large-capacity storage characteristic of the transient lightning impulse. It should be noted that Cb can still charge and store energy from the ac input even in the absence of lightning strike energy. During normal operation, the power switch tubeS is always in a disconnected state, once the input power failure occurs, the auxiliary power supply in the switching power supply continues to work for a period of time, the controller still keeps working normally in the period of time, so that an S opening signal is sent, Cb discharges in the power failure holding time, and the output load R can be supplied with powerLProviding energy for a short period of time. The energy retention time provided is mainly composed of Cb capacitance and RLAnd (5) determining the size. The working principle of other parts is basically the same as that of the figures 3-22, and the description is not repeated here.
When the AC/DC LLC half-bridge resonant converter is used in special situations such as high-end power supplies that require high-precision output, it is generally required to have a low-ripple output requirement. The AC/DC LLC half-bridge resonant converter in fig. 7 is essentially a single-stage conversion topology, and since there is a power difference between the DC output and the AC input, it will inevitably generate a double-frequency ripple at the output, so that it is possible to reduce its output voltage ripple by using the second-stage LC filter circuit. The first low ripple implementation is embodiment three, as shown in fig. 25. The second stage of LC filter circuit is added to the output end of the LC filter circuit, and mainly comprises a filter inductor Ld and a filter capacitor Co2, and the rest of the LC filter circuit is completely the same as that in fig. 7. The lightning protection and retention time extension circuit shown in fig. 23 may be used. The working principle of other parts is basically the same as that of the figures 3-22, and the description is not repeated here.
When the AC/DC LLC half-bridge resonant converter is used in special situations such as high-end power supplies that require high-precision output, it is generally required to have a low-ripple output requirement. The AC/DC LLC half-bridge resonant converter in fig. 7 is essentially a single-stage conversion topology, and since there is a power difference between the DC output and the AC input, it is inevitable to generate twice the power frequency ripple at the output terminal, so that the output ripple can be reduced by using the cascaded buck DC/DC converter. The second low ripple implementation is embodiment four, as shown in fig. 26. A cascaded Buck DC/DC converter (Buck) is added at its output end, mainly composed of a power switch tube Qa and a body diode DQaThe main inductor La, the flywheel diode Da, and the filter capacitors Cb and Co are the same as those in fig. 7. The LLC half-bridge resonant controller and the Buck controller can use two independent chips and can also be integrated into oneA control chip. The lightning protection and retention time extension circuit shown in fig. 23 may be used. The working principle of other parts is basically the same as that of the figures 3-22, and the description is not repeated here. In addition, the cascade DC/DC is not limited to a Buck converter (Buck), and non-isolated DC/DC converters such as a Boost (Boost) converter and a Buck-Boost (Buck-Boost) converter can be used, and even isolated DC/DC converters such as forward, flyback, push-pull or resonant type converters can be used.
The cascaded buck DC/DC converter has two architectures, the first is a direct current bus architecture: in the mode, Cb adopts a large-capacity electrolytic capacitor or a solid capacitor and the like, two ends of the Cb are direct-current voltages, but larger double-power-frequency ripples exist, and the Buck converter can eliminate the power-frequency ripples at the output end through a quick control loop; the second is a sinusoidal bus architecture: namely, Cb adopts high-frequency filter ceramics or a thin-film capacitor, and the two ends of the Cb adopt sinusoidal steamed bread wave voltage. The sine bus structure has two control modes: in the first control mode, the output is fed back to the LLC half-bridge resonant converter for closed-loop regulation to stabilize only the output voltage or output current, and the PFC function is implemented by the Buck controller, which saves the multiplier in the controller of fig. 6. A current error amplifier is added in the Buck controller, the current error amplifier is provided with a proportional-integral-derivative (PID) compensation circuit, a reference signal of the current error amplifier is taken from the voltage at two ends of Cb, and the frequency and the phase of a cascade DC/DC input current following | vin | are adjusted in a closed loop mode, so that the input current of the resonant converter is indirectly controlled to be a sine wave, and the power factor correction function is realized. In the second control mode, the LLC half-bridge resonant converter operates in an open-loop mode, i.e., it operates like a power electronic transformer, and the output is fed back to the Buck converter for closed-loop feedback regulation and stabilization of the output voltage or output current. Instead of using a multiplier in the controller of the original 6, a current error amplifier is added, and the current error amplifier has a proportional-integral-derivative (PID) compensation circuit, the reference signal of which is taken from the rectified voltage | vin |, and the frequency and phase of the input current following | vin |, are adjusted in a closed loop, so as to realize the power factor correction function. The sinusoidal bus architecture can not completely eliminate output power frequency ripples, and a post-stage DC/DC converter can be cascaded, so that the sinusoidal bus architecture has the advantage that the LLC half-bridge resonant converter is close to the open-loop optimal operating point.
When the AC/DC LLC half-bridge resonant converter is used in special situations such as high-end power supplies that require high-precision output, it is generally required to have a low-ripple output requirement. The AC/DC LLC half-bridge resonant converter in fig. 7 is essentially a single-stage conversion topology, and since there is a power difference between the DC output and the AC input, it is inevitable to generate a double-frequency ripple at the output terminal, so that the output voltage ripple can be reduced by using the series step-down DC/DC converter. The third low ripple implementation is embodiment five, as shown in fig. 26. The circuit form is composed of a main converter and an auxiliary converter, and the main converter and the auxiliary converter are corresponding to each other. Main transformer T1 has two secondary windings: the main winding of the main converter forms main output after being rectified and filtered by D5, D6 and Co 1; the auxiliary winding of the auxiliary converter forms a direct current bus voltage Vb through D7, D8 and Cb rectification and filtering, and then passes through a power switch tube Qa and a body diode D thereofQaThe main inductor La, the freewheeling diode Da and the filter capacitor Co2 form a Buck DC/DC converter (Buck) and then form an auxiliary output, and the main output and the auxiliary output are connected in series, so that the total output voltage is obtained and provided for an output load RLThe other parts are identical to those of fig. 7.
Through the corresponding cooperation design of main and auxiliary controller, realize that the auxiliary output is opposite with main output voltage power frequency ripple phase place to total output voltage ripple can reduce even eliminate. The corresponding design of the main controller and the auxiliary controller has multiple implementation modes, for example, the main output voltage or the auxiliary output voltage is fed back to a voltage error amplifier in the main controller, the PID of the main output voltage or the auxiliary output voltage is designed to have a slower response speed, meanwhile, a primary side current sampling signal is fed back to a current error amplifier in the main controller, the PID of the main output voltage is designed to have a faster response speed, so that the input current waveform can conveniently track the alternating current input voltage, the total output voltage is fed back to the voltage error amplifier in the auxiliary controller, and the PID design of the main output voltage or the auxiliary output voltage realizes the output of stabilized current and stabilized voltage. Or the total output voltage is fed back to a voltage error amplifier in the main controller, the PID is designed to have a slower response speed, meanwhile, the primary side current sampling signal is fed back to a current error amplifier in the main controller, the PID is designed to have a faster response speed so as to enable the input current waveform to track the alternating current input voltage conveniently, the main or auxiliary output voltage is fed back to the voltage error amplifier in the auxiliary controller, and the PID design realizes the output of stabilized current and stabilized voltage. In general, the current control loop needs to be fast enough to implement PFC, and there is and only one voltage control loop to adjust the total output voltage and avoid main and auxiliary control conflicts, which are not necessarily exemplified here. The main controller and the auxiliary controller can use two independent chips or can be integrated into one control chip. The lightning protection and retention time extension circuit shown in fig. 23 may be used. The working principle of other parts is basically the same as that of the figures 3-22, and the description is not repeated here. In addition, the series DC/DC is not limited to a Buck converter (Buck), and non-isolated DC/DC converters such as a Boost converter (Boost) and a Buck-Boost converter (Buck-Boost) can be used, and even isolated DC/DC converters such as forward, flyback, push-pull or resonant type can be used.
If the range of the AC input voltage or the DC output voltage is narrow, the automatic switching function of the AC input voltage range is not used, other types of three-level conversion topologies can be used instead, the power factor correction is realized by the resonant converter, and the functions of voltage transformation and isolation are considered at the same time. The power conversion circuit uses a diode-clamped AC/DC three-level LLC half-bridge resonant converter as shown in fig. 27 for example six. Compared with fig. 7, only the wiring form of the basic switching unit is changed. The lightning protection and holding time extension circuit shown in fig. 23 and the output low ripple circuits shown in fig. 24 to 26 may be used. The working principle of other parts is basically similar to that of figures 3-22, and the description is not repeated here. It should be noted that the diode-clamped three-level half-bridge may be replaced by a three-level half-bridge topology such as Flying Capacitor (FC) and cascade, and the basic principle is similar, and the description is not repeated here.
If the range of the AC input voltage or the DC output voltage is narrow, the automatic switching function of the AC input voltage range is not used, other types of three-level conversion topologies can be used instead, the power factor correction is realized by the resonant converter, and the functions of voltage transformation and isolation are considered at the same time. The power conversion circuit uses a diode-clamped AC/DC three-level LLC full-bridge resonant converter as an embodiment seven, as shown in fig. 28. Compared with fig. 7, only the wiring form of the basic switching unit is changed. The lightning protection and holding time extension circuit shown in fig. 23 and the output low ripple circuits shown in fig. 24 to 26 may be used. The working principle of other parts is basically similar to that of figures 3-22, and the description is not repeated here. It should be noted that the diode-clamped three-level full bridge may be replaced by a Flying Capacitor (FC), a cascaded three-level full bridge topology, and the basic principle is similar, and the description is not repeated here.
In order to increase output current or output power, the AC/DC LLC half-bridge resonant converter can also form four multiphase series-parallel circuit forms, namely input parallel/output parallel, input parallel/output series, input series/output series and input series/output parallel, and the input parallel circuit form can also adopt an interleaving parallel technology. The four forms are basically similar in operation principle, wherein the two-phase input series/output parallel circuit form is embodiment eight, as shown in fig. 29, a similar configuration can also be used for the multiphase series parallel circuit form. Compared with the fig. 7, the second set of LLC half-bridge resonant converter is mainly added, and the first and second sets of inputs work directly in series, or alternatively, a phase-interleaved 180 degree control scheme may be used. It should be noted that the lightning protection and hold time extension circuit shown in fig. 23, the output low ripple circuits shown in fig. 24 to 26, and other types of three-level topologies shown in fig. 27 to 28 may also be used. The working principle of other parts is basically the same as that of the figures 3-22, and the description is not repeated here.
The foregoing is described with reference to a single-phase AC input system, and in fact, the AC/DC LLC half-bridge resonant converter can also be used in a three-phase AC input system. The three-phase alternating current has two connection methods of triangle and star, and has two connection modes of three-phase three-wire system and three-phase four-wire system, and the neutral wire (N) of the star connection mode in the three-phase four-wire system is connected to the ground to form a zero wire. The basic concepts of three-phase alternating current triangular connection, star connection, zigzag polygon connection and three-phase three-wire system and three-phase four-wire system connection modes are similar, only phase voltage and line voltage and current amplitude and phase difference exist, and the main change is that corresponding functional units such as three-phase current sampling, conditioning and control and the like of a controller need to be added. The three-phase three-wire star connection is the ninth embodiment, and as shown in fig. 30, a similar configuration can be used for other connection or wiring modes. The three-phase alternating current adopts a star connection method, each phase voltage is Va, Vb and Vc respectively, the common point of the three-phase alternating current is N, Lfa and Cfa, Lfb and Cfb and Lfc and Cfc are an A-phase EMI filter inductor, a B-phase EMI filter inductor and a C-phase EMI filter capacitor respectively, the three-phase EMI filter inductor and the filter capacitor are connected to one input end of an input rectifying circuit respectively, and the Cfa, Cfb and Cfc are connected to a virtual ground 0. Three sets of LLC half-bridge resonant converters are completely symmetrical, the other input ends of the three input rectifying circuits are connected together and connected to a virtual ground 0, and the three direct current outputs form a parallel connection relation. In order to realize three-phase power factor correction, the controller needs to shift the phase of each LLC half-bridge resonant converter by 120 degrees. Optionally, the direct current output can be changed into a series relation, and only the output sampling and control mode needs to be adjusted. When the output of a three-phase alternating current system formed by the AC/DC LLC half-bridge resonant converter works in parallel, the sum of three-phase alternating current power frequency ripples is mostly counteracted, and the output voltage power frequency ripples can be greatly reduced, so that a ripple removing implementation mode is not required to be additionally adopted. It should be noted that the lightning protection and holding time extension circuit shown in fig. 23, the output low ripple circuits shown in fig. 24 to 26, other types of three-level topologies shown in fig. 27 to 28, and the output power increase method shown in fig. 29 may be used. The working principle of other parts is basically the same as that of the figures 3-22, and the description is not repeated here.
The above-mentioned three-level conversion circuit is taken as an example to illustrate, two-level control modes of high-frequency full voltage and low-frequency voltage division are essentially used, and actually, a power conversion circuit can also use a higher level number. The multi-level conversion circuit topology mainly comprises a neutral point clamp, a flying capacitor, a cascade connection and a hybrid type, the multi-level implementation principle is basically similar, wherein the five-level conversion circuit is an embodiment ten, and as shown in fig. 31, similar structures can also be used for other types of multi-level topologies. The power conversion circuit is formed by stacking four basic switch units in series, wherein a high-frequency filter capacitor Ci1, power switch tubes Q1, Q4, and body diodes D of the power switch tubes Q1, Q4Q1、DQ4A first basic switch unit, a high-frequency filter capacitor Ci2, power switch tubes Q5, Q8 and powerBody diode D of switching tube Q5, Q8Q5、DQ8A body diode D constituting a second basic switching cell, flying capacitor Ci3, power switching tubes Q2, Q3, and power switching tubes Q2, Q3Q2、DQ3A body diode D constituting a third basic switching cell, flying capacitor Ci4, power switching tubes Q6, Q7, and power switching tubes Q6, Q7Q6、DQ7Constituting a fourth basic switching unit. The output midpoint of the third basic switch unit is connected to one end of the series resonance inductor Lr, and the output midpoint of the fourth basic switch unit is connected to one end of the series resonance capacitor Cr; the other ends of the Lr and the Cr are respectively connected to two ends of a parallel resonant inductor Lp, and two ends of the Lp are connected with two ends of a primary winding of a main transformer T1 in parallel. The rectifying unit adopts a transformer center tap diode full-wave rectification structure, the anodes of rectifying diodes D5 and D6 are respectively connected to two ends of a T1 secondary winding, the cathodes of the rectifying diodes D5 and D6 are connected with the anode of an output filter capacitor Co, the cathode of Co is connected to a T1 center tap, and an output load RLTwo ends are respectively connected to the Co anode and the Co cathode. The series resonance inductor Lr and the parallel resonance inductor Lp can use independent inductors, and can also respectively use leakage inductance and excitation inductance of the main transformer T1, so that the magnetic integration design is realized. Through different control modes, the two midpoint voltages V34 of the power conversion circuit have different high-frequency square wave level combinations, such as | vin | and 0, | vin | and | vin | 2, | vin | and 0, | vin |/2 and | vin |/4, | vin |/4 and 0, and thus have more control freedom. Due to the fact that the number of the levels is larger, the alternating current input voltage range can be applied to a wider alternating current input voltage range. The multi-level converter is derived from a multi-level inverter circuit, the working principle of which is basically similar and will not be described in detail here. The lightning protection and hold time extension circuit shown in fig. 23, the output low ripple circuits shown in fig. 24 to 26, the other types of three-level topologies shown in fig. 27 to 28, the output power increase method shown in fig. 29, and the three-phase ac input system shown in fig. 30 may be used. The working principle of other parts is basically the same as that of the figures 3-22, and the description is not repeated here.
From the foregoing, it can be seen that a greater number of levels can accommodate a wider ac input voltage range. Correspondingly, the range of the AC input voltage or the DC output voltage is narrow, the function of automatically switching the AC input voltage range can be omitted, the power factor correction can be realized by using the two-level conversion circuit, and the functions of voltage transformation and isolation are simultaneously realized. The power conversion circuit uses an LLC half-bridge resonant converter as an embodiment eleven, as shown in fig. 32. Compared with fig. 7, only one basic switch unit is used, so that the number of power devices is reduced, and the cost of the devices is reduced. The lightning protection and holding time extension circuit shown in fig. 23, the output low ripple circuits shown in fig. 24 to 26, the output power increase method shown in fig. 29, and the three-phase ac input system shown in fig. 30 may be used. The working principle of other parts is basically similar to that of figures 3-22, and the description is not repeated here.
The power conversion circuits are all based on an LLC half-bridge resonant converter, and actually, other types of two-level resonant conversion circuits can be used for realizing power factor correction and simultaneously have the functions of voltage transformation and isolation. The power conversion circuit uses an LCC half-bridge resonant converter as an embodiment twelve, as shown in fig. 33. As compared with fig. 32, it is only necessary to change the parallel resonant inductance Lp to the parallel resonant capacitance Cp. The LCC resonant converter is similar to the LLC basic principle and is generally applied to direct-current output high-voltage occasions. It should be noted that two or more basic switch units can be stacked in series, and two-level control methods as shown in fig. 7 to 22 can be used, and a lightning protection and holding time extension circuit shown in fig. 23, an output low ripple circuit shown in fig. 24 to 26, other types of three-level topologies shown in fig. 27 to 28, an output power increase method shown in fig. 29, a three-phase ac input system shown in fig. 30, and a higher-level implementation method shown in fig. 31 can also be used. The working principle of other parts is basically similar to that of figures 3-22, and the description is not repeated here.
The power conversion circuit uses a CLCL half-bridge resonant converter as embodiment thirteen, as shown in fig. 34. Compared with fig. 32, the parallel resonant inductance Lp is mainly changed to the second series resonant inductance Ls, and a parallel resonant capacitance Cs is added. The CLCL resonant converter is similar to the LLC basic principle, and can further reduce the resonant current amplitude at turn-off. It should be noted that two or more basic switch units can be stacked in series, and two-level control methods as shown in fig. 7 to 22 can be used, and a lightning protection and holding time extension circuit shown in fig. 23, an output low ripple circuit shown in fig. 24 to 26, other types of three-level topologies shown in fig. 27 to 28, an output power increase method shown in fig. 29, a three-phase ac input system shown in fig. 30, and a higher-level implementation method shown in fig. 31 can also be used. The working principle of other parts is basically similar to that of figures 3-22, and the description is not repeated here.
The power conversion circuit uses an LCLC half-bridge resonant converter as an embodiment fourteen, as shown in fig. 35. Compared with fig. 32, the parallel resonant inductor Lp is connected in series with a parallel resonant capacitor Cp and then connected in parallel with the primary winding of the main transformer T1. The CLCL resonant converter, similar to the LLC basic principle, can further extend the input or output voltage variation range. It should be noted that two or more basic switch units can be stacked in series, and two-level control methods as shown in fig. 7 to 22 can be used, and a lightning protection and holding time extension circuit shown in fig. 23, an output low ripple circuit shown in fig. 24 to 26, other types of three-level topologies shown in fig. 27 to 28, an output power increase method shown in fig. 29, a three-phase ac input system shown in fig. 30, and a higher-level implementation method shown in fig. 31 can also be used. The working principle of other parts is basically similar to that of figures 3-22, and the description is not repeated here.
For applications with low power factor requirements, a simplified three-phase resonant conversion circuit form may also be used. If the range of the AC input voltage or the DC output voltage is narrow, the power factor correction can be realized by using the two-level conversion circuit, and the functions of voltage transformation and isolation are considered at the same time. Fifteen embodiments are three-phase AC/DC LLC half-bridge resonant converters, as shown in fig. 36. Compared with the graph 30, only one set of three-phase diode uncontrolled rectifying circuit is used, so that the number of power devices is reduced, and the cost of the devices is reduced. It should be noted that the lightning protection and holding time extension circuit shown in fig. 23, the output low ripple circuits shown in fig. 24 to 26, the other types of three-level topologies shown in fig. 27 to 28, the output power increase mode shown in fig. 29, the higher level implementation mode shown in fig. 31, and the other types of resonant converters shown in fig. 33 to 35 may also be used. The working principle of other parts is basically similar to that of figures 3-22, and the description is not repeated here.
The half-bridge resonant converter adopts a series stacking mode, and actually can also adopt two parallel stacking modes, and simultaneously has the functions of voltage transformation and isolation. The power conversion circuit uses an LLC full-bridge resonant converter as an embodiment sixteen, as shown in fig. 37. Compared with fig. 7, only four series power switch tubes are changed into a two-by-two parallel form, and only one high-frequency filter capacitor Ci is used. The lightning protection and holding time extension circuit shown in fig. 23 may be used, or the output low ripple circuit shown in fig. 24 to 26, the output power increasing method shown in fig. 29, the three-phase ac input system shown in fig. 30, and the other types of resonant converters shown in fig. 33 to 35 may be used.
Similar to fig. 8 to 21, in order to adapt to a wider ac input or dc output voltage range, the full-bridge conversion circuit adopts two control modes, and when the ac input voltage is lower, the resonant converter operates in the full-bridge mode to obtain a larger voltage gain; when the ac input voltage is high, the resonant converter operates in the half-bridge mode, and the voltage gain is reduced to half compared to the full-bridge mode. The same resonance circuit is used in the control mode, and the same set of series resonance inductor Lr, parallel resonance inductor Lp and series resonance capacitor Cr can be shared, so that the design of the resonance tank can be simplified and the cost of the device can be reduced by sharing the same set of resonance device. The resonant converter can work in a low-voltage section and a high-voltage section according to different input voltages. When the ac input low-voltage section operates in the full-bridge mode, the switching mode equivalent circuit is as shown in fig. 38 to 41, and the specific working process is as follows:
during the period from T0 to T1, zero voltage switching-on (ZVS) of the power switching tubes Q2 and Q3 is realized, the Q1 and the Q4 are continuously turned off, the input rectified voltage | vin | is applied to the parallel resonant inductor Lp and the primary winding of the main transformer T1 after passing through the resonant tank series resonant inductor Lr and the series resonant capacitor Cr, the amplitude of the midpoint voltage V34 is | vin |, the resonant current ir is approximate forward sine wave, the secondary winding of the power switching tubes is rectified by the diode D5, filtered by the output filter capacitor Co and then provided for the load RL。
During the period from t1 to t2, Q2 and Q3 start to turn off, Q1 and Q4 continue to turn off, and the magnitudes of drain and source voltages of Q1 and Q4 are equal to | vin |, which is also called the first dead zone. Lp current passes through the primary winding of Ci, Lr, Cr, T1 and the body diode DQ1And DQ4Freewheeling, where the midpoint voltage V34 begins to resonate down to zero and ir begins to fall, where no energy is provided to the output on the primary side and Co provides energy to the load RL. Note that the gate drive signals must be applied to Q1 and Q4 before the Lp current is ramped down to zero to achieve ZVS turn-on.
During the period (T2-T3), the Q1 and the Q4 realize ZVS turn-on, the Q2 and the Q3 are continuously turned off, the input rectified voltage | vin | is connected with the resonant inductor Lr and the series resonant capacitor Cr in series through the resonant tank and then is applied to the parallel resonant inductor Lp and the primary winding of the main transformer T1 in opposite phase, the midpoint voltage V34 is- | vin |, the resonant current ir is approximate reverse sine wave, the secondary winding of the resonant current ir is rectified through the D6 and is provided for the load R after being filtered by CoL。
During the period from t3 to t4, Q1 and Q4 start to turn off, Q2 and Q3 continue to turn off, the magnitudes of drain and source voltages of Q1 and Q4 are equal to | vin |, and this period is the second dead zone. Lp current passes through the primary winding of Ci, Lr, Cr, T1 and the body diode DQ2And DQ3Freewheeling, the midpoint voltage V34 begins to rise from zero at resonance and ir begins to fall in reverse, no energy is provided on the primary side to the output, and Co provides energy to the load RL. Note that the gate drive signals must be applied to Q2 and Q3 before the Lp current reverses to zero to achieve its ZVS turn-on.
From the above, it can be seen that the duty cycles of the four power switching tubes Q1-Q4 are all 0.5, and the illustration adopts frequency conversion control, and optionally uses phase-shift PWM control. In fact, the resonant frequency under the duty ratio setting and the variable frequency control condition is the same as the switching frequency of the switching tube.
When the ac input high-voltage section works in the half-bridge mode, the switching mode equivalent circuit is as shown in fig. 42 to 45, during which the power switch tube Q1 is always turned off and the power switch tube Q2 is always turned on, actually, the full-bridge circuit of fig. 37 is equivalently changed into the half-bridge circuit through a flexible structure, and the specific working process is as follows:
during the period from T0 to T1, the power switch tube Q3 has realized zero voltage turn-on (ZVS), Q4 is continuously turned off, the input rectified voltage | vin | is applied to the parallel resonant inductor Lp and the primary winding of the main transformer T1 after passing through the resonant tank series resonant inductor Lr and the series resonant capacitor Cr, the amplitude of the midpoint voltage V34 is | vin |, the resonant current ir is an approximate forward sine wave, and the secondary winding is rectified by the diode D5 and is filtered by the output filter capacitor Co and then provided to the load RL.
During the period (t 1-t 2), Q3 starts to turn off, Q4 continues to turn off, and the amplitude of the drain and source voltages of Q4 is equal to | vin |, which is also called the first dead zone. Lp current flows through a Ci, Lr, Cr and T1 primary winding and a body diode DQ4, the midpoint voltage V34 starts to resonate and decreases to zero, ir also starts to decrease, no energy is provided to the output from the primary winding, and energy is provided to a load RL by Co. Note that the gate drive signal must be applied to Q4 before the Lp current is ramped down to zero to achieve its ZVS turn-on.
During the period from T2 to T3, the ZVS is turned on by Q4, the Q3 is turned off continuously, the voltage at two ends of Cr is applied to primary windings of Lp and T1 in opposite phases through Lr (namely, the primary windings form the same LLC resonant circuit), the midpoint voltage V34 is zero, the resonant current ir is approximate reverse sine wave, and the secondary winding is rectified through D6, filtered by Co and provided for the load RL.
During the period from t3 to t4, Q4 starts to turn off, Q3 continues to turn off, and the amplitude of the drain and source voltages of Q4 is equal to | vin |, which is the second dead zone. Lp current flows through a Ci, Lr, Cr and T1 primary winding and a body diode DQ3, a midpoint voltage V34 starts to resonate and rises from zero, ir also starts to fall reversely, no energy is provided for the output on the primary winding, and energy is provided for a load RL by Co. Note that the gate drive signal must be applied to Q3 before the Lp current reverses to zero to achieve its ZVS turn-on.
From the above, it can be seen that the duty cycles of the four power switching tubes Q1-Q4 are all 0.5, and the illustration adopts frequency conversion control, and optionally uses phase-shift PWM control. In fact, the resonant frequency under the duty ratio setting and the variable frequency control condition is the same as the switching frequency of the switching tube. The above description is directed to a wide ac input voltage range, and in fact, the basic principles thereof are also applicable to a wide dc output voltage. For example, when high voltage needs to be output, the power conversion circuit works in a full-bridge mode; when the output low voltage is needed, the power conversion circuit works in a half-bridge mode, the basic principle is similar, and the description is not repeated here.
The resonant converter adopts a multilevel conversion circuit to adaptively switch a series-parallel stacking mode through a controller, and actually, a series-parallel stacking mode of two basic LLC units can also be used, and meanwhile, the functions of voltage transformation and isolation are considered. The power conversion circuit uses two LLC half-bridge resonant converters as an embodiment seventeen, as shown in fig. 46. Compared with the fig. 7, the number and the serial form of the four series power switching tubes are not changed, but two sets of independent resonant circuits, a main transformer, an output rectifying circuit, a high-frequency filter capacitor are respectively used, and a voltage section selection switch is additionally added. When the AC input voltage is lower, the selection switch K is closed, and when the AC input sine wave is in a positive half cycle, Vin is filtered by Lf and Cf of the EMI filter and charges Ci1 through D1; when an alternating current input sine wave is in a negative half cycle, Vin is filtered by Lf and Cf of the EMI filter and charges Ci2 through D3, therefore Ci1 and Ci2 can be charged to the peak value of the alternating current input voltage, the two do not form a series voltage division relation, a bridgeless PFC working mode is formed, and conduction loss of a traditional input rectifying circuit D2 and D4 is reduced. When the AC input voltage is higher, the selection switch K is switched off, and when the AC input sine wave is in positive and negative half cycles, Vin is filtered by Lf and Cf of the EMI filter and charges Ci1 and Ci2 through D1, D4, D2 and D3 respectively, so that the sum of the voltages at two ends of Ci1 and Ci2 is the AC input voltage peak value, and the two voltages form a series voltage division relationship. Optionally, the two LLC half-bridge resonant converters adopt an interleaved parallel technology to reduce output high-frequency ripple, and the output connection mode can be changed from parallel to series. Alternatively, the power conversion circuit may also use two LLC full-bridge resonant converters. Further optionally, the direct current output can be changed into a series relation, and only the output sampling and control mode needs to be adjusted. The lightning protection and hold-up time extension circuit shown in fig. 23 may be used, or the output low ripple circuit shown in fig. 24 to 26, the three-level topology shown in fig. 27 to 28, the output power increasing method shown in fig. 29, the three-phase ac input system shown in fig. 30, and the resonant converter shown in fig. 33 to 35 and 37 may be used.
Besides the isolated converter, the method can also be used in a non-isolated AC/DC resonant converter, and the embodiment is shown in FIG. 47. Compared with the figure 7, only the main transformer is omitted, and the parallel resonant inductor Lp still passes through the output rectifier diodes D5-D8 and the load RLAnd (4) connecting in parallel. It should be noted that the lightning protection and holding time extension circuit shown in fig. 23, the output low ripple circuits shown in fig. 24 to 26, the other types of three-level topologies shown in fig. 27 to 28, the output power increase mode shown in fig. 29, the three-phase ac input system shown in fig. 30, the higher level implementation mode shown in fig. 31, and the other types of resonant converters shown in fig. 33 to 37 may be used. The working principle of other parts is basically similar to that of figures 3-22, and the description is not repeated here.
An autotransformer may also be used for flexible adjustment of the voltage gain of the non-isolated AC/DC resonant converter, and a nineteenth embodiment is shown in fig. 48. Compared with fig. 47, the parallel resonant inductor is mainly changed to the self-coupling type, and the output uses voltage-doubling rectification to reduce the number of output rectifying diodes. It should be noted that the lightning protection and holding time extension circuit shown in fig. 23, the output low ripple circuits shown in fig. 24 to 26, the other types of three-level topologies shown in fig. 27 to 30, the output power increase mode shown in fig. 29, the three-phase ac input system shown in fig. 30, the higher level implementation mode shown in fig. 31, and the other types of resonant converters shown in fig. 33 to 37 may be used. The working principle of other parts is basically similar to that of the figures 2-22, and the description is not repeated here.
The invention provides an AC/DC resonance converter which comprises a power conversion circuit and a controller thereof, wherein the power conversion circuit is composed of basic switch units and a resonance tank, the controller adaptively controls the power conversion circuit according to different AC input voltages or different DC output voltages and switches the power conversion circuit into one or more basic switch units to be stacked in series and parallel for working, the power conversion circuit works in a high-frequency full-voltage control mode when AC input voltage or DC output high voltage is input, and works in a low-frequency partial-voltage control mode when AC input voltage or DC output low voltage is output, the two level control modes share the same resonance device to simplify the design of the resonance tank, a multiplier function unit is arranged in the controller, and AC input current tracks the frequency and the phase of the input voltage in real time to realize the power factor correction function. The resonant converter directly converts alternating current input instantaneous voltage into direct current output voltage, so that a conventional PFC converter and a direct current bus capacitor thereof can be saved, the conversion efficiency is improved, and the cost of components is reduced. The power tube is soft switch work under different operating conditions and can reduce turn-off loss, and low frequency partial pressure control mode has lower equivalent switching frequency simultaneously to greatly reduce its switching loss, further improve conversion efficiency, high integration controller design can further reduce the switching power supply volume in addition.
The invention can achieve higher conversion efficiency, lower component cost and smaller volume, and can be widely applied to various switching power supplies, such as chargers, power adapters, LED drives, charging piles, industrial and control power supplies and the like.
The foregoing is merely a preferred embodiment of the invention and is not intended to limit the invention in any manner; those skilled in the art can readily practice the invention as shown and described in the drawings and detailed description herein; however, those skilled in the art should appreciate that they can readily use the disclosed conception and specific embodiments as a basis for designing or modifying other structures for carrying out the same purposes of the present invention without departing from the scope of the invention as defined by the appended claims; meanwhile, any changes, modifications, and evolutions of the equivalent changes of the above embodiments according to the actual techniques of the present invention are still within the protection scope of the technical solution of the present invention.

Claims (10)

1. An AC/DC resonant converter characterized by: the power conversion circuit is connected with an input rectification circuit of a switching power supply, the output rectification filter circuit is connected with the power conversion circuit, the controller is connected with the output rectification filter circuit, the power conversion circuit and the input rectification circuit, the basic switch unit comprises a high-frequency filter capacitor, a plurality of power switch tubes and body diodes of the power switch tubes, the power switch tubes are connected in series, the high-frequency filter capacitor is connected with the power switch tube in series, the basic switch units form a series connection and parallel connection stacking mode, the controller comprises a high-voltage starting and internal power supply unit, an enabling and protecting unit, an output voltage current sampling and feedback unit, a light-load processing mode unit, The voltage-controlled oscillator and pulse modulator, the primary side current sampling and conditioning unit, the multiplier and main control logic unit, the grid signal enabling and driving unit and the AC input voltage sampling and processing unit are connected, the voltage-controlled oscillator and pulse modulator is connected with the grid signal enabling and driving unit, the enabling and protecting unit, the multiplier and the main control logic unit, and the primary side current sampling and conditioning unit, the output voltage current sampling unit, the light load processing mode unit and the feedback unit and the AC input voltage sampling and processing unit are connected with the multiplier and the main control logic unit.
2. An AC/DC resonant converter as recited in claim 1, wherein: the power conversion circuit is formed by stacking two basic switch units in series, the resonant tank comprises a first series resonant inductor, a parallel resonant inductor and a series resonant capacitor, the output rectifying circuit adopts a transformer center tap diode full-wave rectifying structure form and comprises a first rectifying diode, a second rectifying diode, a main transformer and an output filter capacitor, the anodes of the first rectifying diode and the second rectifying diode are respectively connected to two ends of a secondary winding of the main transformer, the cathodes of the first rectifying diode and the second rectifying diode are connected with the anode of the output filter capacitor, the cathode of the output filter capacitor is connected to the center tap of the main transformer, and two ends of an output load are respectively connected to the anode and the cathode of the output filter capacitor; the output midpoint of the first basic switch unit is connected to one end of a first series resonance inductor, the output midpoint of the second basic switch unit is connected to one end of a series resonance capacitor, the other ends of the first series resonance inductor and the series resonance capacitor are respectively connected to two ends of a parallel resonance inductor, and two ends of the parallel resonance inductor are connected with two ends of a primary winding of a main transformer in parallel; the controller comprises an output resistor, a voltage error amplifier, a primary and secondary side isolation optocouplers, a current source, a multiplier, an input resistor, a current error amplifier, a voltage-controlled oscillator, a pulse modulator, a grid signal enable and drive unit and a grid amplifier, wherein the output resistor is connected with the anode of the output filter capacitor, the sampled direct current output voltage of the output resistor is connected with the negative end of the voltage error amplifier in parallel, the positive end of the voltage error amplifier is connected with a voltage reference signal, the output end of the voltage error amplifier is connected to the cathode of a light-emitting diode in the primary and secondary side isolation optocouplers, the anode of the light-emitting diode is connected to the anode of the output filter capacitor through a resistor, the emitter of a triode in the primary and secondary side isolation optocouplers is connected with a primary ground, the collector is connected with the current source in the controller, is connected to the grid signal enable and drive unit, and is connected to one end of the multiplier for input, the input resistor is connected with the output end of the input rectifying circuit, the sampled and rectified voltage is connected to the other end of the multiplier to be input, the output end of the multiplier is connected to the positive end of the current error amplifier to serve as a current reference signal, the primary input current sampling signal is connected to the negative end of the current error amplifier, the output end of the current error amplifier is connected to the voltage-controlled oscillator to generate a sawtooth wave oscillation signal, the sawtooth wave oscillation signal enters the pulse modulator, and a pulse driving signal is generated and passes through the grid amplifier to drive the power switch tube of the basic switch unit.
3. An AC/DC resonant converter as recited in claim 2, wherein: the direct current power supply circuit also comprises a capacitance energy storage branch circuit, wherein the capacitance energy storage branch circuit is connected with the output positive electrode and the output negative electrode of the input rectification circuit in parallel, and the capacitance energy storage branch circuit is formed by connecting a first power switch tube, a body diode of the first power switch tube and a direct current capacitor in series;
the second-stage LC filter circuit is composed of a filter inductor and a second filter capacitor, the second filter capacitor is connected with the output filter capacitor in parallel, and the filter inductor Ld is connected between the anode of the output filter capacitor and the anode of the second filter capacitor;
the cascade step-down DC/DC converter is composed of a second power switch tube, a body diode of the second power switch tube, a main inductor, a freewheeling diode and a third filter capacitor, the freewheeling diode and the output filter capacitor are connected in parallel, the drain electrode of the second power switch tube is connected with the third filter capacitor and the negative electrode of the freewheeling diode, and the main inductor is connected between the negative electrode of the freewheeling diode and the positive electrode of the output filter capacitor.
4. An AC/DC resonant converter as recited in claim 3, wherein: the output rectification circuit comprises a main converter and an auxiliary converter, and is provided with a corresponding main controller and an auxiliary controller, wherein the main converter is provided with two auxiliary windings which are respectively a main winding and an auxiliary winding, the main winding of the main converter forms a main output after being rectified and filtered by the first rectifier diode, the second rectifier diode and the output filter capacitor, the auxiliary winding of the auxiliary converter forms a direct current bus voltage after being rectified and filtered by the third rectifier diode, the fourth rectifier diode and the third filter capacitor, and forms an auxiliary output after being rectified and filtered by the second power switch tube, the body diode of the second power switch tube, the main inductor, the fly-wheel diode and the third filter capacitor, and the main output and the auxiliary output are connected in series to obtain a total output voltage and provide the total output voltage for an output load.
5. An AC/DC resonant converter as recited in claim 2, wherein: the power conversion circuit uses a diode-clamped AC/DC three-level LLC half-bridge resonant converter, and further comprises a first diode and a second diode, the series midpoint of the high-frequency filter capacitor of the first basic switch unit and the high-frequency filter capacitor of the second basic switch unit is connected with the anode of the first diode, the cathode of the second diode and the series resonant capacitor, the cathode of the first diode is connected with the series midpoint of the power switch tube of the first basic switch unit, the anode of the second diode is connected with the series midpoint of the power switch tube of the second basic switch unit, and the series midpoint of the power switch tube of the first basic switch unit and the power switch tube of the second basic switch unit is connected with the first series resonant inductor.
6. An AC/DC resonant converter as recited in claim 5, wherein: the power conversion circuit uses a diode-clamped AC/DC three-level LLC full-bridge resonant converter, the power conversion circuit further comprises a third diode, a fourth diode, a third basic switch unit and a fourth basic switch unit, the series midpoint of a high-frequency filter capacitor of the first basic switch unit and a high-frequency filter capacitor of the second basic switch unit is connected with the anode of the third diode and the cathode of the fourth diode, the cathode of the third diode is connected with the series midpoint of a power switch tube of the third basic switch unit, the anode of the fourth diode is connected with the series midpoint of a power switch tube of the fourth basic switch unit, the series midpoint of the power switch tube of the first basic switch unit and the power switch tube of the second basic switch unit is connected with the series resonant capacitor, and the series midpoint of the power switch tube of the third basic switch unit and the power switch tube of the fourth basic switch unit is connected with the series resonant capacitor The midpoint is connected to the first series resonant inductor.
7. An AC/DC resonant converter as recited in claim 2, wherein: the number of the AC/DC resonant converters is two, and the inputs of the first set of LLC half-bridge resonant converter and the second set of LLC half-bridge resonant converter are connected in series or adopt a phase-staggered 180-degree control mode;
the number of the AC/DC resonant converters is three, and EMI filter inductors and filter capacitors of the A phase, the B phase and the C phase of three-phase alternating current are respectively connected with the AC/DC resonant converters through an input rectification circuit.
8. An AC/DC resonant converter as recited in claim 2, wherein: the power conversion circuit is formed by stacking four basic switch units in series, a power switch tube series branch of a third basic switch unit is connected between power switch tubes of the first basic switch unit, a power switch tube series branch of a fourth basic switch unit is connected between power switch tubes of the second basic switch unit, the output midpoint of the third basic switch unit is connected to one end of the first series resonant inductor, and the output midpoint of the fourth basic switch unit is connected to one end of the series resonant capacitor; the other ends of the first series resonance inductor and the series resonance capacitor are respectively connected to two ends of a parallel resonance inductor, and the parallel resonance inductor is connected with two ends of a primary winding of the main transformer in parallel;
the power conversion circuit is formed by stacking a basic switch unit in series, the resonant tank comprises a first series resonant inductor, a parallel resonant inductor and/or a parallel resonant capacitor and a series resonant capacitor, the middle point of the output of the basic switch unit is connected to the series resonant capacitor, the series resonant capacitor is connected with the first series resonant inductor, a power switch tube branch of the basic switch unit is connected with the parallel resonant inductor and/or the parallel resonant capacitor, and the parallel resonant inductor and/or the parallel resonant capacitor are connected with two ends of a primary winding of the main transformer in parallel;
the resonant tank further comprises a second series resonant inductor, and the second series resonant inductor is connected between the first series resonant inductor and the main transformer;
EMI filter inductors and filter capacitors of the three-phase alternating current phase A, the phase B and the phase C are connected with the basic switch unit through an input rectification circuit;
the power conversion circuit is formed by parallelly stacking two basic switch units, two power switch tube series branches are parallelly connected and parallelly connected with a high-frequency filter capacitor, the output midpoint of one power switch tube series branch is connected with a first series resonance inductor, and the output midpoint of the other power switch tube series branch is connected with a series resonance capacitor.
9. An AC/DC resonant converter as recited in claim 2, wherein: the number of the resonant tanks is two, a first series resonant inductor of a first resonant tank is connected with the output midpoint of the power switch tube series branch of a first basic switch unit, a series resonant capacitor of the first resonant tank is connected with the output midpoint of the power switch tube series branch of the first basic switch unit, a first series resonant inductor of a second resonant tank is connected with the output midpoint of the power switch tube series branch of a second basic switch unit, and a series resonant capacitor of the second resonant tank is connected with the power switch tube series branch of the second basic switch unit; the high-frequency filter circuit also comprises a voltage section selection switch which is connected between the input rectifying circuit and the series midpoint of the high-frequency filter capacitor of the first basic switch unit and the high-frequency filter capacitor of the second basic switch unit.
10. An AC/DC resonant converter as recited in claim 2, wherein: the output rectifying circuit comprises a first rectifying diode, a second rectifying diode, an output filter capacitor, a third rectifying diode and a fourth rectifying diode, the first rectifying diode, the second rectifying diode, the third rectifying diode and the fourth rectifying diode form a rectifying bridge, the input end of the rectifying bridge is connected with the parallel resonant inductor, and the output end of the rectifying bridge is connected with the output filter capacitor;
the anodes of the first rectifier diode and the second rectifier diode are respectively connected to the two ends of the parallel resonant inductor, the cathodes of the first rectifier diode and the second rectifier diode are connected with the anode of the output filter capacitor, the cathode of the output filter capacitor is connected to a center tap of the main transformer, and the two ends of an output load are respectively connected to the anode and the cathode of the output filter capacitor; the output midpoint of the first basic switch unit is connected to one end of a first series resonance inductor, the output midpoint of the second basic switch unit is connected to one end of a series resonance capacitor, the other ends of the first series resonance inductor and the series resonance capacitor are respectively connected to two ends of a parallel resonance inductor, and two ends of the parallel resonance inductor are connected with two ends of a primary winding of a main transformer in parallel;
the output rectifying circuit comprises an autotransformer, a first rectifying diode, a second rectifying diode, an output filter capacitor and a fourth filter capacitor, the first series resonant inductor and the series resonant capacitor are connected with the primary side of the autotransformer, the first rectifying diode and the output filter capacitor are connected to two ends of the secondary side of the autotransformer in parallel, the fourth filter capacitor is connected between the secondary side of the autotransformer and the negative electrode of the first rectifying diode, and the positive electrode of the second rectifying diode is connected between the negative electrode of the first rectifying diode and the positive electrode of the output filter capacitor.
CN202110662362.7A 2021-06-15 2021-06-15 AC/DC resonant converter Pending CN113595415A (en)

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CN115498889A (en) * 2022-10-17 2022-12-20 科威尔技术股份有限公司 Three-phase interleaved bidirectional resonant half-bridge direct-current converter system and control method
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CN115566907A (en) * 2022-11-11 2023-01-03 四川大学 Improved VMC LLC resonant PFC converter control system and design method thereof
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CN115811241A (en) * 2023-02-08 2023-03-17 四川大学 Single-stage bridgeless staggered parallel Boost-LLC AC-DC converter hybrid control method
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