CN113541477B - Boosting modular DC-DC converter for high-voltage direct-current power transmission system - Google Patents

Boosting modular DC-DC converter for high-voltage direct-current power transmission system Download PDF

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CN113541477B
CN113541477B CN202110823800.3A CN202110823800A CN113541477B CN 113541477 B CN113541477 B CN 113541477B CN 202110823800 A CN202110823800 A CN 202110823800A CN 113541477 B CN113541477 B CN 113541477B
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phase
converter
modular
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CN113541477A (en
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刘赫
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Shenzhen Polytechnic
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/36—Arrangements for transfer of electric power between ac networks via a high-tension dc link
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/483—Converters with outputs that each can have more than two voltages levels
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E60/00—Enabling technologies; Technologies with a potential or indirect contribution to GHG emissions mitigation
    • Y02E60/60—Arrangements for transfer of electric power between AC networks or generators via a high voltage DC link [HVCD]

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The invention discloses a boosting modular DC-DC converter for a high-voltage direct-current power transmission system, which is used as a direct-current acquisition point of the high-voltage direct-current power transmission system. The boost modular DC-DC converter of the invention adopts the most advanced MMC at both ends. And a special intermediate frequency decoupling transformer is adopted, so that the miniaturization design is realized. The converter design has the characteristics of modularization, expandability, redundancy, current isolation and the like, and is realized by adopting low-voltage current devices. Compared with two traditional three-phase MMC-based DC-DC converters, the converter achieves similar efficiency, and the number of SMs on the secondary side is reduced by 66%.

Description

Boosting modular DC-DC converter for high-voltage direct-current power transmission system
Technical Field
The invention relates to the technical field of transformers, in particular to a boosting modular DC-DC converter for a high-voltage direct-current power transmission system.
Background
High voltage direct current transmission system (HVDC) is a mature, proven technology for the long distance transmission of large-scale energy. As dc power transmission systems are increasingly used, interconnection between different levels of dc power transmission systems becomes increasingly challenging. The high voltage direct current acquisition network is a promising integrated technology, and aims to eliminate an additional conversion stage and improve the reliability of a system. At present, the key to realize the integration of a system and an HVDC is to construct a high-voltage high-power DC-DC converter by using a multi-module (consisting of a plurality of converter modules) and a modular multilevel topological structure. The high-voltage high-power DC-DC converter can be used as a DC acquisition point with different voltage levels and can also be used as a DC isolator to remove possible faults of one high-voltage direct-current transmission line; for this purpose, the prior art proposes the following topologies of the converter:
a Dual Active Bridge (DAB) converter has: electrical isolation, bi-directional power flow, and high switching frequency operation capability, however, the converter employs a single power module that is difficult to meet high voltage and high power requirements, requiring series and/or parallel combinations of power semiconductor devices and conversion modules.
Furthermore, an Input Parallel Output Series (IPOS) configuration is often preferred because the dc collection point needs to provide high voltage to facilitate connection to the hvdc transmission system. There have been many papers investigating this combined converter as a dc pick-up point for a hvdc system, however, for such converters, full soft switching operation can only be achieved within a limited range of load and input voltage variations. Furthermore, the efficiency and performance of the converter is greatly limited due to increased switching losses and electromagnetic interference. To address this problem, an extra large resonant inductor is usually connected with the transformer to extend the soft switching range, but the large inductor adversely affects the performance of the converter, because it results in increased duty cycle loss and severe ringing voltage. There are also researchers discussing the concept of replacing linear inductance with saturated inductance, which effectively extends the range of soft switching with lower conduction losses and no significant duty cycle losses. But the power density and large-scale application of the overall system is limited due to the large magnetic core required for heat dissipation.
On the other hand, HVDC system dc connection points based on MMC (modular multilevel converter) are advantageous due to its many aspects. The prior art proposes a series of transformerless DC-DC converters consisting of MMCs to adapt to different voltage applications, such as tuned filter modular DC converters and push-pull modular multilevel DC converters. However, these converters lack electrical isolation characteristics due to the absence of a transformer. In order to solve the problem, an MMC-based DC-DC converter is provided and is connected through an intermediate frequency transformer to serve as a direct current collection point of an HVDC system. These converters include a unidirectional converter and a bidirectional converter, but a unidirectional DC/DC converter based on MMC cannot control active and reactive power respectively because a diode is applied as a rectifier module on the secondary side.
From the above, it can be known that the application of the existing DC-DC converter to the high voltage direct current transmission system may cause the voltage stress of the half-bridge sub-module to increase, which may cause the high voltage direct current transmission system to be unstable and reliable, and the existing DC-DC converter needs to use more SM (power sub-module or power component), which may cause the high cost of the DC-DC converter.
Disclosure of Invention
Technical problem to be solved
The invention provides a boost modularized DC-DC converter for a high-voltage direct-current transmission system, which reduces the voltage stress of a half-bridge submodule in the high-voltage direct-current transmission system, improves the stability and reliability of the system and reduces the cost of the boost modularized DC-DC converter.
(II) technical scheme
In order to achieve the purpose, the invention provides the following technical scheme: a boost modular DC-DC converter for a high voltage direct current transmission system, comprising the steps of:
step S1: establishing a topological structure of a boosting modular DC-DC converter, wherein a three-phase MMC inverter generates controllable alternating voltage and is connected to the primary side of a three-phase decoupling intermediate frequency transformer, and each decoupling phase of the secondary side of the boosting modular DC-DC converter is connected with a sub-module of a single-phase MMC;
step S2: constructing a mathematical model of the boost modular DC-DC converter according to the primary side and the secondary side of the boost modular DC-DC converter:
Figure GDA0003704520220000031
step S3: calculating the output power of the boost modular DC-DC converter
Figure GDA0003704520220000032
And obtaining the output power
Figure GDA0003704520220000033
The characteristics of (a);
step S4: calculating the loss of the primary side three-phase MMC inverter of the boost modular DC-DC converter;
step S5: comparing the boost modular DC-DC converter with the conventional two three-phase MMC-based DC-DC converters according to the calculation of steps S3 and S4 results in that the number of SMs used by the boost modular DC-DC converter is less than the number of SMs used by the conventional two three-phase MMC-based DC-DC converters, and the boost modular DC-DC converter exhibits similar efficiency and total loss in a high output power range as the conventional two three-phase MMC-based DC-DC converters.
Preferably, the mathematical models of the a phase, the B phase and the C phase of the primary side of the boost modular DC-DC converter are:
phase A:
Figure GDA0003704520220000041
phase B:
Figure GDA0003704520220000042
and C phase:
Figure GDA0003704520220000043
mathematics of A phase, B phase and C phase at secondary side of boost modular DC-DC converter
The model is as follows:
phase A:
Figure GDA0003704520220000044
phase B:
Figure GDA0003704520220000045
and C phase:
Figure GDA0003704520220000046
wherein N is s The number of each corresponding single-phase MMC rectification module on the secondary side is equal to the number of each corresponding single-phase MMC rectification module on the secondary side;
if the total turn ratio of the primary and secondary coils of each phase is set as R t The turn ratio of primary coil to secondary coil of each phase is T r Then the equivalent voltage V of the secondary side sa 、V sb And V sc The feedback to the primary side can be expressed as:
phase A:
Figure GDA0003704520220000047
phase B:
Figure GDA0003704520220000048
and C phase:
Figure GDA0003704520220000049
will V a 、V b And V c Respectively substituting the voltage-boosting modular DC-DC converter with the mathematical models of the A phase, the B phase and the C phase at the primary side of the voltage-boosting modular DC-DC converter, and then converting the leakage inductance L of each phase of the transformer into the leakage inductance L k And feeding back to the primary side to obtain the mathematical model of the boost modular DC-DC converter.
Preferably, step S3 includes:
step S31: establishing a key voltage current waveform of the boost modular DC-DC converter:
the method comprises the following steps that firstly, an output waveform of a primary side of the boost modular DC-DC converter is assumed to be an approximately sinusoidal output waveform;
secondly, verifying the assumption of the first step through A phases of the primary side and the secondary side of the boost modular DC-DC converter;
thirdly, obtaining that the waveform of the key voltage and current of the boost modular DC-DC converter is a sine waveform and has symmetry;
step S32: calculating the output power of the boost modular DC-DC converter
In a first step, assume that the peak value of the primary equivalent AC voltage of phase A is equal to
Figure GDA0003704520220000051
I.e. ignoring any voltage drop over the circuit elements, the output power is derived only taking into account the key electrical voltage due to the symmetry of the key voltage current waveformHalf cycle of piezoelectric current waveform;
second, respectively calculating phase angles
Figure GDA0003704520220000052
Output power of A phase, B phase and C phase at different time intervals;
third, to the phase angle
Figure GDA0003704520220000053
Normalizing the output power of the A phase, the B phase and the C phase at different time intervals to obtain the total output power, namely the output power of the boost modular DC-DC converter
Figure GDA0003704520220000054
Preferably, the output power of the boost modular DC-DC converter
Figure GDA0003704520220000055
The characteristics of (A): when in use
Figure GDA0003704520220000056
When power is transferred from the high-voltage side to the low-voltage side, when
Figure GDA0003704520220000057
Figure GDA0003704520220000058
At this time, power is transferred from the low-voltage side to the high-voltage side.
(III) advantageous effects
Compared with the prior art, the invention has the beneficial effects that: the invention discloses a boosting modular DC-DC converter as a DC acquisition point of a high-voltage DC transmission system. The boost modular DC-DC converter of the invention adopts the most advanced MMC at both ends. And a special intermediate frequency decoupling transformer is adopted, so that the miniaturization design is realized. The converter design has the characteristics of modularization, expandability, redundancy, current isolation and the like, and is realized by adopting low-voltage current devices. Compared with two traditional three-phase MMC-based DC-DC converters, the converter achieves similar efficiency, and the number of SMs on the secondary side is reduced by 66%.
Drawings
The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention without limiting the invention in which:
FIG. 1 shows a topological structure of a boost modular DC-DC converter and a sub-module of a secondary side of the converter based on a single-phase MMC according to an embodiment of the present invention;
FIG. 2 illustrates a single-phase MMC schematic diagram and its equivalent circuit of an embodiment of the present invention;
FIG. 3 shows an equivalent circuit of a boost modular DC-DC converter of an embodiment of the present invention fed back to a primary side, which includes an A-phase equivalent circuit, a B-phase equivalent circuit, and a C-phase equivalent circuit;
FIG. 4 illustrates a key voltage current waveform diagram for phase A of a boost modular DC-DC converter of an embodiment of the present invention;
FIG. 5 illustrates a normalized output power versus phase angle for a boost modular DC-DC converter of an embodiment of the present invention;
FIG. 6 shows a circuit schematic of the MMC sub-module (SM) of an embodiment of the present invention;
FIG. 7 shows a power loss profile of a converter at a rated power of 12MW according to an embodiment of the present invention;
FIG. 8 is a graph showing a comparison of converter efficiency at different power ratings according to an embodiment of the present invention;
fig. 9 shows a schematic diagram of a d-q vector-based control method of the primary-side three-phase MMC inverter according to an embodiment of the present invention.
FIG. 10 illustrates an MMC-based secondary single-phase rectifier control block diagram of an embodiment of the present invention;
fig. 11 shows a three-phase MMC primary-side output waveform of an embodiment of the present invention, where (a) is a voltage waveform and (b) is a current waveform;
fig. 12 shows secondary side output waveforms of a three-phase MMC according to an embodiment of the present invention, wherein (a) are voltage waveforms of a-phases SM1 and SM2, (B) are voltage waveforms of B-phases SM1 and SM2, and (C) are voltage waveforms of C-phases SM1 and SM 2;
FIG. 13 shows a primary side phase A sub-module voltage diagram (a) and a secondary decoupled phase A sub-module voltage diagram (b) of an embodiment of the invention;
fig. 14 shows output waveforms of the boost modular DC-DC converter of the embodiment of the present invention in a steady state, where (a) is an output DC voltage diagram and (b) is an output DC current diagram;
fig. 15 shows a controller load step change of an embodiment of the present invention, in which (a) is an output dc voltage diagram and (b) is an output dc current diagram.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
Referring to fig. 1-15, the present invention discloses a boost modular DC-DC converter for a high voltage DC power transmission system, comprising the steps of:
step S1: and establishing a topological structure of the boost modular DC-DC converter, wherein a three-phase MMC inverter generates controllable alternating voltage and is connected to the primary side of a three-phase decoupling intermediate frequency transformer, and each decoupling phase of the secondary side of the boost modular DC-DC converter is connected with a sub-module of a single-phase MMC.
Specifically, fig. 1 shows a simplified schematic diagram of a boost modular DC-DC converter in which a three-phase MMC inverter generates a controllable ac voltage connected across the primary side of a three-phase decoupling intermediate frequency transformer. The secondary output voltage is decoupled into three identical 120 degree phase shifted voltages. In addition, each decoupled phase is further split into multiple windings to maximize the output dc voltage. It is noted that the design is fully modular, allowing for easy expansion of the stack of additional modules as needed. Thereby meeting different application occasions.
Step S2: constructing a mathematical model of the boost modular DC-DC converter according to the primary side and the secondary side of the boost modular DC-DC converter:
specifically, firstly, a mathematical model of the primary side of the boost modular DC-DC converter is constructed: FIG. 2 shows an equivalent circuit of A phase of a three-phase MMC inverter, wherein V dc_in And I dc_in The converter dc input voltage and current respectively. V ap And V aN The upper arm voltage and the lower arm voltage of the A-phase cascade submodule are respectively. I.C. A aP And I aN The currents of the upper and lower arms, respectively. E a Is the equivalent output phase voltage, V, shown in FIG. 2 a To output an alternating voltage. I is cir And I a Respectively, circulating current and alternating output current.
As can be seen from fig. 2, the upper arm and lower arm currents of phase a can be represented as:
I aP ＝I a /2+I cir (1)
I aN ＝-I a /2+I cir (2)
here, the current I circulates cir Flows through the upper and lower arms.
It should be noted that the circulating current has no effect on the output phase current and can be expressed as:
I cir ＝(I aP +I aN )/2 (3)
according to (1) and (2), the ac output current Ia can be represented by the upper and lower arm currents as:
I a ＝I aP -I aN (4)
as can be seen from fig. 2, considering that n is a neutral point, applying Kirchhoff's Voltage Law (KVL) can obtain that the upper and lower bridge arm voltages are:
Figure GDA0003704520220000091
Figure GDA0003704520220000092
combining (5) and equation (6), outputting a phase voltage V a Can be expressed as:
Figure GDA0003704520220000093
substituting (4) into (7) to equivalently output the phase voltage E a Comprises the following steps:
Figure GDA0003704520220000094
therefore, the mathematical model for rearranging (8) to obtain a single-phase MMC is:
Figure GDA0003704520220000095
according to (9), the equivalent circuit of phase a is shown in fig. 2.
Similarly, the mathematical models of the primary side B phase and the primary side C phase of the boost modular DC-DC converter are as follows:
Figure GDA0003704520220000101
Figure GDA0003704520220000102
in the formula, E b And E c Equivalent output voltages of B phase and C phase, V b And V c The AC output voltages of the B phase and the C phase are respectively. I is b And I c The alternating output currents of the B phase and the C phase are respectively. It is noted that each bridge arm has the same inductance, and is set to L arm 。
Then, mathematical models of A phase, B phase and C phase of the secondary side of the boost modular DC-DC converter are constructed, and according to the diagram shown in figure 1, a secondary side single-bridge arm of the transformerAn MMC can be viewed as a series of voltage sources (
Figure GDA0003704520220000103
Corresponding to the phase A, the phase A is divided into a phase A,
Figure GDA0003704520220000104
corresponding to the phase B,
Figure GDA0003704520220000105
corresponding to phase C) therefore, the equivalent voltage V of each phase of the secondary side sa , V sb And V sc The mathematical expression of (c) is as follows:
phase A:
Figure GDA0003704520220000106
phase B:
Figure GDA0003704520220000107
and C phase:
Figure GDA0003704520220000108
wherein N is s The number of each corresponding single-phase MMC rectifying module on the secondary side is shown.
If the total turn ratio of the primary and secondary coils of each phase is set as R t The turn ratio of primary coil to secondary coil of each phase is T r Then the equivalent voltage V of the secondary side sa ,V sb And V sc The feedback to the primary side can be expressed as:
phase A:
Figure GDA0003704520220000109
phase B:
Figure GDA00037045202200001010
and C phase:
Figure GDA00037045202200001011
equations (15), (16) and (17) are respectively substituted into equations (9), (10) and (11), and then transformer leakage inductance L is changed per phase k Fed back to the primary side, the equivalent mathematical model of the boost modular DC-DC converter of the present invention can be represented by the following equation:
Figure GDA0003704520220000111
step S3: calculating output power of boost modular DC-DC converter
Figure GDA0003704520220000112
And obtaining output power
Figure GDA0003704520220000113
The characteristic of (c).
Specifically, step S3 includes:
step S31: establishing a key voltage and current waveform of the boost modular DC-DC converter:
the method comprises the following steps that firstly, the output waveform of a primary side of a boost modular DC-DC converter is assumed to be an approximately sinusoidal output waveform;
secondly, verifying the assumption of the first step through A phases of the primary side and the secondary side of the boost modular DC-DC converter;
and thirdly, obtaining that the waveform of the key voltage and current of the boost modular DC-DC converter is a sine waveform and has symmetry.
Specifically, for simplicity of analysis, the output voltage waveform of the primary-side MMC is assumed to have the following characteristics: 1) the voltage balance of the sub-module capacitors is good and ripple free. 2) The converter operates with a uniform modulation index. 3) The MMC consists of a large number of sub-modules to produce an approximately sinusoidal ac output waveform.
Based on the above assumptions, taking phase a as an example, ideal primary and secondary reference voltage waveforms for converter phase a as shown in fig. 4 can be obtained, with power transferred from the low voltage side to the high voltage. As above, V sa Is the equivalent A-phase voltage of the secondary side, equal to
Figure GDA0003704520220000114
While
Figure GDA0003704520220000115
Is a primary side equivalent voltage E of phase A a And the equivalent voltage V of the secondary side feedback to the primary side sa The phase angle between.
Step S32: calculating output power of boost modular DC-DC converter
In a first step, assume that the peak value of the primary equivalent AC voltage of phase A is equal to
Figure GDA0003704520220000121
(i.e., ignoring any voltage drop across the circuit element), the output power is derived taking into account only half cycles of the critical voltage current waveform due to symmetry of the critical voltage current waveform;
second, respectively calculating phase angles
Figure GDA0003704520220000122
Output power of A phase, B phase and C phase at different time intervals;
third, to the phase angle
Figure GDA0003704520220000123
Normalizing the output power of the A phase, the B phase and the C phase at different time intervals to obtain the total output power, namely the output power of the boost modular DC-DC converter
Figure GDA0003704520220000124
Specifically, consider the assumptions that 1) the derivation of output power only takes into account the half-cycle of the AC waveform due to the symmetry of the waveform; 2) for simplicity, the peak value of the primary equivalent AC voltage of phase A is equal to
Figure GDA0003704520220000125
(i.e., ignoring any voltage drop across the circuit element).
As shown in the attached figure 1, the A phase, the B phase and the C phase decoupled from the secondary side of the transformer are connected in series,each phase can withstand one third of the output voltage
Figure GDA0003704520220000126
Therefore, the second order equivalent AC voltage peak value of the A phase is,
Figure GDA0003704520220000127
from the above analysis, the output power can be derived as follows:
interval 1:
Figure GDA0003704520220000128
as can be seen from FIG. 4, during this time interval, E a (θ)，V sa (θ)，I a (θ)
Can be expressed as:
Figure GDA0003704520220000129
Figure GDA00037045202200001210
Figure GDA00037045202200001211
substituting (19) and (20) into (21),
Figure GDA0003704520220000131
from (22) on
Figure GDA0003704520220000132
At that moment, one can get:
Figure GDA0003704520220000133
as can be seen from (21) and (23), the output energy of this period is:
Figure GDA0003704520220000134
interval 2:
Figure GDA0003704520220000135
during this time period, E a (θ)，V sa (θ)，I a (θ), which can be expressed as:
Figure GDA0003704520220000136
Figure GDA0003704520220000137
Figure GDA0003704520220000138
substituting (23), (25) and (26) into (27),
Figure GDA0003704520220000139
from (28), at the time θ ═ pi, one can obtain:
Figure GDA00037045202200001310
similarly, the energy transferred during this time period is:
Figure GDA00037045202200001311
Figure GDA0003704520220000141
therefore, the output power of the decoupled a phase of the boost modular DC-DC converter can be calculated from (24) and (29) as:
Figure GDA0003704520220000142
due to semi-circumferential symmetry, I a (0)＝-I a (π), therefore, from (29), the initial current can be calculated as I a (0)
Figure GDA0003704520220000143
Substituting (32) into (31) to make
Figure GDA0003704520220000144
Output power of boost modular DC-DC converter as phase angle
Figure GDA0003704520220000145
The function of (d) is:
Figure GDA0003704520220000146
where G is defined as the primary side DC voltage gain of the boost modular DC-DC converter, referred to as the DC conversion ratio. It should be noted that equation (33) is derived based on the assumption that the modulation index M is 1, which corresponds to the maximum power transmission capability of the boost modular DC-DC converter. For simplicity, the output power is normalized to the base number
Figure GDA0003704520220000147
The result is:
Figure GDA0003704520220000148
similarly, the output power of the B phase and the C phase can be obtained by:
Figure GDA0003704520220000149
Figure GDA00037045202200001410
thus, the total output power can be calculated as:
Figure GDA00037045202200001411
from equation (37), the normalized total output power versus phase angle of equation (37) can be obtained
Figure GDA0003704520220000151
As shown in fig. 5. It is obvious that
Figure GDA0003704520220000152
When power is transferred from the high-voltage side to the low-voltage side, when
Figure GDA0003704520220000153
At this time, power is transferred from the low-voltage side to the high-voltage side. Power transmission capability of boost modular DC-DC converter is subject to primary reference DC voltage gain G and phase angle
Figure GDA0003704520220000154
The influence of (c). It is apparent from fig. 5 that the maximum output power occurs
Figure GDA0003704520220000155
To (3).
Step S4: and calculating the loss of the primary side three-phase MMC inverter of the boost modular DC-DC converter.
In order to determine the feasibility of the boost modular DC-DC converter proposed by the present invention, its losses were analyzed and calculated to demonstrate its superiority in the boost architecture. To simplify the analysis, the present invention considers the following points: 1) only the losses of the main MMC inverter in the inversion mode are calculated, but it should be noted that the losses of the converter are calculated identically in both operating modes (i.e. rectification/inversion mode). 2) In the power loss calculation, only the fundamental component of the three-phase MMC inverter alternating voltage is considered.
It is noted that the present invention employs carrier phase shift pulse width modulation (CPS-PWM) to control the boost modular DC-DC converter. It is therefore possible to calculate the duty cycle as tau,
Figure GDA0003704520220000156
where M is the modulation index and ω is the fundamental ac frequency of the system.
Table one gives the operation mode of the MMC Submodule (SM).
TABLE-MMC submodule State
Figure GDA0003704520220000157
Figure GDA0003704520220000161
Taking the phase A submodule as an example: 1) when the modulation signal is greater than the carrier signal, T 2 Open, T 1 Close, SM bypass; 2) conversely, when the modulated signal is less than the carrier signal, T 2 Is turned off, T 1 Is turned on, which means that the SM is inserted into the circuit. Therefore, if the carrier period is set to T c The duty ratio τ (T) derived from equation (38) is such that when the SM is bypassed, the bypass time is τ (T) × T c I.e. T 2 The on-time is tau (T) x T c . On the contrary, when inserting SM, the insertion time is [ 1-tau (t)]×T c I.e. T 1 The conduction time is [ 1-tau (t)]×T c . Therefore, according to the above analysis, by the upper and lower IGBT modules (S) 1 ,S 2 ) Are respectively expressed as:
I sm1 ＝[1-τ(t)]*I aP_normal (t) (39)
I sm2 ＝τ(t)*I aP_normal (t) (40)
wherein I aP_normal (t) represents the upper arm current of the A phase.
According to the working principle of MMC in normal state, the input direct current I of each phase dc_in The AC output currents are uniformly distributed between the three phases, respectively, and also between the upper and lower arms, respectively. The upper arm current for phase a can therefore be expressed as:
Figure GDA0003704520220000162
substituting (38) and (41) into (39) and (40) yields:
Figure GDA0003704520220000163
Figure GDA0003704520220000164
from the above analysis, the effective value (RMS) and the average value of the current flowing through the switch and the diode can be obtained as follows (note that the following analysis is based on the current defined in fig. 6):
through diode D 1 The average current of (d) is:
Figure GDA0003704520220000171
substituting (42) into (44) to obtain:
Figure GDA0003704520220000172
through diodeD 1 The square of the effective current of (c) is:
Figure GDA0003704520220000173
substituting (42) into (46) to obtain:
Figure GDA0003704520220000174
by means of a switch T 1 The average current of (d) is:
Figure GDA0003704520220000175
substituting (42) into (48) to obtain:
Figure GDA0003704520220000176
through switch T 1 Has an effective current squared of
Figure GDA0003704520220000177
Substituting (42) into (50) to obtain:
Figure GDA0003704520220000178
Figure GDA0003704520220000181
by means of a switch T 2 The average current of (d) is:
Figure GDA0003704520220000182
substituting (43) into (52) to obtain:
Figure GDA0003704520220000183
by means of a switch T 2 The square of the effective current of (c) is:
Figure GDA0003704520220000184
substituting (43) into (54) to obtain:
Figure GDA0003704520220000185
through diode D 2 The average current of (d) is:
Figure GDA0003704520220000186
substituting (43) into (56) to obtain:
Figure GDA0003704520220000187
through diode D 2 The square of the effective current of (c) is:
Figure GDA0003704520220000188
substituting (43) into (58) to obtain:
Figure GDA0003704520220000189
Figure GDA0003704520220000191
the average/effective current derived above is used to calculate the conduction loss over a fundamental ac cycle, as follows:
Figure GDA0003704520220000192
Figure GDA0003704520220000193
in the formula P T_con And P D_con Respectively the conduction losses of the switch and the diode of the SM during one basic ac cycle. V D_0 And V T_0 r D_0 The threshold voltages of the diode and the switch, respectively. r is D_0 And r T_0 Respectively diode forward and switch forward resistors.
As known from the prior art, the switching loss is approximately linearly related to the average current flowing through the switching tube; therefore, the switching loss in a fundamental ac cycle is calculated by the formula:
Figure GDA0003704520220000194
Figure GDA0003704520220000195
where f is the switching frequency, E on And E off Respectively turn-on and turn-off losses. V T_ref And I T_avg Respectively, the reference voltage and current of the switch. E rec Is the reverse recovery energy loss of the diode. V D_ref And I D_avg Respectively, the reference voltage and current of the diode. It is to be noted that E on ， E off ，V T_ref ，I T_avg ，E rec ，V D_ref ，I D_avg The value of (d) can be obtained from a data table of the selected component.
Further, the loss of the intermediate frequency transformer of the boost modular DC-DC converter is evaluated:
generally, a transformerThere are two types of losses in the machine, copper losses and iron core losses. Copper losses are mainly caused by conductor resistivity and skin and proximity effects, and these losses increase with frequency. However, the use of Litz wire greatly reduces these effects, so the skin and proximity effects are not considered for simplicity of analysis. Therefore, P is lost in copper copper Is defined as:
Figure GDA0003704520220000201
wherein R is the equivalent resistance of each winding of the transformer, I rms Is the effective current through each winding.
Core loss is proportionally affected by the maximum magnetic flux density, which is an important factor in designing transformers. However, for a given magnetic flux, the flux density is determined only by the cross-sectional area of the core. In general, iron loss is predicted using Steinmetz equation [22], which can be expressed as:
Figure GDA0003704520220000202
wherein P is core And V core Core loss and core volume, respectively; b is pk Is the peak core flux density, the coefficients K, α, β are given by the properties of the core material. f is the ac frequency of the system.
Step S5: comparing the boost modular DC-DC converter with the conventional two three-phase MMC-based DC-DC converters according to the calculation of steps S3 and S4 results in that the number of SMs used by the boost modular DC-DC converter is less than the number of SMs used by the conventional two three-phase MMC-based DC-DC converters, and the boost modular DC-DC converter exhibits similar efficiency and total loss in a high output power range as the conventional two three-phase MMC-based DC-DC converters.
Specifically, referring to table two, table two illustrates the parameters of the boost modular DC-DC converter and the conventional two-phase MMC based DC-DC converter. Since both configurations employ exactly the same three-phase MMC inverter on the primary side (primary side), they have the same number of SMs, the same device rating, and the same switching and conduction losses, as shown in table two. In the invention, each decoupling phase (a phase, B phase and C phase) on the secondary side (secondary side) has two series connected rectifier modules based on single-tube MMC. Each leg of the single-phase MMC rectifier module contains five SMs. Thus, the total number of secondary side SMs is equal to 60, each SM having the capability of withstanding a voltage of 5 kV. According to similar design criteria, the conventional two three-phase MMC based DC-DC converters require 30SM per arm, which means that the total number of SMs required on the secondary side is 180 SM, while the boost modular DC-DC converter of the present invention is 60 SM. Thus, the boost modular DC-DC converter of the present invention can reduce the required number of SMs by 66% compared to two conventional three-phase MMC based DC-DC converters with the same design requirements. This greatly reduces cost and control complexity, resulting in a more reliable and cost-effective solution.
TABLE two different DC-DC CONVERTER PARAMETERS
Figure GDA0003704520220000211
Figure GDA0003704520220000221
For the sake of completeness, the losses of both topologies are calculated from the analysis described above to evaluate the efficiency of both converters in the output power range. For this purpose, FZ1200R12HE4 IGBT modules are used as primary side MMC for both converters. The FZ250R65KE3 [24] IGBT module was evaluated as the secondary side MMC for both converters. The decoupling three-phase transformer used by the boost modular DC-DC converter is constructed by three single-phase transformers, the primary windings of the three single-phase transformers are connected in a star shape, and the secondary windings of the three single-phase transformers are connected in a delta shape. Thus, only one third of the total power flows through each transformer. The loss of each single-phase transformer is calculated according to the parameters listed in table three, which are publicly reported in the prior art, and the invention is not explained in verification. From the calculation results, the boost modular DC-DC converter of the present invention and the conventional two three-phase MMC based DC-DC converters show similar efficiency and total loss, especially for the range of high output power.
Parameters of table three single-phase transformer
Figure GDA0003704520220000222
Figure GDA0003704520220000231
Based on the scheme, as shown in fig. 1, the boost modular DC-DC converter of the invention is composed of a primary-side three-phase MMC inverter and a series of single-phase MMC rectifier modules connected to the secondary side. The three-phase MMC inverter is used as a voltage source to generate alternating voltage with constant amplitude and frequency, and the secondary side single-phase MMC rectifying module controls the output power of the converter.
FIG. 9 is a schematic diagram of a control method of a primary side three-phase MMC inverter based on a d-q vector. In the formula I iabc And V iabc Primary three-phase alternating current and voltage respectively; i is idq And V idq Components of the three-phase alternating current and voltage transferred to the d-axis and q-axis, respectively. f. of i And L i The alternating current frequency and the bridge arm inductance of the main MMC inverter are respectively. The d-q transformation matrix of the primary side alternating current signal from the static coordinate system to the rotating coordinate system is as follows:
Figure GDA0003704520220000232
as can be seen from fig. 9, the control system comprises two control loops, a voltage outer loop and a current inner loop. The ac voltage is fed back from the primary side of the transformer and compared to its reference value. The resulting ac voltage error is then used as an input to an outer loop proportional-integral controller. PI outer loop controlThe output of the controller is used as a reference signal of the inner loop current controller. In the control method, V is set id_ref ) Set as the rated peak value of the AC voltage, V iq_ref Set to zero and set the AC frequency to f i . MMC-based single-pin secondary side rectifier control system:
fig. 10 is a control block diagram of a secondary single-phase rectifier based on MMC. In the formula I ra And V ra Are respectively single-phase MMC 1 Actual ac voltage and current of the module; i is rab And V rab Is a current and voltage quadrature pair. L is i Is a single leg arm inductor (note that all single leg MMC modules of a second have the same arm inductor). MMC 1 The d-q transformation matrix of the master control system from the stationary coordinate system to the rotating coordinate system is:
Figure GDA0003704520220000241
as can be seen from fig. 10, the single-leg MMC module is controlled using a similar d-q vector control method. q-axis current I rq It can simply be set to zero, which means that unity power factor can be achieved. The whole control system is realized by two control loops. In decoupled A-phase, an external voltage loop is used to regulate a single-leg MMC module MMC 1 Output DC voltage, the internal current loop being used to regulate MMC 1 The input alternating current of (2). The control signals of other modules can be directly transmitted from a module MMC 1 And (4) obtaining. Similarly, the control signals for the B-phase and C-phase single-phase modules can be derived from the A-phase MMC 1 The module acquires, but phase shifts 120 ° and 240 °, respectively.
To verify the theoretical analysis and effectiveness of the boost modular DC-DC converter disclosed in the present invention, the present example uses MATLAB/SIMULINK software to develop a simulation model with a rating of 12 MW/150 kV, and the parameters are listed in table four. In this operation, each phase of the primary-side MMC converter consists of five half-bridges SM on each arm. On the secondary side, each decoupled phase has two single-arm MMC rectifier modules in series, each having five half-bridges SM per arm.
Parameters of the Table four simulation System
Figure GDA0003704520220000251
As shown in fig. 11, the primary-side MMC generates an ac voltage of 400 Hz. The decoupling voltage on the secondary side is shown in fig. 12, where two identical voltages are obtained from each input phase. It should also be noted that each voltage pair is offset by 120 deg., as shown in fig. 12. It is worth mentioning that if a higher voltage is required at the output side, it may be considered to use a higher number of windings (with more single-leg MMC rectifier modules) on each decoupled secondary phase.
The performance of the proposed controller is also confirmed and proven in fig. 13 (i.e. phase a as an example). As shown in fig. 13(a), the capacitor voltage of the SM is strictly controlled around its reference value (i.e., the capacitor voltage of the upper arm SM). Similarly, the same voltage balance control method is applied to the single-leg MMC module of the secondary side. For example, fig. 13(b) shows controlling the upper arm SMs voltage of MMC _1 of decoupled a-phase at 150kV/6 x 5 (i.e., where 6 is the number of single-leg MMC rectifier modules and 5 is the number of SMs) per arm of each single-leg MMC rectifier module).
The total direct current output voltage and current of the boost modular DC-DC converter disclosed by the invention are shown in figure 14, and under the condition of adopting the design, the output voltage is kept at 150 kV.
Fig. 15 shows the dynamic response of the controller when the load is changed at t-0.5 s, and it can be seen that the output current is reduced from 80A to 60A, while the output voltage is kept constant, which demonstrates the effectiveness of the proposed method.
Based on the above, the invention discloses a boost modularized DC-DC converter as a direct current acquisition point of a high-voltage direct current transmission system. The boost modular DC-DC converter of the invention adopts the most advanced MMC at both ends. And a special intermediate frequency decoupling transformer is adopted, so that the miniaturization design is realized. The converter design has the characteristics of modularization, expandability, redundancy, current isolation and the like, and is realized by adopting low-voltage current devices. Compared with two traditional three-phase MMC-based DC-DC converters, the converter achieves similar efficiency, and the number of SMs on the secondary side is reduced by 66%.
It should be noted that although embodiments of the present invention have been shown and described, it would be appreciated by those skilled in the art that changes, modifications, substitutions and alterations can be made in these embodiments without departing from the principles and spirit of the invention, the scope of which is defined in the appended claims and their equivalents.

Claims (2)

1. A boost modular DC-DC converter for a high voltage direct current transmission system, comprising the steps of:
step S1: establishing a topological structure of a boost modular DC-DC converter, wherein the boost modular DC-DC converter generates controllable alternating voltage and is connected to the primary side of a three-phase decoupling intermediate frequency transformer, and each decoupling phase at the secondary side of the boost modular DC-DC converter is connected with a submodule of a single-phase MMC;
step S2: constructing a mathematical model of the boost modular DC-DC converter according to the primary side and the secondary side of the boost modular DC-DC converter:
Figure FDA0003711671430000011
wherein L is arm For the inductance of each leg, L k For per phase transformer leakage inductance, I a 、I b And I c For each phase of AC output current, E a 、E b And E c For each equivalent output voltage, V sa ，V sb And V sc Is the equivalent voltage of each phase of the secondary side, R t The total turn ratio of each phase of the primary side coil and the secondary side coil is obtained;
step S3: calculating the output power of the boost modular DC-DC converter
Figure FDA0003711671430000012
And obtaining the output power
Figure FDA0003711671430000013
The characteristics of (a);
step S31: establishing a key voltage current waveform of the boost modular DC-DC converter:
the method comprises the following steps that firstly, an output waveform of a primary side of the boost modular DC-DC converter is assumed to be an approximately sinusoidal output waveform;
secondly, verifying the assumption of the first step through A phases of the primary side and the secondary side of the boost modular DC-DC converter;
thirdly, obtaining that the waveform of the key voltage and current of the boost modular DC-DC converter is a sine waveform and has symmetry;
step S32: calculating the output power of the boost modular DC-DC converter:
in a first step, assume that the peak value of the primary equivalent AC voltage of phase A is equal to
Figure FDA0003711671430000021
I.e. ignoring any voltage drop over the circuit elements, the output power is derived considering only half cycles of the critical voltage current waveform due to the symmetry of the critical voltage current waveform, where V dc_in A DC input voltage for the converter;
second, respectively calculating phase angles
Figure FDA0003711671430000022
Output power of A phase, B phase and C phase at different time intervals;
third, to the phase angle
Figure FDA0003711671430000023
Normalizing the output power of the A phase, the B phase and the C phase at different time intervals to obtain the total output power, namely the output power of the boost modular DC-DC converter
Figure FDA0003711671430000024
Step S4: calculating the loss of the primary side three-phase MMC inverter of the boost modular DC-DC converter;
output power of the boost modular DC-DC converter
Figure FDA0003711671430000025
The characteristics of (A): when in use
Figure FDA0003711671430000026
When power is transferred from the high-voltage side to the low-voltage side, when
Figure FDA0003711671430000027
When the power is transferred from the low-voltage side to the high-voltage side;
step S5: comparing the boost modular DC-DC converter with the conventional two three-phase MMC-based DC-DC converters according to the calculation of steps S3 and S4 results in that the number of SMs used by the boost modular DC-DC converter is less than the number of SM used by the conventional two three-phase MMC-based DC-DC converters, and the boost modular DC-DC converter exhibits similar efficiency and total loss in a high output power range as the conventional two three-phase MMC-based DC-DC converters.
2. A boost modular DC-DC converter for an HVDC transmission system according to claim 1,
the mathematical models of the A phase, the B phase and the C phase of the primary side of the boost modular DC-DC converter are as follows:
phase A:
Figure FDA0003711671430000031
phase B:
Figure FDA0003711671430000032
and C phase:
Figure FDA0003711671430000033
the mathematical models of the A phase, the B phase and the C phase of the secondary side of the boost modular DC-DC converter are as follows:
phase A:
Figure FDA0003711671430000034
phase B:
Figure FDA0003711671430000035
and C phase:
Figure FDA0003711671430000036
wherein N is s The number V of each corresponding single-phase MMC rectification module at the secondary side sa ，V sb And V sc Is the equivalent voltage of each phase of the secondary side, V a1 ，V a2 ，...，
Figure FDA0003711671430000037
Is a phase A voltage source, V b1 ，V b2 ，...，
Figure FDA0003711671430000038
Is a B-phase voltage source, V c1 ，V c2 ，...，
Figure FDA0003711671430000039
Is a C-phase voltage source;
if the total turn ratio of the primary and secondary coils of each phase is set as R t The turn ratio of primary coil to secondary coil of each phase is T r Then the equivalent voltage V of the secondary side sa 、V sb And V sc The feedback to the primary side can be expressed as:
phase A:
Figure FDA00037116714300000310
phase B:
Figure FDA00037116714300000311
and C phase:
Figure FDA00037116714300000312
will V a 、V b And V c Respectively substituting the voltage-boosting modular DC-DC converter with the mathematical models of the A phase, the B phase and the C phase at the primary side of the voltage-boosting modular DC-DC converter, and then converting the leakage inductance L of each phase of the transformer into the leakage inductance L k And feeding back to the primary side to obtain the mathematical model of the boost modular DC-DC converter.
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