CN113422556A - MMC-PET-based passive control driving system of permanent magnet synchronous motor - Google Patents
MMC-PET-based passive control driving system of permanent magnet synchronous motor Download PDFInfo
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
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- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
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- H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
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- H02P21/22—Current control, e.g. using a current control loop
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- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
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- H—ELECTRICITY
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- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
- H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
- H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
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Abstract
The invention relates to a permanent magnet synchronous motor passive control driving system based on MMC-PET, which adopts a 3-stage structure of an input stage, an intermediate stage and an output stage to supply power to a passive network; the input-stage high-voltage electricity rectifies three-phase alternating current through a modular multilevel converter MMC; the intermediate stage adopts a double-active bridge converter with serial input and parallel output to carry out isolation voltage reduction on the direct current output by the input stage; the output stage adopts a three-phase full-bridge inverter, and the output of the three-phase full-bridge inverter is connected with the permanent magnet synchronous motor. The 3-stage MMC-PET system is suitable for high-voltage and high-power occasions while completing basic voltage conversion, has wider application range, and can provide application prospects for higher-voltage occasions by combining with PMSM. Compared with PID control, the passive control strategy provided by the invention has more excellent dynamic and static performances, and the parameter selection is simpler and more convenient.
Description
Technical Field
The invention relates to a driving Control technology of a Permanent Magnet Synchronous Motor, in particular to a Permanent Magnet Synchronous Motor (PMSM) passive-Based (PBC) driving system Based on a Modular Multilevel Converter (MMC) -Power Electronic Transformer (PET) (namely MMC-PET).
Background
The power electronic transformer is a novel intelligent power transformer based on power electronic technology, can complete flexible voltage conversion and energy flow functions, has the advantages of small size, high power density, low noise, less pollution and the like compared with the traditional transformer, is concerned about the world to develop energy Internet vigorously today, and has become an inevitable trend when being widely applied to a power distribution network. Many structures have been proposed for the conventional PET topology, and the adoption of the modular multilevel converter type three-stage structure has the advantages of improving the voltage quality, enabling the active power exchange, and increasing the application range. The permanent magnet synchronous motor has the advantages of low inertia, simple structure, economical operation, high efficiency and the like, and is increasingly widely applied to a plurality of fields such as train traction, wind power generation and the like.
Disclosure of Invention
In order to further improve the control performance of the permanent magnet synchronous motor, the MMC-PET-based passive control driving system of the permanent magnet synchronous motor is provided, the advantages that the MMC-PET is suitable for a high-voltage and high-power system are combined with PMSM passive control, and the system is more stable while normal voltage conversion is ensured.
The technical scheme of the invention is as follows: a permanent magnet synchronous motor passive control driving system based on MMC-PET adopts a 3-stage structure of an input stage, an intermediate stage and an output stage to supply power to a passive network;
an input stage: the high-voltage electricity rectifies three-phase alternating current through a modular multilevel converter MMC;
an intermediate stage: the double-active bridge converter with series input and parallel output is adopted to carry out isolation voltage reduction on the direct current output by the input stage;
an output stage: a three-phase full-bridge inverter is adopted, and the output of the three-phase full-bridge inverter is connected with a permanent magnet synchronous motor.
Preferably, the MMC of the input stage is composed of 6 bridge arms of an upper 2 bridge arms and a lower 2 bridge arms of each phase, and each bridge arm is composed of N sub-modules SM and a bridge arm inductor LsEquivalent resistance R of sum bridge armsAre connected in series; each SM all adopts the structure of half-bridge, contains 2 IGBT that have parallelly connected the diode in the anti-parallel and 1 parallelly connected energy storage capacitor C at 2 IGBT's that establish ties both ends.
Preferably, the method for controlling the MMC of the input stage comprises:
according to MMC topological structure, according to Kirchhoff law, can obtain the mathematical model of MMC transverter and be:
in the formula: u. ofsa、usb、uscRespectively inputting high-voltage end three-phase alternating-current voltages; i.e. isa、isb、iscThree-phase alternating current input into the MMC current converter respectively; l is0、R0The three-phase high-voltage end is connected with each phase of MMC current converter respectively to form a line inductor and a line resistor; u. ofa、ub、ucOutputting three-phase direct current voltage for the MMC current converter respectively; u. ofdcOutputting direct current voltage for the MMC current converter; u. ofjp、ujnUpper and lower arm voltages of j-th phase; i.e. ijp、ijnAre respectively provided withIs the upper and lower arm current of j phase; j is three phases of a, b and c;
obtaining an electromagnetic transient equation of the MMC alternating-current side by the following formula:
wherein,
in the formula: 1,2, N; j ═ a, b, c; n is the number of submodules on a single bridge arm; u. ofsj、isjThe voltage and the current of a j-th phase three-phase alternating current power supply are respectively; req、LeqRespectively an equivalent resistance and an equivalent inductance of the MMC; sjA j-th phase switch control variable is related to the quantity of the SMs input by the upper and lower bridge arms, and aims to control the output voltage of the MMC at the alternating current side; sjpi、SjniThe switching function of the ith sub-module of the jth phase upper and lower bridge arms is obtained;
the electromagnetic transient equation of the direct current side under the abc coordinate system can be obtained by a mathematical model of the MMC current converter and is as follows:
in the formula:
the expected value of the DC side voltage; l issBridge arm inductance; ceqEquivalent capacitors of an upper bridge arm and a lower bridge arm; i.e. idcIs direct current side current;
the mathematical model of the MMC current converter is transformed by abc-dq0, and the mathematical model under a dq rotation coordinate system is as follows:
in the formula: u. ofsd、usqAnd isd、isqThe components of three-phase voltage and current on d and q axes; sd、SqIs SjComponents on d, q axes; omega is angular velocity;
for the derivative calculation of the electromagnetic transient equation of the direct current side under the abc coordinate system, on the premise of omitting the fluctuation of the direct current, namely the derivative of the derivative term of the direct current side current can be obtained by the following steps:
transforming the above equation to dq axis coordinate system to obtain:
and D, carrying out PID control on a current inner ring and a voltage outer ring of the MMC converter mathematical model under the dq coordinate system.
Preferably, the intermediate-stage dual-active bridge converter adopts input voltage-sharing control: the double-active bridge converter comprises 3 sub-modules of an input end DC-AC converter, an AC-AC converter and an output end AC-DC converter, UinnThe input voltage of each submodule on the serial side is n equal to 1,2 and 3; u shapeout、Uout_refRespectively outputting a measured value and a reference value of the voltage at the DC side, and outputting a moving direction comparison reference value d of each submodule after the difference value of the measured value and the reference value is regulated by PIsH_ref；Uin_avThe input voltage of each submodule is the average value of the input voltage of the submoduleinnAnd the average value U of the input voltagein_avThe difference value of the correction coefficient is regulated by PI and then output the shift ratio correction quantity delta d of each sub-modulesHkK is 1,2, 3; shift correction amount Δ d for each sub-modulesHkAnd a reference value dsH_refAs a control value for the phase shift ratio of each sub-module.
Preferably, the PMSM model based on the dissipative Hamilton PCHD is established by the permanent magnet synchronous motor connected with the output stage, and the PMSM model adopts passive control.
Preferably, the PMSM model passive control design method based on dissipative hamilton PCHD comprises: for PMSM at equilibrium point x*Reaching a steady state, and constructing a closed loop expected energy function H by using feedbackd(x) Let it be at x*Taking the minimum value, i.e. at x*Within a neighborhood of
Satisfy Hd(x)>Hd(x*) The feedback control law u is designed to be β (x), and the closed loop system is expressed as:
in the formula: j. the design is a squared(x)、Rd(x) The desired interconnection matrix, damping matrix, satisfies the following relationships:
if the feedback law u is β (x), Ra(x)、Ja(x) And K (x) satisfies the relationship:
in the formula: ra(x)、Ja(x) Respectively, as a function of system performance; when R isa(x)、Ja(x) When 0 is selected, the system convergence rate is uncontrollable, and the system performance is not good; when J isa(x) 0 and Ra(x) When not equal to 0, the system convergence rate is represented by Ra(x) The control performance is better, so that the invention selects J when designing the passive controllera(x) 0 and Ra(x) Scheme not equal to 0; and the following conditions are satisfied:
then the closed loop system is a PCHD system, x*Is a locally asymptotically stable equilibrium point of the closed loop system,
in the formula: ha(x) A function of the pending energy injected into the system for feedback.
The invention has the beneficial effects that: the MMC-PET-based passive control driving system of the permanent magnet synchronous motor combines MMC-PET and PMSM drive, and has the advantages of small size, light weight, high conversion efficiency, good system performance and the like.
Drawings
FIG. 1 is a schematic structural diagram of a MMC-PET-based passive control driving system of a permanent magnet synchronous motor of the invention;
FIG. 2 is a block diagram of the MMC in the system of the present invention;
FIG. 3 is a control block diagram of input voltage sharing in the system of the present invention;
FIG. 4 is a simulation diagram of DC side output voltage in the balanced and unbalanced states of the power grid;
FIG. 5 is a simulation diagram of the capacitance and voltage of an MMC input stage sub-module according to an embodiment of the present invention;
FIG. 6 is a voltage simulation diagram of an input module according to an embodiment of the present invention;
FIG. 7 is a simulation diagram of the intermediate stage output voltage according to an embodiment of the present invention;
FIG. 8 is a diagram illustrating the simulation of the rotation speed of the motor during no-load operation according to the embodiment of the present invention;
FIG. 9 is a motor torque simulation diagram during no-load operation according to an embodiment of the present invention;
FIG. 10 is a simulation diagram of a-phase stator current during no-load operation according to the embodiment of the present invention;
FIG. 11 is a motor speed simulation graph during constant speed operation according to an embodiment of the present invention;
FIG. 12 is a motor torque simulation graph during constant speed operation according to an embodiment of the present invention;
FIG. 13 is a simulation of a-phase stator current at constant speed operation according to an embodiment of the present invention;
FIG. 14 is a motor speed simulation diagram during variable speed operation according to an embodiment of the present invention;
FIG. 15 is a motor torque simulation diagram during a variable speed operation of the embodiment of the present invention;
FIG. 16 is a simulation diagram of a-phase stator current during variable speed operation according to the embodiment of the present invention;
FIG. 17 is a motor rotation speed simulation diagram during variable load operation according to an embodiment of the present invention;
FIG. 18 is a motor torque simulation graph during variable load operation according to an embodiment of the present invention;
fig. 19 is a simulation diagram of a-phase stator current during variable load operation according to the embodiment of the present invention.
Detailed Description
The invention is described in detail below with reference to the figures and specific embodiments. The present embodiment is implemented on the premise of the technical solution of the present invention, and a detailed implementation manner and a specific operation process are given, but the scope of the present invention is not limited to the following embodiments.
The invention relates to a permanent magnet synchronous motor passive control driving system based on MMC-PET, and the system composition and the control method are explained one by one.
1. The MMC-PET system structure and the working principle of supplying power to the passive network are as follows:
FIG. 1 is a block diagram of an MMC-PET system connected to a PMSM. As can be seen, the PET system employs a 3-stage architecture with an input stage, an intermediate stage, and an output stage, wherein: firstly, an input type three-phase rectifier adopting an MMC structure, a high-voltage generator, a fan and the like can be connected to an input stage; the intermediate stage adopts a double-active bridge converter with serial input and parallel output; and thirdly, the output stage adopts a three-phase full-bridge inverter and is connected with a permanent magnet synchronous motor.
FIG. 2 is a topology of an MMC. As can be seen from fig. 2, the MMC is composed of 6 bridge arms of 2 bridge arms of each phase, each of which is composed of n Sub-modules (SM) and a bridge arm inductor LsEquivalent resistance R of sum bridge armsAre connected in series. Each SM has a half-bridge structure and comprises 2 IGBTs (S) with diodes connected in anti-parallelp、Sn) And 1 energy storage capacitor C connected in parallel at two ends of the 2 series IGBTs. According to 2IGBT Sp、SnThe difference of the switch states, when in normal operation, the SM can present two working states, namely an input state and a cutting state. When SM is in the on state, Sp(T1) Conduction, Sn(T2) And (4) switching off, wherein the output voltage of the submodule is the capacitor voltage Uc. When SM is in the off state, Sp(T1) Off, Sn(T2) And conducting, wherein the output voltage of the sub-module is the capacitor voltage 0.
2. According to MMC topological structure, according to Kirchhoff law, can obtain the mathematical model of MMC transverter and be:
in the formula: u. ofsa、usb、uscRespectively inputting high-voltage end three-phase alternating-current voltages; i.e. isa、isb、iscThree-phase alternating current input into the MMC current converter respectively; l is0、R0The three-phase high-voltage end is connected with each phase of MMC current converter respectively to form a line inductor and a line resistor; u. ofa、ub、ucOutputting three-phase direct current voltage for the MMC current converter respectively; u. ofdcOutputting direct current voltage for the MMC current converter; u. ofjp、ujnUpper and lower arm voltages of j-th phase; (ii) a i.e. ijp、ijnThe upper arm current and the lower arm current of the j phase are respectively; j is three phases of a, b and c;
the MMC alternating-current side electromagnetic transient equation obtained by the formula (1) and the formula (2) is as follows:
wherein,
in the formula: 1,2, N; j ═ a, b, c; n is the number of submodules on a single bridge arm; u. ofsj、isjThe voltage and the current of a j-th phase three-phase alternating current power supply are respectively; req、LeqRespectively an equivalent resistance and an equivalent inductance of the MMC; sjA j-th phase switch control variable is related to the quantity of the SMs input by the upper and lower bridge arms, and aims to control the output voltage of the MMC at the alternating current side; sjpi、SjniAnd the switching function of the ith sub-module of the jth phase upper and lower bridge arms is obtained.
The electromagnetic transient equation on the direct current side under the abc coordinate system can be obtained from the formula (2) as follows:
in the formula:
the expected value of the DC side voltage; l issBridge arm inductance; ceqEquivalent capacitors of an upper bridge arm and a lower bridge arm; i.e. idcIs a direct side current.
3. Conversion of three-phase stationary coordinate system to two-phase rotating coordinate system: the mathematical model of the MMC current converter is transformed by abc-dq0, and the mathematical model under a dq rotation coordinate system is as follows:
in the formula: u. ofsd、usqAnd isd、isqThe components of three-phase voltage and current on d and q axes; sd、SqIs SjComponents on d, q axes; ω is the angular velocity.
For the derivative calculation of equation (4), on the premise of omitting the dc current fluctuation, i.e. omitting the derivative of the derivative term of the dc side current in equation (4), it can be deduced that:
transformation of equation (6) to the dq axis coordinate system yields:
and D, carrying out PID control on a current inner ring and a voltage outer ring of the MMC converter mathematical model under the dq coordinate system.
The MMC adopts double closed-loop control, the voltage outer loop is controlled by constant direct current voltage, and the inner loop is controlled by a PID control method, so that the direct current voltage is kept stable, and the active power is kept stable. The control process comprises the following steps: the positive and negative sequence separation system carries out dq coordinate conversion on the collected voltage and current signals, the output signals are transmitted to a voltage outer loop controller, current reference signals and other quantities are transmitted to a PID controller, a switch function is constructed, and the signals are transmitted to a carrier phase shift modulation module in cooperation with circulation suppression control so as to maintain direct-current voltage and active power at the rectifying side.
4. And aiming at the middle-level DAB, input voltage-sharing control is adopted.
The block diagram of the input voltage equalization control structure is shown in fig. 3. In the figure, Uinn(n is 1,2 and 3) is input voltage of each submodule (an input end DC-AC converter, an AC-AC converter and an output end AC-DC converter) at the series side; u shapeout、Uout_refRespectively outputting a measured value and a reference value of the voltage at the DC side, and outputting a moving direction comparison reference value d of each submodule after the difference value of the measured value and the reference value is regulated by PIsH_ref；Uin_avThe input voltage of each submodule is the average value of the input voltage of the submoduleinnAnd the average value U of the input voltagein_avThe difference value of the correction coefficient is regulated by PI and then output the shift ratio correction quantity delta d of each sub-modulesHk(k ═ 1,2, 3); shift correction amount Δ d for each sub-modulesHkAnd a reference value dsH_refAs a control value for the phase shift ratio of each sub-module. The controller finally outputs the actual shift ratio of each submodule. The control method is simple and effective in control and can achieve the aim ofBetter voltage-sharing effect.
5. For a PMSM (permanent magnet synchronous motor) connected with an output stage, a PMSM model is established based on a dissipative Hamilton model (PCHD).
The PCHD model under the state equation form of the permanent magnet synchronous motor is as follows:
in the formula: x is a state variable, x belongs to Rn(ii) a u and y are input and output variables, u and y are belonged to Rm(ii) a R (x) is a port damping matrix, R (x) is-RT(x) Not less than 0; j (x) is an intra-system interconnection matrix, J (x) is-JT(x) (ii) a H (x), f (x), g (x) are energy storage, state variable, and input variable coefficient functions, respectively.
The PMSM voltage equation under the three-phase abc static coordinate system is as follows:
the PMSM flux linkage equation is:
in the formula: the subscript 3s represents the three-phase abc stationary frame; u. of3sIs a three-phase winding phase voltage; r is a motor resistor; i.e. i3sIs the motor current; psi3s、L3s、F3s(θe) Respectively a winding flux linkage, an inductor and a magnetomotive force; thetae、
The rotor electrical position angle and the stator flux linkage are respectively. The variables above can be expressed as:
in the formula: l ism3、Ll3The mutual inductance and the leakage inductance of the stator are respectively.
The expressions of the electromagnetic torque and the motion equation are as follows:
in the formula: omegar、npAnd thetamThe mechanical rotating speed, the pole pair number and the mechanical position angle of the rotor are respectively; J. b is rotational inertia and a damping coefficient respectively, wherein B is small and can be ignored.
The transformation matrix P that transforms the model into dq rotation coordinate system is:
the mathematical model of the PMSM under the dq coordinate system obtained through Park transformation is as follows:
in the formula: u. ofd、uqAre stator voltage d, q axis components; i.e. id、iqAre stator current d, q axis components;
the components of the stator flux linkage axes d and q are shown; and R is the stator resistance.
The stator flux linkage equation is:
by substituting equation (15) for equation (14), the voltage equation in dq coordinate system can be obtained as:
in the formula: l isd、LqThe stator inductances in d and q coordinate systems are respectively.
The electromagnetic torque equation is:
the damping coefficient B is omitted, and a mechanical equation can be deduced as follows:
in the formula: t ise、TLElectromagnetic torque and load torque, respectively.
The state variable x, the input variable u, and the output variable y of the PMSM are respectively defined as:
in the formula: d is a diagonal matrix, D ═ diag { Ld,Lq,J}。
The PMSM energy storage function h (x) is expressed as:
the dq mathematical model of PMSM is expressed as PCHD model form by equation (8):
wherein,
6. and carrying out passive control design on the PMSM based on the PCHD model.
For PMSM system at balance point x*Reaching a steady state, and constructing a closed loop expected energy function H by using feedbackd(x) Let it be at x*Taking the minimum value, i.e. at x*Within a neighborhood of
Satisfy Hd(x)>Hd(x*). The feedback control law u is designed as β (x), and the closed loop system is represented as:
in the formula: j. the design is a squared(x)、Rd(x) The desired interconnection matrix, damping matrix. They satisfy the following relationship:
if the feedback law u is β (x), Ra(x)、Ja(x) And K (x) satisfies the relationship:
in the formula: ra(x)、Ja(x) Respectively, as a function of the system performance, taking R in the present inventiona(x)≠0，Ja(x)＝0。
And the following conditions are satisfied:
then the closed loop system is a PCHD system, x*Is a locally asymptotically stable equilibrium point of the closed loop system.
In the formula: ha(x) A function of the pending energy injected into the system for feedback.
7. In order to verify the advantages of the method of the present invention, the present embodiment performs a simulation comparison experiment by building a simulation model based on MATLAB/Simulink according to the MMC-PET system connected to PMSM. The PMSM and MMC-PET parameters of the model are as follows:
TABLE 1
The specific simulation effect is as follows:
according to the set value of the system, the high-voltage side of the input stage is connected with a 20kV alternating-current system, the balanced state of the power grid and the unbalanced state of reducing the A-phase voltage of the power grid to 18kV are set, and the output voltage of the direct-current bus is shown in figure 4 in the two states. As can be seen from FIG. 4, the output voltage of the DC side is basically stabilized at 20kV no matter whether the network side voltage is in an unbalanced state or not, which provides conditions for the DC/DC conversion of the subsequent intermediate stage. The voltage of the sub-modules of the MMC input stage is shown in figure 5, the voltage is stable at 4.0kV, and the number of the sub-modules is 10.
Through the input voltage-sharing control, the voltage waveform of the direct current side of the single DC-DC converter in a steady state is shown in FIG. 6. As can be seen from FIG. 6, the voltages of the 3 serially-connected input and parallelly-connected output submodules are all distributed between 6666.66V and 6666.68V, the fluctuation amplitude is less than 1%, and the input voltage-sharing effect is ideal.
The intermediate stage output voltage is shown in fig. 7. As can be seen from fig. 7, the DC-DC isolation stage stabilizes the low-voltage side DC voltage at 700V, providing conditions for inversion of the following output stage.
8. In order to verify that the PMSM can stably operate under various operating conditions, the passive control (PBC) drive and the PID control drive of the PMSM based on the MMC-PET are respectively subjected to simulation comparison under 4 working conditions of no-load operation, constant-speed operation, variable-speed operation and variable-load operation of the motor.
(1) No load operation
After voltage conversion of MMC-PET, the voltage of an output stage is stabilized at 700V, and the load torque T isLThe entire system was simulated at 0N · m (no load) and at a rotation speed N of 1000 r/min. Simulation curves of the motor rotating speed, the motor torque and the a-phase stator current in no-load operation are shown in figures 8-10.
As can be seen from FIG. 8, compared with PID control, the rotation speed overshoot of PMSM under passive control is obviously reduced, almost approaches to zero, the stabilization time is shortened, and the speed of the PMSM is increased by 0.002s compared with the former; as can be seen from fig. 9, compared with PID control, PMSM under passive control has smaller torque fluctuation, smoother waveform, and better dynamic performance; as can be seen from fig. 10, the waveform of the a-phase stator current under passive control is smooth, and stabilizes at 0A after 0.015 s.
(2) Constant speed operation
Setting System 0.7s input load Torque TLThe entire system was simulated with the number of revolutions set to 10N · m and N1000 r/min. At this time, simulation curves of the motor rotation speed, the torque and the a-phase stator current are shown in fig. 11-13.
As can be seen from fig. 11, compared with PID control, the overshoot of the rotation speed of the PMSM under passive control is significantly reduced, almost zero, and the stabilization time is significantly shortened, the latter being accelerated by 0.03s compared with the former; as can be seen from fig. 12, compared with PID control, PMSM torque under passive control has almost no fluctuation, shorter stabilization time, and better dynamic performance; as can be seen from fig. 13, the waveform of the a-phase stator current under the passive control is smooth, and stabilizes at the set current value after 0.01 s.
(3) Variable speed operation
Setting the constant load torque T of the systemLWhen the rotation speed is reduced from N1000 r/min to N800 r/min at 1.0s, the whole system is simulated, and the simulation curves of the motor rotation speed, the torque and the a-phase stator current are shown in fig. 14 to 16.
As can be seen from fig. 14, compared with PID control, the overshoot of the rotation speed of PMSM under passive control is significantly reduced, almost zero, stronger anti-interference capability, and significantly shortened settling time, which is increased by 0.025s compared with the former; as can be seen from fig. 15, compared with PID control, PMSM torque fluctuation under passive control is significantly reduced, overshoot is lower, and better dynamic performance is achieved; as can be seen from fig. 16, the waveform of the a-phase stator current under passive control is relatively smooth, and is stabilized at the set current value after 0.004 s.
(4) Variable load operation
The system load torque is set to 1.1s by TL10N · m increased to TLThe whole system is simulated under the condition that the rotating speed is constantly 1000r/min at 15 N.m, and the simulation curves of the rotating speed, the torque and the a-phase stator current of the motor are shown in the graphs of 17-19.
As can be seen from fig. 17, compared with PID control, the overshoot of the rotation speed of PMSM under passive control is significantly reduced, almost zero, stronger anti-interference capability, and significantly shortened settling time, which is accelerated by 0.03s compared with the former; as can be seen from fig. 18, compared with PID control, PMSM torque under passive control has little fluctuation, shorter stabilization time, and better dynamic performance; as can be seen from fig. 19, the a-phase stator current waveform under passive control has no jitter, and a smooth switching effect can be achieved.
In conclusion, the invention provides a passive control driving system based on the MMC-PET permanent magnet synchronous motor, combines the advantages of MMC-PET suitable for a high-voltage and high-power system with PMSM passive control, and theoretically proves the stability of the control system. Finally, simulation verification is carried out on 4 different working conditions of no-load operation, constant-speed operation, variable-speed operation and variable-load operation on a Matlab/Simulink simulation platform, and the following conclusion is obtained through theoretical and experimental analysis:
1) the 3-stage MMC-PET system is suitable for high-voltage and high-power occasions while completing basic voltage conversion, has wider application range, and can provide application prospects for higher-voltage occasions by combining with PMSM.
2) Compared with PID control, the passive control strategy provided by the invention has more excellent dynamic and static performances, and the parameter selection is simpler and more convenient.
The above-mentioned embodiments only express several embodiments of the present invention, and the description thereof is more specific and detailed, but not construed as limiting the scope of the invention. It should be noted that, for a person skilled in the art, several variations and modifications can be made without departing from the inventive concept, which falls within the scope of the present invention. Therefore, the protection scope of the present patent shall be subject to the appended claims.
Claims (6)
1. A permanent magnet synchronous motor passive control driving system based on MMC-PET is characterized in that a 3-stage structure of an input stage, an intermediate stage and an output stage is adopted to supply power to a passive network;
an input stage: the high-voltage electricity rectifies three-phase alternating current through a modular multilevel converter MMC;
an intermediate stage: the double-active bridge converter with series input and parallel output is adopted to carry out isolation voltage reduction on the direct current output by the input stage;
an output stage: a three-phase full-bridge inverter is adopted, and the output of the three-phase full-bridge inverter is connected with a permanent magnet synchronous motor.
2. The MMC-PET-based passive control driving system for the PMSM of claim 1, wherein the MMC of the input stage consists of 6 bridge arms of an upper bridge arm and a lower bridge arm of each phase, and each bridge arm consists of N sub-modules SM and a bridge arm inductor LsEquivalent resistance R of sum bridge armsAre connected in series; each SM isThe half-bridge structure is adopted, and the half-bridge structure comprises 2 IGBTs which are connected with diodes in anti-parallel and 1 energy storage capacitor C which is connected with two ends of the 2 series IGBTs in parallel.
3. The MMC-PET based PMSM passive control drive system of claim 2, characterized in that the control method of the MMC of the input stage:
according to MMC topological structure, according to Kirchhoff law, the mathematical model who gets the MMC transverter is:
in the formula: u. ofsa、usb、uscRespectively inputting high-voltage end three-phase alternating-current voltages; i.e. isa、isb、iscThree-phase alternating current input into the MMC current converter respectively; l is0、R0The three-phase high-voltage end is connected with each phase of MMC current converter respectively to form a line inductor and a line resistor; u. ofa、ub、ucOutputting three-phase direct current voltage for the MMC current converter respectively; u. ofdcOutputting direct current voltage for the MMC current converter; u. ofjp、ujnUpper and lower arm voltages of j-th phase; i.e. ijp、ijnThe upper arm current and the lower arm current of the j phase are respectively; j is three phases of a, b and c;
obtaining an electromagnetic transient equation of the MMC alternating-current side by the following formula:
wherein,
in the formula: 1,2, N; j ═ a, b, c; n is the number of submodules on a single bridge arm; u. ofsj、isjThe voltage and the current of a j-th phase three-phase alternating current power supply are respectively; req、LeqRespectively an equivalent resistance and an equivalent inductance of the MMC; sjA j-th phase switch control variable is related to the quantity of the SMs input by the upper and lower bridge arms, and aims to control the output voltage of the MMC at the alternating current side; sjpi、SjniThe switching function of the ith sub-module of the jth phase upper and lower bridge arms is obtained;
the electromagnetic transient equation of the direct current side under the abc coordinate system can be obtained by a mathematical model of the MMC current converter and is as follows:
in the formula:
the expected value of the DC side voltage; l issBridge arm inductance; ceqEquivalent capacitors of an upper bridge arm and a lower bridge arm; i.e. idcIs direct current side current;
converting the mathematical model of the MMC current converter through abc-dq0 to obtain the mathematical model under a dq rotation coordinate system as follows:
in the formula: u. ofsd、usqAnd isd、isqThe components of three-phase voltage and current on d and q axes; sd、SqIs SjComponents on d, q axes; omega is angular velocity;
for the derivative calculation of the electromagnetic transient equation of the direct current side under the abc coordinate system, on the premise of omitting the fluctuation of the direct current, namely the derivative of the derivative term of the direct current side current can be obtained by the following steps:
transforming the above equation to dq axis coordinate system to obtain:
and D, carrying out PID control on a current inner ring and a voltage outer ring of the MMC converter mathematical model under the dq coordinate system.
4. The MMC-PET based PMSM passive control drive system of claim 1, wherein the mid-stage dual active bridge converter employs input voltage sharing control: the double-active bridge converter comprises 3 sub-modules of an input end DC-AC converter, an AC-AC converter and an output end AC-DC converter, UinnThe input voltage of each submodule on the serial side is n equal to 1,2 and 3; u shapeout、Uout_refRespectively outputting a measured value and a reference value of the voltage at the DC side, and outputting a moving direction comparison reference value d of each submodule after the difference value of the measured value and the reference value is regulated by PIsH_ref；Uin_avThe input voltage of each submodule is the average value of the input voltage of the submoduleinnAnd the average value U of the input voltagein_avThe difference value of the correction coefficient is regulated by PI and then output the shift ratio correction quantity delta d of each sub-modulesHkK is 1,2, 3; shift correction amount Δ d for each sub-modulesHkAnd a reference value dsH_refAs a control value for the phase shift ratio of each sub-module.
5. The MMC-PET-based passive control driving system for the PMSM of claim 1, wherein the output-stage-connected PMSM establishes a PMSM model based on dissipative Hamilton PCHD, the PMSM model using passive control.
6. The MMC-PET based PMSM passive control drive system of claim 5, wherein the MMC-PET based PMSM passive control drive systemThe PMSM model passive control design method for dissipating Hamilton PCHD comprises the following steps: for PMSM at equilibrium point x*Reaching a steady state, and constructing a closed loop expected energy function H by using feedbackd(x) Let it be at x*Taking the minimum value, i.e. at x*Within a neighborhood of
Satisfy Hd(x)>Hd(x*) The feedback control law u is designed to be β (x), and the closed loop system is expressed as:
in the formula: j. the design is a squared(x)、Rd(x) The desired interconnection matrix, damping matrix, satisfies the following relationships:
if the feedback law u is β (x), Ra(x)、Ja(x) And K (x) satisfies the relationship:
in the formula: ra(x)、Ja(x) Respectively, as a function of system performance; when R isa(x)、Ja(x) When 0 is selected, the system convergence rate is uncontrollable, and the system performance is not good; when J isa(x) 0 and Ra(x) When not equal to 0, the system convergence rate is represented by Ra(x) The control performance is better, so that the invention selects J when designing the passive controllera(x) 0 and Ra(x) Scheme not equal to 0; and the following conditions are satisfied:
then the closed loop system is a PCHD system, x*Is a locally asymptotically stable equilibrium point of the closed loop system,
in the formula: ha(x) A function of the pending energy injected into the system for feedback.
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