CN113364377A - Active-disturbance-rejection position servo control method for permanent magnet synchronous motor - Google Patents

Active-disturbance-rejection position servo control method for permanent magnet synchronous motor Download PDF

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CN113364377A
CN113364377A CN202110252531.XA CN202110252531A CN113364377A CN 113364377 A CN113364377 A CN 113364377A CN 202110252531 A CN202110252531 A CN 202110252531A CN 113364377 A CN113364377 A CN 113364377A
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permanent magnet
synchronous motor
magnet synchronous
axis current
signal
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CN113364377B (en
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陈益广
刘宏旭
苏江
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Tianjin University
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Tianjin University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/13—Observer control, e.g. using Luenberger observers or Kalman filters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/12—Stator flux based control involving the use of rotor position or rotor speed sensors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H02P25/024—Synchronous motors controlled by supply frequency
    • H02P25/026—Synchronous motors controlled by supply frequency thereby detecting the rotor position
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

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  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)

Abstract

The invention discloses an active disturbance rejection control method based on an improved extended state observer, aiming at the defects of large tracking error, poor anti-interference capability, poor robustness, low response speed and the like in a position servo system of a traditional permanent magnet synchronous motor, firstly, by introducing speed into the extended state observer as input quantity, obtaining a better observation effect on a second-order state variable; secondly, by introducing the finite time state observer, the state observer not only has the characteristic that the traditional extended state observer can observe disturbance, but also has the advantage that the finite time state observer can be converged in finite time. Compared with the traditional position-speed-current three-closed-loop control method, the method has the advantages of small tracking error, strong anti-interference capability and capability of following a higher-frequency position signal, and improves the performance of a permanent magnet synchronous motor position servo system.

Description

Active-disturbance-rejection position servo control method for permanent magnet synchronous motor
Technical Field
The invention relates to the technical field of control of permanent magnet synchronous motors, in particular to an active disturbance rejection position servo control method based on an improved extended state observer.
Background
In recent years, with the progress in the fields of power electronics technology, modern control theory and the like, the position servo performance of the motor is further improved, so that the servo motor is widely applied in various fields. The permanent magnet synchronous motor is widely applied to servo drive due to the advantages of high efficiency, high reliability, high power density, easiness in control and the like. A traditional permanent magnet synchronous motor position servo system adopts a vector control method, specifically comprises three closed-loop control of a position loop, a speed loop and a current loop, the position loop usually adopts proportional control, the speed loop and the current loop adopt proportional integral control, but the permanent magnet synchronous motor is a multivariable, high-coupling and nonlinear high-order system, the traditional proportional integral control method has the defects of low response speed, large overshoot, poor control performance and the like, and the high-precision and high-response requirements of the modern society on gradual improvement of the position servo system cannot be met.
The active disturbance rejection control can realize the compensation of signals by estimating the internal disturbance and the external disturbance of the system, and the system is set into an integral series system. The method has low dependence degree on the model and strong robustness, and is widely applied.
Active disturbance rejection control also has a number of disadvantages that limit its application to engineering. Because the extended state observer is a core link of the active disturbance rejection control, the number of parameters of the extended state observer is reduced, the tracking precision of the extended state observer on system input is improved, and the state convergence speed is accelerated, so that the method becomes a main direction for researching the current active disturbance rejection control method.
Disclosure of Invention
The invention aims to overcome the defects of the existing method, provides an active disturbance rejection position servo control method based on an improved extended state observer, and realizes the purposes of reducing tracking error, improving disturbance rejection capability, following a position signal with higher frequency and improving the performance of a position servo system by constructing an improved second-order active disturbance rejection controller to replace a position loop and a speed loop in the traditional three-loop control method.
The invention provides a permanent magnet synchronous motor position servo control method which is characterized in that the method is an active disturbance rejection position servo control method based on an improved extended state observer, and comprises the following steps:
step one, toSampling and resolving signals of the permanent magnet synchronous motor, and sampling and resolving by using an absolute position encoder coaxially connected with a permanent magnet synchronous motor rotor to obtain a mechanical angle theta of the permanent magnet synchronous motor rotor position and an electrical angle theta of the permanent magnet synchronous motor rotor positioneAnd mechanical angular velocity of rotor of permanent magnet synchronous motor
Figure BSA0000235517680000011
Method for controlling A, B and C three-phase stator current i of permanent magnet synchronous motor by using non-contact Hall current sensorA、iBAnd iCSampling is carried out;
step two, sampling the current signals i of the permanent magnet synchronous motor A, B and the C three-phase stator obtained in the step oneA、iBAnd iCObtaining alpha axis current i under a two-phase alpha beta static coordinate system through Clark changeαAnd beta axis current iβAnd applying the alpha axis current iαAnd beta axis current iβObtaining the direct axis current i under the dq synchronous rotation coordinate system through Park positive changedAnd quadrature axis current iq；
Step three, giving an external position command theta*Inputting the signal into a differential tracker, and outputting a rotor position reference signal theta of the permanent magnet synchronous motor through the differential trackerrefAnd permanent magnet synchronous motor rotor speed reference signal
Figure BSA0000235517680000021
Step four, the permanent magnet synchronous motor rotor position reference signal theta obtained in the step three is usedrefAnd permanent magnet synchronous motor rotor speed reference signal
Figure BSA0000235517680000022
Respectively obtaining permanent magnet synchronous motor rotor position observation signals Z by the improved extended state observer1And permanent magnet synchronous motor rotor speed observation signal Z2Making difference to obtain rotor position error signal e1And rotor speed error signal e2Then the rotor position error signal e is used1And rotor speed error signal e2Inputting the signal into a nonlinear feedback link, and obtaining a quadrature axis current reference signal without considering the influence of the total disturbance error of the system by the nonlinear feedback link
Figure BSA0000235517680000023
Step five, the mechanical angular speed of the rotor of the permanent magnet synchronous motor obtained in the step one
Figure BSA0000235517680000024
Rotor mechanical angle theta and quadrature axis current reference signal of permanent magnet synchronous motor
Figure BSA0000235517680000025
Inputting the signal into an improved extended state observer, and obtaining a rotor position observation signal Z of the permanent magnet synchronous motor by the improved extended state observer1Permanent magnet synchronous motor rotor speed observation signal Z2And the total disturbance error estimate Z3The expression of the improved extended state observer is:
Figure BSA0000235517680000026
in the formula (1), epsilon1Is a permanent magnet synchronous motor rotor mechanical angle observed value Z1Error of mechanical angle theta with rotor of permanent magnet synchronous motor2Is the observed value Z of the mechanical angular velocity of the rotor of the permanent magnet synchronous motor2Mechanical angular velocity of rotor of permanent magnet synchronous motor
Figure BSA0000235517680000027
Sign () is a sign function, β01、β02、β03And beta04Is the gain coefficient of the system, h is the sampling period, a is the nonlinear coefficient, in the method, a takes the fraction, b0To improve the extended state observer compensation coefficients;
in the process, Z3The method comprises the following steps of (1) including the influence of factors such as unmodeled errors, parameter errors and external interference on the load torque of the motor;
in the method, the compensation coefficient b of the extended state observer is improved0Solving according to the following formula;
Figure BSA0000235517680000028
in the formula (2), pnIs the pole pair number psi of the permanent magnet synchronous motorfThe permanent magnet flux linkage is adopted, J is lumped rotational inertia of a motor end system, and R is resistance of a motor stator phase winding; step six, according to the total disturbance error estimated value Z obtained in the step five3Calculating to obtain a correction current considering the influence of the total disturbance error
Figure BSA0000235517680000029
Step seven, according to the quadrature axis current reference signal obtained in the step four without considering the influence of the total disturbance error of the system
Figure BSA00002355176800000210
And the corrected current obtained in the step six and considering the influence of the total disturbance error
Figure BSA00002355176800000211
Calculating to obtain quadrature axis current reference signal
Figure BSA00002355176800000212
Step eight, the quadrature axis current reference signal in the step seven
Figure BSA00002355176800000213
And the quadrature axis current i obtained in the step twoqMaking a comparison, i.e. obtaining quadrature-axis current reference signals
Figure BSA00002355176800000214
And quadrature axis current iqThe obtained difference is input into a current controller with Proportional Integral (PI) regulation characteristic, and quadrature axis reference voltage is obtained after the current controller regulates the difference
Figure BSA0000235517680000031
Taking a direct axis current reference signal
Figure BSA0000235517680000032
To 0, the direct axis current is referenced to the signal
Figure BSA0000235517680000033
And the direct axis current i obtained in the step twodMaking a comparison, i.e. obtaining a direct-axis current reference signal
Figure BSA0000235517680000034
And the direct axis current idThe obtained difference is input into a current controller with Proportional Integral (PI) regulation characteristic, and a direct-axis reference voltage is obtained after the adjustment of the current controller
Figure BSA0000235517680000035
Step nine, the quadrature axis reference voltage obtained in the step eight
Figure BSA0000235517680000036
And a direct axis reference voltage
Figure BSA0000235517680000037
Obtaining alpha axis reference voltage under stator two-phase alpha beta static coordinate system through Park inverse transformation
Figure BSA0000235517680000038
And beta axis reference voltage
Figure BSA0000235517680000039
Step ten, the alpha axis reference voltage under the stator two-phase alpha beta static coordinate system
Figure BSA00002355176800000310
And beta axis reference voltage
Figure BSA00002355176800000311
Completing SVPWM pulse width calculation in an SVPWM pulse generator to generate SVPWM pulses;
step eleven, inputting the SVPWM pulse generated in the step eleven into an inverter, controlling the inverter to provide corresponding SVPWM pulse width modulation voltage for the permanent magnet synchronous motor, changing the motion trend of the permanent magnet synchronous motor, and realizing the position servo control of the permanent magnet synchronous motor. Compared with the prior art, the invention has the beneficial effects that:
(1) the invention introduces the mechanical speed of the motor into the extended state observer
Figure BSA00002355176800000312
As the input quantity, the speed signal can obtain better following effect, and the observation effect of the extended state observer is improved.
(2) The invention combines the traditional extended state observer and the finite time controller, and introduces a fractional term into the observer, so that the controller has the characteristic of convergence in the finite time, and the controller has better robustness and anti-interference performance.
(3) The invention optimizes the parameter design of the extended state observer, provides a feasible parameter setting method, and reduces the complexity of parameter setting, so that the extended state observer has higher practical application value.
(4) According to the invention, through a series of improvements on the extended state observer, the tracking error of the motor position signal is reduced, the bandwidth of the controller is increased, the system can track the motor position signal with higher frequency, and the control effect of the permanent magnet synchronous motor position servo system is improved.
Drawings
FIG. 1 is a control system block diagram of an auto-disturbance-rejection position servo control method based on an improved extended state observer;
FIG. 2 is a block diagram of a differential tracker;
FIG. 3 is a block diagram of a process for a non-linear feedback loop;
FIG. 4 is a block diagram of a process for improving the extended state observer as set forth herein.
Detailed Description
Embodiments of the present invention will be described in further detail below with reference to the accompanying drawings.
The active disturbance rejection position servo control method based on the improved extended state observer is realized on the basis of hardware of a general permanent magnet synchronous motor digital control driving system. The most basic hardware comprises a digital signal processor, a contactless Hall current sensor, an absolute position encoder (Encorder), an inverter, and a direct current power supply (U)DC) And a Permanent Magnet Synchronous Motor (PMSM). The system control algorithm is implemented in a digital signal processor.
The block diagram of a control system for implementing the active disturbance rejection position servo control method based on the improved extended state observer is shown in fig. 1. The invention is realized by discrete control algorithm, which is executed by digital signal processor.
Firstly, sampling and resolving are carried out on signals of the permanent magnet synchronous motor. Sampling and resolving by using an absolute position encoder coaxially connected with the permanent magnet synchronous motor to obtain a permanent magnet synchronous motor rotor position mechanical angle theta (k) and a permanent magnet synchronous motor rotor position electrical angle thetae(k) And mechanical angular velocity of rotor of permanent magnet synchronous motor
Figure BSA0000235517680000041
Then, a k-th operation period A, B of the permanent magnet synchronous motor and a C-phase three-phase stator current i are performed by using the non-contact Hall current sensorA(k)、iB(k) And iC(k) Sampling is performed.
Then the k-th operation period A, B of the permanent magnet synchronous motor obtained by sampling and a C-phase three-phase stator current signal iA(k)、iB(k) And iC(k) Obtaining alpha axis current i under a two-phase alpha beta static coordinate system through Clark changeα(k) And beta axis current iβ(k) The specific coordinate change expression is as follows:
Figure BSA0000235517680000042
then the alpha axis current i under the two-phase alpha beta static coordinate system is usedα(k) And beta axis current iβ(k) Obtaining the direct axis current i under the dq synchronous rotation coordinate system through Park positive changed(k) And quadrature axis current iq(k) The specific coordinate change expression is as follows:
Figure BSA0000235517680000043
since the measured signal usually introduces a series of noises, which causes errors in the signal, the differential calculation further increases the noise effect.
Therefore, the differential Tracker (TD) is adopted to realize the tracking of the measurement signal and the differential signal, reduce the noise pollution and simultaneously reduce the overshoot of the tracking signal on the premise of ensuring the quick response of the system.
Then, the motor position is commanded to θ*(k) Inputting the signal into a differential tracker, and outputting a position reference signal theta of the permanent magnet synchronous motor through the differential trackerref(k) And a permanent magnet synchronous motor speed reference signal
Figure BSA0000235517680000044
The mathematical expression for the differential tracker is:
Figure BSA0000235517680000045
in the formula (5), r is a maximum value that can be controlled; h is a sampling period; when x is ordered1＝θref(k)-θ*，
Figure BSA0000235517680000046
When, fhan (x)1，x2R, h) is a discrete steepest control integral function; the mathematical expression of the discrete steepest control synthesis function is:
Figure BSA0000235517680000051
in the formula (6), a and a0、a1、a2、d、sa、sgAnd g are both intermediate variables; sign (·) is a sign function.
Then, the obtained position reference signal theta of the permanent magnet synchronous motorref(k) And a permanent magnet synchronous motor speed reference signal
Figure BSA0000235517680000052
And a permanent magnet synchronous motor position observation signal Z obtained by an improved extended state observer1(k) And permanent magnet synchronous motor speed observation signal Z2(k) Differencing to obtain a position error signal e1(k) And the error signal e of the velocity2(k)。
Figure BSA0000235517680000053
In order to enable the control system to better relieve the contradiction between overshoot and quick response, the invention adopts a nonlinear feedback link (NLSEF).
The nonlinear error feedback link introduces nonlinear control on the basis of the traditional control, the method has strong robustness, and the dynamic performance of the system can be improved.
Will position error signal e1(k) And the error signal e of the velocity2(k) Input to a nonlinear feedback link to obtain a quadrature axis current reference signal without considering the influence of the total disturbance error of the system
Figure BSA0000235517680000054
The mathematical expression of the nonlinear feedback link is as follows:
Figure BSA0000235517680000055
in the formula (8) < beta >1、β2Is the gain factor of the system. a is1And a2Is the nonlinear coefficient of the system; fal (-) is a blockThe mathematical expression of the speed optimal control comprehensive function is as follows:
Figure BSA0000235517680000056
in order to enable the position servo system to obtain a better position tracking effect, the invention designs an improved extended state observer (FT-ESO), and the principle of the improved state observer is as follows:
for a system, the second order kinetic equation can be written:
Figure BSA0000235517680000061
in the formula (10), y is the output of the system; y is(t)The derivatives of the orders of the output g; u is the input of the system; b is a constant representing the effect of the input on the output; w (t) is an external perturbation; f [ g (t), w (t), t)]The total disturbance of the system is represented;
representing the system state as x1＝y，
Figure BSA0000235517680000062
The mathematical expression of equation (10) is:
Figure BSA0000235517680000063
in formula (11), H (t) is the derivative of f [ g (t), w (t), t ].
Let the observer state be Z1(k)、Z2(k) And Z3(k) The mathematical expression of the conventional second-order extended state observer in a discrete form is obtained as follows:
Figure BSA0000235517680000064
in the formula (12), ε (k) represents an error; z3(k) The total disturbance error estimated value is obtained; and b is the compensation coefficient of the extended state observer.
The angle, rotating speed, torque, current and voltage equations of the permanent magnet synchronous motor under the dq coordinate system are as follows:
Figure BSA0000235517680000065
in the formula (19), θ is a rotor mechanical angle; omega is the mechanical angular speed of the rotor; j is lumped moment of inertia of the motor end system; b is a friction coefficient; p is a radical ofnThe number of pole pairs of the permanent magnet synchronous motor is; t iseIs an electromagnetic torque; t isLIs the load torque; psifIs a permanent magnet flux linkage; u. ofqAnd udRespectively an alternating voltage and a direct-axis voltage; i.e. iqAnd idQuadrature axis current and direct axis current, respectively; l isqAnd LdA quadrature axis inductor and a direct axis inductor respectively; and R is the resistance of the stator phase winding.
Using idControl is 0, can obtain
Figure BSA0000235517680000071
From formula (14) can be obtained
Figure BSA0000235517680000072
In formula (15), iq0The quadrature axis current when the total disturbance of the system is not considered; i.e. iq1To account for the modified current of the total disturbance of the system.
As can be seen from equation (15), the state variable is the mechanical angular velocity of the rotor of the magnetic synchronous motor
Figure BSA0000235517680000073
And the mechanical angle theta, x of the rotor of the permanent magnet synchronous motor1＝θ，
Figure BSA0000235517680000074
Expanding the system to obtain a traditional second-order expansion state suitable for the systemAn observer having a mathematical expression:
Figure BSA0000235517680000075
wherein Z is1(k) The observed value of the mechanical angle theta (k) of the rotor position of the permanent magnet synchronous motor in the kth operation period is obtained; z2(k) The mechanical angular speed of the rotor of the permanent magnet synchronous motor in the kth operation period
Figure BSA0000235517680000076
The observed value of (a); z3(k) Is the total disturbance error estimated value of the kth operation period, Z in the system3(k) The method comprises the influence of factors such as unmodeled errors, parameter errors, external interference and the like on the load torque of the motor, wherein epsilon (k) is the mechanical angle observation value Z of the rotor position of the permanent magnet synchronous motor in the kth operation period1(k) The error of the mechanical angle theta (k) of the rotor position of the permanent magnet synchronous motor in the kth calculation period;
Figure BSA0000235517680000077
is the quadrature current reference signal of the k-th operation period.
In the process b0The mathematical expression of (a) is:
Figure BSA0000235517680000078
with respect to the present system, it is,
Figure BSA0000235517680000081
the method has practical significance, namely the mechanical angular speed of the rotor of the permanent magnet synchronous motor is obtained; derived by simultaneous differential trackers
Figure BSA0000235517680000082
Also according to the physical meaning of the rotational speed, the speed quantity should be strictly followed. In order to obtain a more optimal tracking condition of the speed, introducing an error of the speed observed quantity into an extended state observer, and rewritingA discrete form of the extended state observer, whose mathematical expression in discrete form is:
Figure BSA0000235517680000083
in formula (18), ε1(k) And the observed value Z of the mechanical angle of the rotor position of the permanent magnet synchronous motor in the kth operation period1(k) The error of the mechanical angle theta (k) of the rotor position of the permanent magnet synchronous motor in the kth calculation period; epsilon2(k) For k operation period permanent magnet synchronous motor rotor mechanical angular velocity observed value Z2(k) And the mechanical angular speed of the rotor of the permanent magnet synchronous motor in the kth operation period
Figure BSA0000235517680000084
The error of (2).
Since the state convergence speed and the state convergence accuracy are important indexes for evaluating the quality of a control system, the finite time control can converge the system from an initial state to a target state in a finite time. The invention introduces the finite time observer into the traditional extended state observer, so that the controller has the characteristic of convergence in finite time and has better robustness and anti-interference performance.
Finite time control is characterized by a fractional term, and the mathematical expression of a discrete form of the discrete form that rewrites a sheet state observer (FT-ESO) is:
Figure BSA0000235517680000085
in the formula (19), a is a fraction in the present invention.
Compared with the formula (18), the formula (19) reduces the number of system parameters and reduces the difficulty of system parameter setting.
Then, the mechanical angular speed of the rotor of the permanent magnet synchronous motor is measured
Figure BSA0000235517680000086
The mechanical angle theta (k) of the rotor position of the permanent magnet synchronous motor and the last period are stored and registeredQuadrature current reference signal of device
Figure BSA0000235517680000087
Inputting the signal into an improved extended state observer to obtain a permanent magnet synchronous motor rotor position observation signal Z1(k) And permanent magnet synchronous motor rotor speed observation signal Z2(k) And the total disturbance error estimate Z3(k)。
Then according to the total disturbance error estimated value Z3(k) Calculating to obtain a correction current considering the influence of the total disturbance error
Figure BSA0000235517680000088
The mathematical expression is as follows:
Figure BSA0000235517680000091
from equation (15), the quadrature current reference signal
Figure BSA0000235517680000092
The mathematical expression is:
Figure BSA0000235517680000093
then, will
Figure BSA0000235517680000094
Stored in a register and the next cycle is input to the modified extended state observer.
Then, the quadrature axis current reference signal in the formula (21)
Figure BSA0000235517680000095
And quadrature axis current i obtained through a coordinate transformation moduleq(k) Making a comparison, i.e. obtaining quadrature-axis current reference signals
Figure BSA0000235517680000096
And quadrature axis current iq(k) Difference of (2)Inputting the obtained difference value into a current loop PI controller to obtain a quadrature axis reference voltage
Figure BSA0000235517680000097
Taking a direct axis current reference signal
Figure BSA0000235517680000098
To 0, the direct axis current is referenced to the signal
Figure BSA0000235517680000099
And the direct axis current i obtained by the coordinate transformation moduled(k) Making a comparison, i.e. obtaining a direct-axis current reference signal
Figure BSA00002355176800000910
And the direct axis current id(k) The obtained difference is input into a current loop PI controller to obtain a direct axis reference voltage
Figure BSA00002355176800000911
Then, the obtained quadrature axis reference voltage is used
Figure BSA00002355176800000912
And a direct axis reference voltage
Figure BSA00002355176800000913
Obtaining alpha axis reference voltage under stator two-phase static alpha beta coordinate system through Park inverse transformation
Figure BSA00002355176800000914
And beta axis reference voltage
Figure BSA00002355176800000915
The specific coordinate change expression is as follows:
Figure BSA00002355176800000916
then stator two-phase stationary alpha betaReference voltage of alpha axis under coordinate system
Figure BSA00002355176800000917
And beta axis reference voltage
Figure BSA00002355176800000918
And completing SVPWM pulse width calculation in an SVPWM pulse generator to generate SVPWM pulses.
And finally, inputting the generated SVPWM pulse into an inverter, controlling the inverter to provide corresponding SVPWM pulse width modulation voltage for the permanent magnet synchronous motor, changing the motion trend of the permanent magnet synchronous motor, and realizing the position servo control of the permanent magnet synchronous motor.
The invention introduces the mechanical angular velocity of the rotor of the permanent magnet synchronous motor into the extended state observer
Figure BSA00002355176800000919
As the input quantity, the speed signal can obtain better following effect; the traditional extended state observer is combined with a finite time controller, and a fractional term is introduced into the observer, so that the controller has the characteristic of convergence in finite time, and the controller has better robustness and anti-interference performance; by optimizing the parameter design of the extended state observer, a feasible parameter setting method is provided, the complexity of parameter setting is reduced, and the active disturbance rejection control has higher practical application value; the tracking error of the motor position signal is reduced, the bandwidth of the controller is increased, the system can track the motor position signal with higher frequency, and the control effect of the permanent magnet synchronous motor position servo system is improved.
The foregoing embodiments illustrate and describe the general principles, principal features, and advantages of the invention. Those of ordinary skill in the art will understand that: the discussion of the above embodiments is merely exemplary. Therefore, any omissions, modifications, substitutions, improvements and the like that may be made without departing from the spirit and principles of the invention are intended to be included within the scope of the invention.

Claims (1)

1. A permanent magnet synchronous motor position servo control method is characterized in that the method is an active disturbance rejection position servo control method based on an improved extended state observer, and comprises the following steps:
sampling and resolving signals of the permanent magnet synchronous motor, and sampling and resolving by using an absolute position encoder coaxially connected with a permanent magnet synchronous motor rotor to obtain a permanent magnet synchronous motor rotor position mechanical angle theta and a permanent magnet synchronous motor rotor position electrical angle thetaeAnd mechanical angular velocity of rotor of permanent magnet synchronous motor
Figure FSA0000235517670000011
Method for controlling A, B and C three-phase stator current i of permanent magnet synchronous motor by using non-contact Hall current sensorA、iBAnd iCSampling is carried out;
step two, sampling the current signals i of the permanent magnet synchronous motor A, B and the C three-phase stator obtained in the step oneA、iBAnd iCObtaining alpha axis current i under a two-phase alpha beta static coordinate system through Clark changeαAnd beta axis current iβAnd applying the alpha axis current iαAnd beta axis current iβObtaining the direct axis current i under the dq synchronous rotation coordinate system through Park positive changedAnd quadrature axis current iq；
Step three, giving an external position command theta*Inputting the signal into a differential tracker, and outputting a rotor position reference signal theta of the permanent magnet synchronous motor through the differential trackerrefAnd permanent magnet synchronous motor rotor speed reference signal
Figure FSA0000235517670000012
Step four, the permanent magnet synchronous motor rotor position reference signal theta obtained in the step three is usedrefAnd permanent magnet synchronous motor rotor speed reference signal
Figure FSA0000235517670000013
Respectively connected with a permanent magnet synchronous motor rotor position obtained by an improved extended state observerObservation signal Z1And permanent magnet synchronous motor rotor speed observation signal Z2Making difference to obtain rotor position error signal e1And rotor speed error signal e2Then the rotor position error signal e is used1And rotor speed error signal e2Inputting the signal into a nonlinear feedback link, and obtaining a quadrature axis current reference signal without considering the influence of the total disturbance error of the system by the nonlinear feedback link
Figure FSA0000235517670000014
Step five, the mechanical angular speed of the rotor of the permanent magnet synchronous motor obtained in the step one
Figure FSA0000235517670000015
Rotor mechanical angle theta and quadrature axis current reference signal of permanent magnet synchronous motor
Figure FSA0000235517670000016
Inputting the signal into an improved extended state observer, and obtaining a rotor position observation signal Z of the permanent magnet synchronous motor by the improved extended state observer1Permanent magnet synchronous motor rotor speed observation signal Z2And the total disturbance error estimate Z3The expression of the improved extended state observer is:
Figure FSA0000235517670000017
in the formula (1), epsilon1Is a permanent magnet synchronous motor rotor mechanical angle observed value Z1Error of mechanical angle theta with rotor of permanent magnet synchronous motor2Is the observed value Z of the mechanical angular velocity of the rotor of the permanent magnet synchronous motor2Mechanical angular velocity of rotor of permanent magnet synchronous motor
Figure FSA0000235517670000018
Sign () is a sign function, β01、β02、β03And beta04Is the gain of the systemCoefficient, h is sampling period, a is nonlinear coefficient, in the method a is taken as fraction, b0To improve the extended state observer compensation coefficients;
in the process, Z3The method comprises the following steps of (1) including the influence of factors such as unmodeled errors, parameter errors and external interference on the load torque of the motor;
in the method, the compensation coefficient b of the extended state observer is improved0Solving according to the following formula;
Figure FSA0000235517670000021
in the formula (2), pnIs the pole pair number psi of the permanent magnet synchronous motorfThe permanent magnet flux linkage is adopted, J is lumped rotational inertia of a motor end system, and R is resistance of a motor stator phase winding;
step six, according to the total disturbance error estimated value Z obtained in the step five3Calculating to obtain a correction current considering the influence of the total disturbance error
Figure FSA0000235517670000022
Step seven, according to the quadrature axis current reference signal obtained in the step four without considering the influence of the total disturbance error of the system
Figure FSA0000235517670000023
And the corrected current obtained in the step six and considering the influence of the total disturbance error
Figure FSA0000235517670000024
Calculating to obtain quadrature axis current reference signal
Figure FSA0000235517670000025
Step eight, the quadrature axis current reference signal in the step seven
Figure FSA0000235517670000026
And step twoQuadrature axis current i obtained inqMaking a comparison, i.e. obtaining quadrature-axis current reference signals
Figure FSA0000235517670000027
And quadrature axis current iqThe obtained difference is input into a current controller with Proportional Integral (PI) regulation characteristic, and quadrature axis reference voltage is obtained after the current controller regulates the difference
Figure FSA0000235517670000028
Taking a direct axis current reference signal
Figure FSA0000235517670000029
To 0, the direct axis current is referenced to the signal
Figure FSA00002355176700000210
And the direct axis current i obtained in the step twodMaking a comparison, i.e. obtaining a direct-axis current reference signal
Figure FSA00002355176700000211
And the direct axis current idThe obtained difference is input into a current controller with Proportional Integral (PI) regulation characteristic, and a direct-axis reference voltage is obtained after the adjustment of the current controller
Figure FSA00002355176700000212
Step nine, the quadrature axis reference voltage obtained in the step eight
Figure FSA00002355176700000213
And a direct axis reference voltage
Figure FSA00002355176700000214
Obtaining alpha axis reference voltage under stator two-phase alpha beta static coordinate system through Park inverse transformation
Figure FSA00002355176700000215
And beta axis reference voltage
Figure FSA00002355176700000216
Step ten, the alpha axis reference voltage under the stator two-phase alpha beta static coordinate system
Figure FSA00002355176700000217
And beta axis reference voltage
Figure FSA00002355176700000218
Completing SVPWM pulse width calculation in an SVPWM pulse generator to generate SVPWM pulses;
step eleven, inputting the SVPWM pulse generated in the step eleven into an inverter, controlling the inverter to provide corresponding SVPWM pulse width modulation voltage for the permanent magnet synchronous motor, changing the motion trend of the permanent magnet synchronous motor, and realizing the position servo control of the permanent magnet synchronous motor.
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CN114123881A (en) * 2021-11-30 2022-03-01 深圳市麦格米特驱动技术有限公司 Motor position estimation method, motor control method and device
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CN114257132A (en) * 2021-12-27 2022-03-29 江苏大学 Optimized anti-interference controller for permanent magnet synchronous motor of electric vehicle
CN114310911A (en) * 2022-02-08 2022-04-12 天津大学 Neural network-based dynamic error prediction and compensation system and method for driving joint
CN114310911B (en) * 2022-02-08 2023-06-30 天津大学 Driving joint dynamic error prediction and compensation system and method based on neural network
CN114488782A (en) * 2022-04-18 2022-05-13 中国科学院西安光学精密机械研究所 Turntable double-position ring control method and system based on harmonic speed reducing mechanism
CN114928295A (en) * 2022-05-25 2022-08-19 福州大学 Permanent magnet synchronous motor variable structure active disturbance rejection control method based on error insertion

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