CN113364330B - Output harmonic wave minimized 3D-SVPWM modulation strategy overmodulation method and system - Google Patents

Output harmonic wave minimized 3D-SVPWM modulation strategy overmodulation method and system Download PDF

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CN113364330B
CN113364330B CN202110747675.2A CN202110747675A CN113364330B CN 113364330 B CN113364330 B CN 113364330B CN 202110747675 A CN202110747675 A CN 202110747675A CN 113364330 B CN113364330 B CN 113364330B
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modulation
voltage vector
svpwm
reference voltage
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CN113364330A (en
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杨淑英
王顺
谢震
刘芳
张兴
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Hefei University of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
    • H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53875—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
    • H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H02P25/024—Synchronous motors controlled by supply frequency
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/02—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using supply voltage with constant frequency and variable amplitude
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control

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Abstract

An overmodulation method and a system of an output harmonic minimization 3D-SVPWM modulation strategy belong to the technical field of inverter modulation, solve the problem of how to calculate the maximum linear modulation degree of the 3D-SVPWM modulation strategy, obtain the maximum linear modulation degree of the 3D-SVPWM modulation strategy by calculation while not changing the characteristics of simultaneously modulating a differential mode component and a common mode component of the 3D-SVPWM modulation strategy and provide a spatial modulation body judgment of a spatial modulation reference vector, combine a corresponding compression plane constraint equation and an output harmonic minimization compression scheme constraint equation during overmodulation, improve the modulation range of the 3D-SVPWM modulation strategy by the overmodulation technology on the premise of not changing the characteristics of simultaneously modulating the differential mode component and the common mode component of the 3D-SVPWM modulation strategy, the planar phase angle tracking during overmodulation is kept, output harmonic waves during overmodulation are minimized, the overmodulation process is smooth, and the applicability of a 3D-SVPWM (three-dimensional space vector pulse width modulation) strategy is effectively enhanced.

Description

Output harmonic wave minimized 3D-SVPWM modulation strategy overmodulation method and system
Technical Field
The invention belongs to the technical field of inverter modulation, and relates to an output harmonic minimization 3D-SVPWM (space vector pulse width modulation) modulation strategy overmodulation method and system.
Background
The 3D-SVPWM (three-dimensional space vector pulse width modulation) strategy can simultaneously modulate the differential mode component and the common mode component of the reference voltage vector, so that the method is widely applied to zero-sequence circulating current control needing to modulate common mode voltage, such as open winding topological structures of a common direct current bus and a common neutral line double inverter, a three-phase four-leg inverter and the like. Compared with the traditional SVPWM (space vector pulse width modulation) strategy only focusing on differential mode component modulation, the 3D-SVPWM strategy is limited in the modulation output range of the differential mode component because the common mode component is modulated at the same time, so that the modulation range is expanded by related linear modulation range calculation and a corresponding over-modulation scheme, and the applicability of the 3D-SVPWM strategy is improved.
In the prior art, a document 3D-SVPWM modulation strategy of a four-leg three-level inverter (3D-SVPWM modulation strategy) with a publication date of 10 months in 2019 (anyhow, etc., power system and its automatic chemical report, 31 vol. 10, 125, 132) discloses that a 3D-SVPWM modulation strategy is applied to a four-leg three-level inverter, but the document does not specifically provide the maximum linear modulation degree of the 3D-SVPWM modulation strategy and a related overmodulation strategy. The document 'flux weakening control strategy of a common direct current bus open-winding permanent magnet synchronous motor' with publication date of 2018, 11 and 5 (year honing, etc., China Motor engineering reports, volume 38, No. 21, 6461, 6469 in 2018) discloses that a 3D-SVPWM modulation strategy is used for realizing zero-sequence circulating current closed-loop control of a common direct current bus double-inverter open-winding system, but the linear modulation range and related overmodulation strategy of the 3D-SVPWM modulation strategy are not given.
In summary, the prior art has the following problems: 1) for a 3D-SVPWM modulation strategy, the prior art only provides a basic synthesis principle and an implementation process, and does not provide a linear modulation range, namely a maximum linear modulation degree, of the 3D-SVPWM modulation strategy; 2) and a constraint scheme when the 3D-SVPWM modulation strategy is overmodulatied is not given, so that the overmodulation constraint scheme cannot be adopted to effectively expand the application range of the 3D-SVPWM modulation strategy.
Disclosure of Invention
The invention aims to calculate the maximum linear modulation degree of a 3D-SVPWM (space vector pulse width modulation) strategy, and on the premise of not changing the characteristic that the 3D-SVPWM strategy simultaneously modulates a differential mode component and a common mode component, plane phase angle tracking during overmodulation is kept through an overmodulation technology, output harmonic during overmodulation is minimized, and the modulation range of the 3D-SVPWM strategy is expanded.
The invention solves the technical problems through the following technical scheme:
an output harmonic minimization 3D-SVPWM modulation strategy overmodulation method comprises the following steps:
step S1, calculating a reference voltage vector V ref Of the alpha-beta plane component m 1 Phase of the magnetic flux
Figure BDA0003143507290000021
And modulation degree M 1 Calculating a reference voltage vector V ref Gamma axis component V of γ Amplitude m of 3 And phase
Figure BDA0003143507290000022
Calculating a reference voltage vector V ref Characteristic phase difference of
Figure BDA0003143507290000023
Step S2, according to the amplitude m in step S1 3 And characteristic phase difference
Figure BDA0003143507290000024
Calculating the maximum linear modulation degree M of the 3D-SVPWM modulation strategy max1 And maximum compression modulation degree M max2 ；
Step S3, according to the modulation degree M 1 Maximum linear modulation M max1 And maximum compression modulation degree M max2 And performing overmodulation judgment:
when calculating M 1 <M max1 In the time, the linear modulation region is adopted, the wave-sending control of the linear modulation region is adopted, and the method comprises the following steps: according to V α 、V β And V γ Calculating the base voltage vector action time t by using a 3D-SVPWM (space vector pulse width modulation) modulation strategy 0 、t 1 、t 2 And t 7 Wave generation control is carried out;
when calculating M max1 ≤M 1 ≤M max2 In the time, the wave generation control of the overmodulation region is adopted as the overmodulation region, and the method comprises the following steps:
when t is 0 T is not less than 0 7 When the value is more than or equal to 0, the value is a circular arc area of an overmodulation area according to V α 、V β And V γ Calculating the base voltage vector action time t by using a 3D-SVPWM (space vector pulse width modulation) modulation strategy 0 、t 1 、t 2 And t 7 Wave generation control is carried out;
when t is 0 <0 or t 7 <At 0, for the boundary region of the overmodulation region, for the reference voltage vector V ref The α - β plane component of (a) is modified:
first according to V α 、V β And V γ For reference voltage vector V ref Judging the serial number of the modulation body in which the modulation body is positioned, and selecting a corresponding compression plane constraint equation according to the judged serial number of the modulation body;
then, the modified reference voltage vector alpha axis component is obtained by simultaneous calculation with the constraint equation of the output harmonic wave minimum compression scheme
Figure BDA0003143507290000025
Reference voltage vector beta axis component
Figure BDA0003143507290000026
The constraint equation of the output harmonic minimization compression scheme is as follows:
Figure BDA0003143507290000027
finally, according to the modified reference voltage vector alpha axis component
Figure BDA0003143507290000028
Modified reference voltage vector beta axis component
Figure BDA0003143507290000029
And V γ Calculating the action time of the modified base voltage vector by using a 3D-SVPWM (space vector pulse width modulation) strategy
Figure BDA0003143507290000031
And
Figure BDA0003143507290000032
wave generation control is performed.
According to the technical scheme, the maximum linear modulation degree of the 3D-SVPWM modulation strategy is obtained through calculation, the judgment of a space modulation body where a space modulation reference vector is located is given, a corresponding compression plane constraint equation and an output harmonic wave minimum compression scheme constraint equation are simultaneously established during overmodulation, on the premise that the characteristic that the 3D-SVPWM modulation strategy simultaneously modulates a differential mode component and a common mode component is not changed, the modulation range of the 3D-SVPWM modulation strategy is improved through an overmodulation technology, plane phase angle tracking during overmodulation is kept, output harmonic waves during overmodulation are minimized, the overmodulation process is smooth, and the applicability of the 3D-SVPWM modulation strategy is effectively enhanced.
As a further improvement of the technical scheme of the invention, the reference voltage vector V is calculated in step S1 ref Of the alpha-beta plane component m 1 Phase of
Figure BDA0003143507290000033
And modulation degree M 1 The formula of (1) is:
Figure BDA0003143507290000034
wherein, V α 、V β Are respectively a reference voltage vector V ref Projection components of coordinate axes alpha and beta in a three-dimensional space coordinate system are converted into direct current voltage U dc Per-unit value.
As a further improvement of the technical scheme of the invention, the reference voltage vector V is calculated in step S1 ref Gamma axis component V of γ Amplitude m of 3 And phase
Figure BDA0003143507290000035
The formula of (1) is:
Figure BDA0003143507290000036
wherein, V γ，1 Is a first orthogonal component, V γ，2 Is a second orthogonal component; the first and second orthogonal components are pairs V γ Orthogonal decomposition is performed to obtain two components which are 90 degrees apart.
As a further improvement of the technical scheme of the invention, the reference voltage vector V is calculated in step S1 ref Characteristic phase difference of
Figure BDA0003143507290000037
The calculation formula is as follows:
Figure BDA0003143507290000041
wherein the content of the first and second substances,
Figure BDA0003143507290000042
is a reference voltage vector V ref Gamma axis component V of γ The phase of (a) is determined,
Figure BDA0003143507290000043
is a reference voltage vector V ref The phase of the alpha-beta plane component of (a).
As a further improvement of the technical scheme of the invention, in step S2, the maximum linear modulation degree M of the 3D-SVPWM modulation strategy is calculated max1 And maximum compression modulation degree M max2 The method specifically comprises the following steps:
defining a functional formula W 1 ，
Figure BDA0003143507290000044
Defining a functional formula W 2 ，
Figure BDA0003143507290000045
Wherein, theta 1 Has a value range of
Figure BDA0003143507290000046
At theta 1 Within the value range of (A) calculating the function formula W 1 Minimum value of W 1min Calculating a functional formula W 2 Minimum value of W 2min When W is 1min ≤W 2min When M is in contact with max1 ＝W 1min When W is 1min >W 2min When M is in contact with max1 ＝W 2min Calculating a functional curve W 1 Sum function formula W 2 The function value of the curve intersection point is M max2 。
As a further improvement of the technical solution of the present invention, the specific manner of judging the modulation entity number is as follows:
defining the intermediate variables judged by the serial number of the modulation body as a first variable A, a second variable B, a third variable C, a fourth variable N and a fifth variable A 1 A sixth variable B 1 The seventh variable C 1 The eighth variable N 1 Define the functional formula F 1 ，
Figure BDA0003143507290000047
Definition function formula F 2 ，F 2 ＝2V γ -V α Define the functional formula F 3 ，
Figure BDA0003143507290000048
Definition function formula F 4 ，F 4 ＝V β Define the functional formula F 5 ，
Figure BDA0003143507290000049
Definition function formula F 6 ，
Figure BDA00031435072900000410
Then:
when F is present 1 When the value is more than or equal to 0, A is 1; when F is present 1 <When 0, A is 0; when F is present 2 When the value is more than or equal to 0, B is 1; when F is present 2 <When 0, B is 0; when F is present 3 When the content is more than or equal to 0, C is 1; when F is present 3 <When 0, C is 0; n ═ a +2B + 4C;
when F is present 4 When the value is more than or equal to 0, A 1 1 is ═ 1; when F is 4 <At 0, A 1 0; when F is present 5 When not less than 0, B 1 1 is ═ 1; when F is present 5 <At 0, B 1 0; when F is present 6 When not less than 0, C 1 1 is ═ 1; when F is present 6 <At 0 time, C 1 ＝0；N 1 ＝A 1 +2B 1 +4C 1 ；
Each value of the fourth variable N corresponds to a modulation entity number, which is as follows: n ═ 5 corresponds to modulator 1; n1 corresponds to modulator 2; n ═ 3 corresponds to the modulator 3; n ═ 2 corresponds to the modulator 4; n ═ 6 corresponds to the modulator 5; n-4 corresponds to the modulator 6; when N is 0, an eighth variable N is required to be combined 1 Making a further determination of the value, wherein when N is present 1 1 or N 1 When the value is 3, the value corresponds to the modulation body 2, and when N is 1 4 or N 1 When the number is 5, the number is N, the modulation body 4 is corresponded 1 2 or N 1 When it is 6, it corresponds to the modulator 6; when N is 7, an eighth variable N is required to be combined 1 Making a further determination of the value, wherein when N 1 2 or N 1 When the value is 3, the value corresponds to the modulation body 1, and when N is 1 1 or N 1 When the number is 5, the number is N, the modulation body 3 is concerned 1 Or 4 or N 1 When 6, the corresponding modulator 5 is used.
As a further improvement of the technical solution of the present invention, the specific way of selecting the corresponding compression plane constraint equation by the judged serial number of the modulation body is as follows:
the compression plane constraint equation for the modulator 1 is:
Figure BDA0003143507290000051
the compression plane constraint equation of the modulation volume 2 is:
Figure BDA0003143507290000052
the compression plane constraint equation of the modulator body 3 is:
Figure BDA0003143507290000053
the compression plane constraint equation of the modulation volume 4 is:
Figure BDA0003143507290000054
the compression plane constraint equation of the modulator 5 is:
Figure BDA0003143507290000055
the compression plane constraint equation for the modulator 6 is:
Figure BDA0003143507290000056
a system applied to the output harmonic minimization 3D-SVPWM modulation strategy overmodulation method comprises the following steps: first DC source U dc1 A second DC source U dc2 The three-phase two-level inverter comprises a first three-phase two-level inverter VSI1, a second three-phase two-level inverter VSI2, a three-phase stator winding OEWIM, a neutral line I, a capacitor C1, a capacitor C2, a capacitor C3 and a capacitor C4;
the capacitor C1 and the capacitor C2 are connected in series and then connected to a first direct current source U dc1 Between the direct current positive bus P and the direct current negative bus N, a common node of the capacitor C1 and the capacitor C2 is marked as a point O; the capacitor C3 and the capacitor C4 are connected in series and then connected to a second direct current source U dc2 Between the positive dc bus P 'and the negative dc bus N', the common node of the capacitors C3 and C4 is denoted as point O ', the neutral line I connects the point O and the point O', and the first dc source U dc1 And a second DC source U dc2 All direct current voltages are U dc ；
In the three-phase bridge arm of the first three-phase two-level inverter VSI1, each phase of bridge arm includes 2 switching tubes with anti-parallel diodes, that is, the first three-phase two-level inverter VSI1 includes 6 switching tubes with anti-parallel diodes in total, and 6 switching tubes are respectively marked as S n1j N represents a phase sequence, n is a, b, c, j represents the serial number of the switching tube, and j is 1, 2; the three-phase bridge arms of the first three-phase two-level inverter VSI1 are connected in parallel between the direct current positive bus P and the direct current negative bus N, namely a switch tube S a11 、S b11 、S c11 The collectors are connected in parallel and then are connected with a direct current positive bus P and a switching tube S a12 、S b12 、S c12 The emitting electrodes are connected in parallel and then connected with a direct current negative bus N; in the three-phase leg of the first three-phase two-level inverter VSI1, the switching tube S a11 And a switching tube S a12 Series, switch tube S b11 And a switching tube S b12 Series, switch tube S c11 And a switching tube S c12 The connection points of the series connection are respectively marked as three-phase bridge arm middle points a of the first three-phase two-level inverter VSI1 1 、b 1 、c 1 ；
In the three-phase bridge arm of the second three-phase two-level inverter VSI2, each phase of bridge arm includes 2 switching tubes with anti-parallel diodes, that is, the second three-phase two-level inverter VSI2 includes 6 switching tubes with anti-parallel diodes in total, and 6 switching tubes are respectively marked as S n2j (ii) a The three-phase bridge arms of the second three-phase two-level inverter VSI2 are connected in parallel between the direct current positive bus P 'and the direct current negative bus N', namely a switch tube S a21 、S b21 、S c21 The collectors are connected in parallel and then connected with a direct current positive bus P', and a switching tube S a22 、S b22 、S c22 The emitting electrodes are connected in parallel and then connected with a direct current negative bus N'; in the three-phase leg of the second three-phase two-level inverter VSI2, the switching tube S a21 And a switching tube S a22 Series, switch tube S b21 And a switching tube S b22 Series, switch tube S c21 And a switching tube S c22 The connection points of the series connection are respectively marked as three-phase bridge arm middle points a of the second three-phase two-level inverter VSI2 2 、b 2 、c 2 ；
The three-phase stator winding OEWIM comprises three-phase windings, and the left ports of the A-phase winding, the B-phase winding and the C-phase winding are respectively connected with the three-phase bridge arm midpoint a of the first three-phase two-level inverter VSI1 1 、b 1 、c 1 The right ports of the A-phase winding, the B-phase winding and the C-phase winding are respectively connected with the three-phase bridge arm midpoint a of the second three-phase two-level inverter VSI2 2 、b 2 、c 2 。
The invention has the advantages that:
according to the technical scheme, the maximum linear modulation degree of the 3D-SVPWM modulation strategy is obtained through calculation, the judgment of a space modulation body where a space modulation reference vector is located is given, a corresponding compression plane constraint equation and an output harmonic wave minimum compression scheme constraint equation are simultaneously established during overmodulation, the modulation range of the 3D-SVPWM modulation strategy is expanded through the overmodulation technology while the characteristics that the differential mode component and the common mode component are simultaneously modulated by the 3D-SVPWM modulation strategy are not changed, plane phase angle tracking during overmodulation is kept, the output harmonic wave during overmodulation is minimized, the overmodulation process is smooth, and the applicability of the 3D-SVPWM modulation strategy is effectively enhanced.
Drawings
FIG. 1 is a common-neutral open-winding topology as referred to in the present invention;
FIG. 2 is a flow chart of an over-modulation operation in any of the modulators of the embodiments of the present invention;
FIG. 3 is an illustration of a general modulation scheme of a 3D-SVPWM modulation strategy according to an embodiment of the present invention;
FIG. 4 is a diagram illustrating a 3D-SVPWM modulation strategy modulator according to an embodiment of the present invention;
FIG. 5 shows the gamma component V of the reference voltage vector calculated in step S1 when the parameters are accurate γ Amplitude m of 3 Schematic diagram of the variation situation of (1);
FIG. 6 shows the reference voltage vector V calculated in step S1 when the parameters are accurate in the experiment ref Characteristic phase difference of
Figure BDA0003143507290000071
Schematic diagram of the variation of (1);
FIG. 7 is a functional curve W plotted in step S2 when the parameters are accurate in the experiment 1 Sum function formula W 2 Curve and maximum linear modulation degree M of 3D-SVPWM modulation strategy obtained by calculation max1 And maximum compression modulation degree M max2 A schematic diagram;
fig. 8 is a schematic diagram of a change in amplitude of a fundamental voltage of a total output of a common-neutral open-winding electric drive system when modulation degree M is 0.6 to 0.8 is modulated by using a 3D-SVPWM strategy under the condition that each parameter is accurate in an experiment;
fig. 9 shows the common-mode voltage amplitude output by the common-center open-winding electric drive system when the modulation degree M is 0.6 to 0.8 in the experiment when the modulation degree M is 0.6;
fig. 10 is a schematic diagram of changes in fundamental voltage THD output by the common-neutral open-winding electric drive system when modulation is performed using the 3D-SVPWM strategy from the modulation degree M of 0.6 to the modulation degree M of 0.8 when the parameters are accurate in the experiment;
fig. 11 is a graph showing the trend of the variation of the fourth variable N in the judgment of the modulator measured in the experiment.
Detailed Description
In order to make the objects, technical solutions and advantages of the embodiments of the present invention clearer, the technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the embodiments of the present invention, and it is obvious that the described embodiments are some embodiments of the present invention, but not all embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
The technical scheme of the invention is further described by combining the drawings and the specific embodiments in the specification:
example one
FIG. 1 is a three-phase two-level voltage source inverter topology as referred to in the present invention, from which it can be seen that the common neutral open winding electric drive system topology as referred to in the present strategy comprises a first DC source U dc1 A second DC source U dc2 The three-phase two-level inverter comprises a first three-phase two-level inverter VSI1, a second three-phase two-level inverter VSI2, a three-phase stator winding OEWIM, a neutral line I, a capacitor C1, a capacitor C2, a capacitor C3 and a capacitor C4;
the capacitor C1 and the capacitor C2 are connected in series and then connected to a first direct current source U dc1 Between the direct current positive bus P and the direct current negative bus N, a common node of the capacitor C1 and the capacitor C2 is marked as a point O; the capacitor C3 and the capacitor C4 are connected in series and then connected to a second direct current source U dc2 Between the positive dc bus P 'and the negative dc bus N', the common node of the capacitors C3 and C4 is denoted as point O ', the neutral line I connects the point O and the point O', and the first dc source U dc1 And a second DC source U dc2 All direct current voltages are U dc ；
Of said first three-phase two-level inverter VSI1In the three-phase bridge arms, each phase of bridge arm comprises 2 switching tubes with anti-parallel diodes, that is, the first three-phase two-level inverter VSI1 comprises 6 switching tubes with anti-parallel diodes in total, and 6 switching tubes are respectively marked as S n1j N represents a phase sequence, n is a, b, c, j represents the serial number of the switching tube, and j is 1, 2; the three-phase bridge arms of the first three-phase two-level inverter VSI1 are connected in parallel between the direct current positive bus P and the direct current negative bus N, namely a switch tube S a11 、S b11 、S c11 The collectors are connected in parallel and then are connected with a direct current positive bus P and a switching tube S a12 、S b12 、S c12 The emitting electrodes are connected in parallel and then connected with a direct current negative bus N; in the three-phase leg of the first three-phase two-level inverter VSI1, the switching tube S a11 And a switching tube S a12 Series, switch tube S b11 And a switching tube S b12 Series, switch tube S c11 And a switching tube S c12 The connection points of the series connection are respectively marked as three-phase bridge arm middle points a of the first three-phase two-level inverter VSI1 1 、b 1 、c 1 ；
In the three-phase bridge arm of the second three-phase two-level inverter VSI2, each phase of bridge arm includes 2 switching tubes with anti-parallel diodes, that is, the second three-phase two-level inverter VSI2 includes 6 switching tubes with anti-parallel diodes in total, and 6 switching tubes are respectively marked as S n2j (ii) a The three-phase bridge arms of the second three-phase two-level inverter VSI2 are connected in parallel between the direct current positive bus P 'and the direct current negative bus N', namely a switch tube S a21 、S b21 、S c21 The collectors are connected in parallel and then are connected with a direct current positive bus P', and a switching tube S a22 、S b22 、S c22 The emitting electrodes are connected in parallel and then connected with a direct current negative bus N'; in the three-phase leg of the second three-phase two-level inverter VSI2, the switching tube S a21 And a switching tube S a22 Series, switch tube S b21 And a switching tube S b22 Series, switch tube S c21 And a switching tube S c22 Are connected in series, and the connection points of the three-phase bridge arms are respectively marked as three-phase bridge arm midpoint a of the second three-phase two-level inverter VSI2 2 、b 2 、c 2 ；
The three-phase stator winding OEWIM comprises a three-phase winding, an A-phase winding and a B-phase windingThe left ports of the group and the C-phase winding are respectively connected with the three-phase bridge arm midpoint a of the first three-phase two-level inverter VSI1 1 、b 1 、c 1 The right ports of the A-phase winding, the B-phase winding and the C-phase winding are respectively connected with the midpoint a of a three-phase bridge arm of a second three-phase two-level inverter VSI2 2 、b 2 、c 2 。
The invention comprises the following steps:
fig. 2 is a flow chart of overmodulation operation in any of the modulators according to the embodiments of the present invention, corresponding to steps 1 to 3.
Step 1, setting a reference voltage vector to be modulated of a three-phase two-level inverter as V ref The reference voltage vector is V ref Corresponding to the reference voltage vector output by any one of the two inverters VSI1 and VSI2, and referring to the voltage vector V ref Direct current voltage U is used for projection components of coordinate axes alpha, beta and gamma in a three-dimensional space coordinate system dc Performing per unit, and respectively marking as the alpha-axis component V of the reference voltage vector α Reference voltage vector beta axis component V β Reference voltage vector gamma axis component V γ Using the reference voltage vector alpha-axis component V α Reference voltage vector beta axis component V β Calculating to obtain a reference voltage vector V ref Of the alpha-beta plane component m 1 Reference voltage vector V ref Phase of the alpha-beta plane component of (1)
Figure BDA0003143507290000091
And a reference voltage vector V ref Modulation degree M corresponding to the alpha-beta plane component of 1 The calculation formula is as follows:
Figure BDA0003143507290000092
for reference voltage vector gamma axis component V γ Orthogonal decomposition is carried out to obtain two components with 90-degree phase difference, which are respectively marked as gamma-axis components V of reference voltage vectors γ First orthogonal component V of γ，1 And a reference voltage vector gamma-axis component V γ Second orthogonal component V of γ，2 Meter for measuringCalculating to obtain a gamma-axis component V of the reference voltage vector γ Amplitude m of 3 And a reference voltage vector gamma-axis component V γ Phase of
Figure BDA0003143507290000093
The calculation formula is as follows:
Figure BDA0003143507290000101
calculating a reference voltage vector V ref Characteristic phase difference of
Figure BDA0003143507290000102
The calculation formula is as follows:
Figure BDA0003143507290000103
step 2, obtaining the gamma axis component V of the reference voltage vector according to the step 1 γ Amplitude m of 3 And reference voltage vector V ref Characteristic phase difference of
Figure BDA0003143507290000104
Calculating the maximum linear modulation degree M of the 3D-SVPWM modulation strategy max1 And maximum compression modulation degree M max2 ：
Defining a functional formula W 1 The following were used:
Figure BDA0003143507290000105
defining a functional formula W 2 The following were used:
Figure BDA0003143507290000106
wherein theta is 1 Has a value range of
Figure BDA0003143507290000107
At theta 1 Within the value range of (A) calculating the function formula W 1 Minimum value of W 1min Calculating a functional formula W 2 Minimum value of W 2min When W is 1min ≤W 2min When M is in contact with max1 ＝W 1min When W is 1min >W 2min When M is in contact with max1 ＝W 2min Calculating a functional curve W 1 Sum function formula W 2 The function value of the curve intersection point is M max2 。
Step 3, obtaining a reference voltage vector V according to the step 1 and the step 2 ref Modulation degree M corresponding to the alpha-beta plane component of 1 And the maximum linear modulation degree M of the 3D-SVPWM modulation strategy max1 And maximum compression modulation degree M max2 Carrying out overmodulation judgment;
when calculating M 1 <M max1 If so, the method is a linear modulation region and enters the step 3.1;
when calculating M max1 ≤M 1 ≤M max2 If so, entering a step 3.2 for an overmodulation region;
step 3.1, calculate M 1 <M max1 Linear modulation region of time according to the reference voltage vector alpha-axis component V α Reference voltage vector beta axis component V β And a reference voltage vector gamma-axis component V γ Calculating the base voltage vector action time t by using a 3D-SVPWM modulation strategy 0 Base voltage vector action time t 1 Base voltage vector action time t 2 And base voltage vector action time t 7 Wave generation control is carried out; time t 0 、t 1 、t 2 、t 7 The specific calculation process of (2) is referred to pages 6461 and 6469 of a literature "flux weakening control strategy of a common direct current bus open winding permanent magnet synchronous motor" (yearly honing, etc., China Motor engineering, vol 38, No. 21 in 2018), the publication date of which is 2018, 11, month and 5.
Step 3.2, calculate M max1 ≤M 1 ≤M max2 An overmodulation region of time;
step 3.21, according to the reference voltage vector alpha axis component V α Reference voltage vector beta axis component V β Root of HeshenGamma component V of reference voltage vector γ Calculating the base voltage vector action time t by using a 3D-SVPWM (space vector pulse width modulation) modulation strategy 0 Base voltage vector action time t 1 Base voltage vector action time t 2 And base voltage vector action time t 7 ；
When t is 0 T is not less than 0 7 When the value is more than or equal to 0, the value is a circular arc area of the overmodulation area, and the step 3.22 is carried out;
when t is 0 <0 or t 7 <When 0, the boundary area of the overmodulation area is reached, and the step 3.23 is carried out;
step 3.22, the action time t of the basic voltage vector is obtained by calculation 0 Base voltage vector action time t 1 Base voltage vector action time t 2 And base voltage vector action time t 7 Wave generation control is carried out;
step 3.23, for reference voltage vector V ref Is modified according to the alpha-axis component V of the reference voltage vector α Reference voltage vector beta axis component V β And a reference voltage vector gamma-axis component V γ For reference voltage vector V ref The modulation body is used for judging the serial number of the modulation body, the corresponding compression plane constraint equation is selected according to the judged serial number of the modulation body, and then the modified reference voltage vector alpha-axis component is obtained by simultaneous calculation with the constraint equation of the output harmonic wave minimum compression scheme
Figure BDA0003143507290000111
Reference voltage vector beta axis component
Figure BDA0003143507290000112
According to the alpha-axis component of the modified reference voltage vector
Figure BDA0003143507290000113
Reference voltage vector beta axis component
Figure BDA0003143507290000114
And a reference voltage vector gamma-axis component V γ Calculating a base voltage vector using a 3D-SVPWM modulation strategyTime of measurement
Figure BDA0003143507290000115
Base voltage vector action time
Figure BDA0003143507290000116
Base voltage vector action time
Figure BDA0003143507290000117
And base voltage vector action time
Figure BDA0003143507290000118
Wave generation control is carried out; time of day
Figure BDA0003143507290000119
The specific calculation process of (2) is referred to 6461-6469 pages in the literature "flux weakening control strategy of common direct current bus open winding permanent magnet synchronous motor" (yearly honing, etc., china motor engineering report, vol 38, vol 21, in 2018), whose publication date is 2018, 11/5/11/5.
The specific way of judging the modulation body serial number is as follows: defining the intermediate variables judged by the serial number of the modulation body as a first variable A, a second variable B, a third variable C, a fourth variable N and a fifth variable A 1 A sixth variable B 1 The seventh variable C 1 The eighth variable N 1 Define the functional formula F 1 ，
Figure BDA0003143507290000121
Definition function formula F 2 ，F 2 ＝2V γ -V α Define the functional formula F 3 ，
Figure BDA0003143507290000122
Definition function formula F 4 ，F 4 ＝V β Define the functional formula F 5 ，
Figure BDA0003143507290000123
Definition function formula F 6 ，
Figure BDA0003143507290000124
Then:
when F is present 1 When the value is more than or equal to 0, A is 1,
when F is 1 <When the number is 0, A is 0,
when F is present 2 When the value is more than or equal to 0, B is 1,
when F is present 2 <When the number of the carbon atoms is 0, B is 0,
when F is present 3 When the carbon content is more than or equal to 0, C is 1,
when F is present 3 <When the ratio is 0, C is 0,
N＝A+2B+4C，
when F is present 4 When the value is more than or equal to 0, A 1 ＝1，
When F is present 4 <At 0, A 1 ＝0，
When F is present 5 When not less than 0, B 1 ＝1，
When F is 5 <At 0, B 1 ＝0，
When F is present 6 When not less than 0, C 1 ＝1，
When F is present 6 <At 0 time, C 1 ＝0，
N 1 ＝A 1 +2B 1 +4C 1 ，
Each value of the fourth variable N corresponds to a modulation entity number, which is as follows:
n ═ 5 corresponds to modulator 1; n1 corresponds to modulator 2; n ═ 3 corresponds to the modulator 3; n ═ 2 corresponds to the modulator 4; n ═ 6 corresponds to the modulator 5; n-4 corresponds to the modulator 6; when N is 0, an eighth variable N is required to be combined 1 Making a further determination of the value, wherein when N is present 1 1 or N 1 When the value is 3, the value corresponds to the modulation body 2, and when N is 1 4 or N 1 When the number is 5, the number is N, the modulation body 4 is corresponded 1 2 or N 1 When it is 6, it corresponds to the modulator 6; when N is 7, an eighth variable N is required to be combined 1 Making a further determination of the value, wherein when N is present 1 2 or N 1 When the value is 3, the value corresponds to the modulation body 1, and when N is 1 1 or N 1 When the number is 5, the number is N, the modulation body 3 is concerned 1 Or 4 or N 1 When 6, the corresponding modulator 5 is used.
The corresponding relation between different values of the fourth variable N and the serial number of the modulation body is shown in the following table:
Figure BDA0003143507290000131
fig. 3 is an illustration diagram of a total modulation body of a 3D-SVPWM modulation strategy in an embodiment of the present invention, which is the total modulation body of the 3D-SVPWM modulation strategy on an α - β - γ three-dimensional space.
Fig. 4 is a diagram illustrating a 3D-SVPWM modulation strategy modulation entity separately according to an embodiment of the present invention, where the total modulation entity in fig. 3 is divided into six modulation entities and numbered.
The specific way of selecting the corresponding compression plane constraint equation according to the judged modulation body serial number is as follows:
the compression plane constraint equation of the modulator 1 is:
Figure BDA0003143507290000132
the compression plane constraint equation of the modulation volume 2 is:
Figure BDA0003143507290000133
the compression plane constraint equation of the modulator body 3 is:
Figure BDA0003143507290000134
the compression plane constraint equation of the modulation volume 4 is:
Figure BDA0003143507290000135
the compression plane constraint equation of the modulator 5 is:
Figure BDA0003143507290000136
the compression plane constraint equation for the modulator 6 is:
Figure BDA0003143507290000137
the output harmonic minimization compression scheme constraint equation is as follows:
Figure BDA0003143507290000138
namely, the wave-sending control of the 3D-SVPWM modulation strategy with overmodulation is realized.
In order to verify the effectiveness of the invention, the invention was experimentally verified. Topological structure first direct current source U of common neutral open winding electric drive system dc1 And a second DC source U dc2 D.c. voltage U dc 280V, the main circuits of the first three-phase two-level inverter VSI1 and the second three-phase two-level inverter VSI2 are composed of Mitsubishi intelligent IGBT power module PM100CLA120, and the switching frequency f s The dead band is set at 3 mus at 9600 Hz. Using a three-phase asynchronous motor as a load, the asynchronous motor parameters: rated power p n 3kW, rated phase voltage U N 220V, stator resistance R s 1.93 omega, mutual inductance L m 0.19H, stator inductance L s 0.21H, pole pair number P2, operating frequency f e 50 Hz. The reference voltage vectors needing to be modulated of the common neutral open-winding electric drive system are decoupled by 180 degrees and are evenly distributed to the first three-phase two-level inverter VSI1 and the second three-phase two-level inverter VSI2 for modulation, namely the reference voltage vectors needing to be modulated by the two three-phase two-level inverters are equal in size and opposite in direction.
Fig. 5 shows the gamma component V of the reference voltage vector calculated in step 1 for the first three-phase two-level inverter VSI1 γ Amplitude m of 3 Approximately 0.212, corresponding to a common mode voltage requirement of 59.4V for the first three-phase two-level inverter VSI1, and 118.72V for the total common mode voltage requirement of the common neutral open winding electric drive system.
Fig. 6 shows the calculation of the reference voltage vector V of the first three-phase two-level inverter VSI1 in step 1 ref Characteristic phase difference
Figure BDA0003143507290000141
About 2.4.
FIG. 7 shows the gamma component V at the reference voltage vector γ Amplitude m of 3 About 0.212, reference voltage vectorQuantity V ref Characteristic phase difference of
Figure BDA0003143507290000142
About 2.4 by step 2 1 Formula of sum function W 2 Curve, maximum linear modulation degree M of the corresponding first three-phase two-level inverter VSI1 obtained through calculation by using 3D-SVPWM modulation strategy max1 0.6096, namely, the maximum output of the first three-phase two-level inverter VSI1 is linearly modulated by using a 3D-SVPWM modulation strategy, and the maximum compression modulation degree M is calculated max2 Is 1.04.
Fig. 8 shows a reference voltage vector V of a first three-phase two-level inverter VSI1 ref Modulation degree M corresponding to the alpha-beta plane component of 1 And when the voltage rises from 0.6 to 0.8, the amplitude of the fundamental wave voltage of the total output of the common neutral open-winding electric drive system is modulated by using a 3D-SVPWM modulation strategy. Therefore, the over-modulation strategy can effectively break through the traditional linear modulation maximum constraint of 216.8V and increase the amplitude of the fundamental wave voltage of the total output to about 255.4V, and effectively increase the modulation range of the 3D-SVPWM modulation strategy.
Fig. 9 shows a reference voltage vector V of a first three-phase two-level inverter VSI1 ref Modulation degree M corresponding to the alpha-beta plane component of 1 When the voltage rises from 0.6 to 0.8, the common-mode voltage amplitude value of the total output of the common-neutral open-winding electric drive system is modulated by using a 3D-SVPWM modulation strategy, so that the output common-mode voltage amplitude value can be kept unchanged while overmodulation is realized, and the modulation of common-mode components required by zero-sequence loop closed-loop control such as common-neutral open winding is met.
Fig. 10 shows a reference voltage vector V of a first three-phase two-level inverter VSI1 ref Modulation degree M corresponding to the alpha-beta plane component of 1 When the voltage rises from 0.6 to 0.8, the change situation of the fundamental voltage THD of the total output of the common neutral open winding electric drive system is modulated by using a 3D-SVPWM modulation strategy, and the fundamental voltage THD is smaller and the harmonic wave performance is better in the scheme compared with other overmodulation schemes.
In the experiment, the determination of the modulation entity number was also verified in the following manner.
As can be seen from fig. 11, the value of the fourth variable N varies sequentially within one fundamental wave period, corresponding to the reference voltage vector V ref The modulation body 1 to the modulation body 6 rotate continuously for one circle, and the judgment of the modulation body serial number is accurate and effective.
The above examples are only intended to illustrate the technical solution of the present invention, but not to limit it; although the present invention has been described in detail with reference to the foregoing embodiments, it will be understood by those of ordinary skill in the art that: the technical solutions described in the foregoing embodiments may still be modified, or some technical features may be equivalently replaced; and such modifications or substitutions do not depart from the spirit and scope of the corresponding technical solutions of the embodiments of the present invention.

Claims (8)

1. An output harmonic minimization 3D-SVPWM modulation strategy overmodulation method is characterized by comprising the following steps:
step S1, calculating a reference voltage vector V ref Of the alpha-beta plane component m 1 Phase of the magnetic flux
Figure FDA0003627643390000011
And modulation degree M 1 Calculating a reference voltage vector V ref Gamma axis component V of γ Amplitude m of 3 And phase
Figure FDA0003627643390000012
Calculating a reference voltage vector V ref Characteristic phase difference of
Figure FDA0003627643390000013
Step S2, according to the amplitude m in step S1 3 And characteristic phase difference
Figure FDA0003627643390000014
Calculating the maximum linear modulation degree M of the 3D-SVPWM modulation strategy max1 And maximum compression modulation degree M max2 ；
Step S3, according to the modulation degree M 1 Maximum linear modulation M max1 And maximum compression modulation degree M max2 And performing overmodulation judgment:
when calculating M 1 <M max1 In the time, the linear modulation area is adopted for wave sending control, and the method comprises the following steps: according to V α 、V β And V γ Calculating the base voltage vector action time t by using a 3D-SVPWM (space vector pulse width modulation) modulation strategy 0 、t 1 、t 2 And t 7 Wave generation control is carried out;
when calculating M max1 ≤M 1 ≤M max2 In the time, the wave generation control of the overmodulation region is adopted as the overmodulation region, and the method comprises the following steps:
when t is 0 T is not less than 0 7 When the value is more than or equal to 0, the value is a circular arc area of an overmodulation area according to V α 、V β And V γ Calculating the base voltage vector action time t by using a 3D-SVPWM (space vector pulse width modulation) modulation strategy 0 、t 1 、t 2 And t 7 Wave generation control is carried out;
when t is 0 <0 or t 7 <At 0, for the boundary region of the overmodulation region, for the reference voltage vector V ref The α - β plane component of (a) is modified:
first according to V α 、V β And V γ For reference voltage vector V ref Judging the serial number of the modulation body in which the modulation body is positioned, and selecting a corresponding compression plane constraint equation according to the judged serial number of the modulation body;
and then, the modified reference voltage vector alpha-axis component is obtained by simultaneous calculation with the constraint equation of the output harmonic minimization compression scheme
Figure FDA0003627643390000015
Reference voltage vector beta axis component
Figure FDA0003627643390000016
The constraint equation of the output harmonic minimization compression scheme is as follows:
Figure FDA0003627643390000017
finally, according to the modified reference voltage vector alpha axis component
Figure FDA0003627643390000018
Modified reference voltage vector beta axis component
Figure FDA0003627643390000019
And V γ Calculating the action time of the modified base voltage vector by using a 3D-SVPWM (space vector pulse width modulation) strategy
Figure FDA0003627643390000021
And
Figure FDA0003627643390000022
wave generation control is performed.
2. The output harmonic minimization 3D-SVPWM modulation strategy overmodulation method according to claim 1, characterized in that in step S1, a reference voltage vector V is calculated ref Of the alpha-beta plane component m 1 Phase of
Figure FDA0003627643390000023
And modulation degree M 1 The formula of (1) is:
Figure FDA0003627643390000024
wherein, V α 、V β Are respectively a reference voltage vector V ref Projection components of coordinate axes alpha and beta in a three-dimensional space coordinate system are converted into direct current voltage U dc Per-unit values.
3. The output harmonic minimization 3D-SVPWM modulation strategy overmodulation method of claim 2, characterized in thatIn step S1, a reference voltage vector V is calculated ref Gamma axis component V of γ Amplitude m of 3 And phase
Figure FDA0003627643390000025
The formula of (1) is:
Figure FDA0003627643390000026
wherein the content of the first and second substances,
Figure FDA0003627643390000027
is a first orthogonal component, V γ，2 Is a second orthogonal component; the first and second orthogonal components are pairs V γ Orthogonal decomposition is carried out to obtain two components which are 90 degrees different in phase.
4. The overmodulation method of output harmonic minimization 3D-SVPWM modulation strategy according to claim 2, characterized in that the reference voltage vector V is calculated in step S1 ref Characteristic phase difference of
Figure FDA0003627643390000028
The calculation formula is as follows:
Figure FDA0003627643390000029
wherein the content of the first and second substances,
Figure FDA00036276433900000210
as a reference voltage vector V ref Gamma axis component V of γ The phase of (a) is determined,
Figure FDA00036276433900000211
as a reference voltage vector V ref The phase of the alpha-beta plane component of (a).
5. Root of herbaceous plantThe overmodulation method of claim 2 for an output harmonic minimization 3D-SVPWM modulation scheme, wherein the maximum linear modulation degree M of the 3D-SVPWM modulation scheme is calculated in step S2 max1 And maximum compression modulation degree M max2 The method specifically comprises the following steps:
definition function formula W 1 ，
Figure FDA0003627643390000031
Defining a functional formula W 2 ，
Figure FDA0003627643390000032
Wherein, theta 1 Has a value range of
Figure FDA0003627643390000033
At theta 1 Within a value range of (c) calculating a functional formula W 1 Minimum value of W 1min Calculating a functional formula W 2 Minimum value of W 2min When W is 1min ≤W 2min When M is in contact with max1 ＝W 1min When W is 1min >W 2min When M is in contact with max1 ＝W 2min Calculating a functional curve W 1 Sum function formula W 2 The function value of the curve intersection point is M max2 。
6. The output harmonic minimization 3D-SVPWM modulation strategy overmodulation method according to claim 1, characterized in that the specific way of judging the modulation order number is:
defining the intermediate variables judged by the serial number of the modulation body as a first variable A, a second variable B, a third variable C, a fourth variable N and a fifth variable A 1 Sixth variable B 1 The seventh variable C 1 The eighth variable N 1 Define the functional formula F 1 ，
Figure FDA0003627643390000034
Definition function formula F 2 ，F 2 ＝2V γ -V α Define the functional formula F 3 ，
Figure FDA0003627643390000035
Definition function formula F 4 ，F 4 ＝V β Define the functional formula F 5 ，
Figure FDA0003627643390000036
Definition function formula F 6 ，
Figure FDA0003627643390000037
Then:
when F is present 1 When the value is more than or equal to 0, A is 1; when F is present 1 <When 0, A is 0; when F is 2 When the value is more than or equal to 0, B is 1; when F is present 2 <When 0, B is 0; when F is present 3 When the carbon content is more than or equal to 0, C is 1; when F is present 3 <When 0, C is 0; n ═ a +2B + 4C;
when F is present 4 When the value is more than or equal to 0, A 1 1; when F is present 4 <At 0, A 1 0; when F is present 5 When not less than 0, B 1 1 is ═ 1; when F is present 5 <At 0, B 1 0; when F is present 6 When not less than 0, C 1 1; when F is present 6 <At 0 time, C 1 ＝0；N 1 ＝A 1 +2B 1 +4C 1 ；
Each value of the fourth variable N corresponds to a modulation entity number, which is as follows: n ═ 5 corresponds to modulator 1; n1 corresponds to modulator 2; n ═ 3 corresponds to the modulator 3; n ═ 2 corresponds to the modulator 4; n ═ 6 corresponds to the modulator 5; n-4 corresponds to the modulator 6; when N is 0, an eighth variable N is required to be combined 1 Making a further determination of the value, wherein when N is present 1 1 or N 1 When the value is 3, the value corresponds to the modulation body 2, and when N is 1 4 or N 1 When the number is 5, the number is N, the modulation body 4 is corresponded 1 2 or N 1 When it is 6, it corresponds to the modulator 6; when N is 7, an eighth variable N is required to be combined 1 Making a further determination of the value, wherein when N is present 1 2 or N 1 When the value is 3, the value corresponds to the modulation body 1, and when N is 1 1 or N 1 When the number is 5, the number is N, the modulation body 3 is concerned 1 4 or N 1 When 6, the corresponding modulator 5 is used.
7. The overmodulation method of an output harmonic minimization 3D-SVPWM modulation strategy according to claim 1, characterized in that the specific way to select the corresponding compression plane constraint equation from the judged modulation body number is:
the compression plane constraint equation of the modulator 1 is:
Figure FDA0003627643390000041
the compression plane constraint equation of the modulation volume 2 is:
Figure FDA0003627643390000042
the compression plane constraint equation of the modulator body 3 is:
Figure FDA0003627643390000043
the compression plane constraint equation of the modulation volume 4 is:
Figure FDA0003627643390000044
the compression plane constraint equation for the modulator 5 is:
Figure FDA0003627643390000045
the compression plane constraint equation for the modulator 6 is:
Figure FDA0003627643390000046
8. a system for applying the output harmonic minimization 3D-SVPWM modulation strategy overmodulation method according to any of claims 1-7, comprising: first DC source U dc1 A second DC source U dc2 A first three-phase two-level inverter VThe three-phase two-level inverter comprises SI1, a second three-phase two-level inverter VSI2, a three-phase stator winding OEWIM, a neutral wire I, a capacitor C1, a capacitor C2, a capacitor C3 and a capacitor C4;
the capacitor C1 and the capacitor C2 are connected in series and then connected to a first direct current source U dc1 Between the direct current positive bus P and the direct current negative bus N, a common node of the capacitor C1 and the capacitor C2 is marked as a point O; the capacitor C3 and the capacitor C4 are connected in series and then connected to a second direct current source U dc2 Between the positive dc bus P 'and the negative dc bus N', the common node of the capacitors C3 and C4 is denoted as point O ', the neutral line I connects the point O and the point O', and the first dc source U dc1 And a second DC source U dc2 All direct current voltages are U dc ；
In the three-phase bridge arm of the first three-phase two-level inverter VSI1, each phase of bridge arm includes 2 switching tubes with anti-parallel diodes, that is, the first three-phase two-level inverter VSI1 includes 6 switching tubes with anti-parallel diodes in total, and 6 switching tubes are respectively marked as S n1j Wherein n represents the phase sequence, n is a, b, c, j represents the serial number of the switching tube, and j is 1, 2; the three-phase bridge arms of the first three-phase two-level inverter VSI1 are connected in parallel between the direct current positive bus P and the direct current negative bus N, namely a switch tube S a11 、S b11 、S c11 The collectors are connected in parallel and then are connected with a direct current positive bus P and a switching tube S a12 、S b12 、S c12 The emitting electrodes are connected in parallel and then connected with a direct current negative bus N; in the three-phase leg of the first three-phase two-level inverter VSI1, the switching tube S a11 And a switching tube S a12 Series, switch tube S b11 And a switching tube S b12 Series, switch tube S c11 And a switching tube S c12 The connection points of the series connection are respectively marked as three-phase bridge arm middle points a of the first three-phase two-level inverter VSI1 1 、b 1 、c 1 ；
In the three-phase bridge arm of the second three-phase two-level inverter VSI2, each phase of bridge arm includes 2 switching tubes with anti-parallel diodes, that is, the second three-phase two-level inverter VSI2 includes 6 switching tubes with anti-parallel diodes in total, and 6 switching tubes are respectively marked as S n2j (ii) a Three-phase bridge arms of the second three-phase two-level inverter VSI2 are connected in parallel between a direct-current positive bus P' and a direct-current negative busBetween N', i.e. switching tubes S a21 、S b21 、S c21 The collectors are connected in parallel and then connected with a direct current positive bus P', and a switching tube S a22 、S b22 、S c22 The emitting electrodes are connected in parallel and then connected with a direct current negative bus N'; in the three-phase leg of the second three-phase two-level inverter VSI2, the switching tube S a21 And a switching tube S a22 Series, switch tube S b21 And a switching tube S b22 Series, switch tube S c21 And a switching tube S c22 The connection points of the series connection are respectively marked as three-phase bridge arm middle points a of the second three-phase two-level inverter VSI2 2 、b 2 、c 2 ；
The three-phase stator winding OEWIM comprises three-phase windings, and the left ports of the A-phase winding, the B-phase winding and the C-phase winding are respectively connected with the three-phase bridge arm midpoint a of the first three-phase two-level inverter VSI1 1 、b 1 、c 1 The right ports of the A-phase winding, the B-phase winding and the C-phase winding are respectively connected with the three-phase bridge arm midpoint a of the second three-phase two-level inverter VSI2 2 、b 2 、c 2 。
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