CN113346770A - Sliding mode control method of three-level NPC converter - Google Patents

Sliding mode control method of three-level NPC converter Download PDF

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CN113346770A
CN113346770A CN202110692178.7A CN202110692178A CN113346770A CN 113346770 A CN113346770 A CN 113346770A CN 202110692178 A CN202110692178 A CN 202110692178A CN 113346770 A CN113346770 A CN 113346770A
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active power
level npc
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npc converter
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CN113346770B (en
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吴立刚
刘健行
孙光辉
沈肖宁
殷允飞
高亚斌
姚蔚然
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Harbin Institute of Technology Shenzhen
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/02—Conversion of AC power input into DC power output without possibility of reversal
    • H02M7/04—Conversion of AC power input into DC power output without possibility of reversal by static converters
    • H02M7/12—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/145—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means
    • H02M7/155—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for AC mains or AC distribution networks
    • H02J3/01—Arrangements for reducing harmonics or ripples
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for AC mains or AC distribution networks
    • H02J3/36—Arrangements for transfer of electric power between AC networks via a high-tension DC link
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for AC mains or AC distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from AC input or output
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
    • Y02E40/40—Arrangements for reducing harmonics
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E60/00—Enabling technologies; Technologies with a potential or indirect contribution to GHG emissions mitigation
    • Y02E60/60—Arrangements for transfer of electric power between AC networks or generators via a high voltage DC link [HVCD]

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  • Power Engineering (AREA)
  • Control Of Electrical Variables (AREA)
  • Dc-Dc Converters (AREA)

Abstract

一种三电平NPC变换器的滑模控制方法，属于电力电子控制技术领域，解决了三电平NPC变换器采用现有基于观测器的滑模控制存在抖振大，对测量噪声十分敏感的问题。本发明采用直流电压调节环，获取当前采样点时刻负载的有功功率参考值p*；采用瞬时功率跟踪环，获取三电平NPC变换器平均占空比号δαβ；采用电压平衡环，获得平衡占空比δba；对平衡占空比δba和电平NPC变换器平均占空比δ′abc相加后，通过脉冲宽度调制器，获得三电平NPC变换器开关管控制信号，实现对三电平NPC变换器控制。本发明适用于三电平NPC变换器的控制。

Figure 202110692178

A sliding mode control method for a three-level NPC converter belongs to the technical field of power electronic control, and solves the problem that the three-level NPC converter adopts the existing observer-based sliding mode control, which has large chattering and is very sensitive to measurement noise. question. The present invention adopts the DC voltage regulation loop to obtain the active power reference value p * of the load at the current sampling point; adopts the instantaneous power tracking loop to obtain the average duty cycle number δαβ of the three-level NPC converter; adopts the voltage balance loop to obtain a balanced Duty ratio δ ba ; after adding the balance duty ratio δ ba and the average duty ratio δ′ abc of the level NPC converter, the control signal of the switch tube of the three-level NPC converter is obtained through the pulse width modulator, and the control signal of the three-level NPC converter is obtained. Three-level NPC inverter control. The invention is suitable for the control of the three-level NPC converter.

Figure 202110692178

Description

一种三电平NPC变换器的滑模控制方法A sliding mode control method for a three-level NPC converter

技术领域technical field

本发明属于电力电子控制技术领域，具体涉及一种三电平NPC变换器的滑模控制方法。The invention belongs to the technical field of power electronic control, and in particular relates to a sliding mode control method of a three-level NPC converter.

背景技术Background technique

在过去的几十年中，，许多种多电平功率变换器拓扑被提出以满足中高压应用的需求。三电平中性点钳位(NPC)功率变换器作为一种高性能和低损耗的多电平功率变换器，于1979年被首次提出。与传统的两电平变换器相比，NPC功率变换器具有更高的电压等级和更好的输出电压波形等优势。目前它已经成熟的应用于有源前端或变速驱动等几种中高压工业应用中。例如，直流微电网(MG)，光伏发电，风力涡轮机，电机驱动和能量存储系统等。Over the past few decades, many multilevel power converter topologies have been proposed to meet the needs of medium and high voltage applications. The three-level neutral-point clamped (NPC) power converter was first proposed in 1979 as a high-performance and low-loss multi-level power converter. Compared with the traditional two-level converter, the NPC power converter has the advantages of higher voltage level and better output voltage waveform. At present, it has been maturely used in several medium and high voltage industrial applications such as active front end or variable speed drive. For example, DC microgrids (MGs), photovoltaics, wind turbines, motor drives and energy storage systems, etc.

目前，这些应用中的NPC变换器使用的控制方法有许多，包括传统的PI控制和近期提出的基于观测器的滑模控制等。虽然它们都可以实现基本的控制目标，但是仍然存在一些缺点，可以总结为：Currently, many control methods are used for NPC converters in these applications, including traditional PI control and recently proposed observer-based sliding mode control. While they can all achieve basic control goals, there are still some disadvantages, which can be summarized as:

(1)使用PI控制，虽然可以保证NPC的正常运行，但是它的动态和稳态性能却无法保证；此外，当出现外部干扰时，例如加载的情况，PI控制器无法迅速的抑制扰动，导致直流母线电压出现较大的过冲，对整个系统造成破坏，抗干扰性较差。(1) Using PI control, although the normal operation of NPC can be guaranteed, its dynamic and steady-state performance cannot be guaranteed; in addition, when there is external disturbance, such as loading, the PI controller cannot quickly suppress the disturbance, resulting in The DC bus voltage has a large overshoot, causing damage to the entire system and poor anti-interference.

(2)使用基于观测器的滑模控制，虽然可以提升动态性能，但是会产生较大的抖振现象，对系统造成不良的影响。此外，扰动观测器虽然可以提升系统的抗干扰能力，但是它对测量噪声十分敏感，从而限制了它的性能。(2) Using the observer-based sliding mode control, although the dynamic performance can be improved, it will produce a large chattering phenomenon, which will have an adverse effect on the system. In addition, the disturbance observer can improve the anti-interference ability of the system, but it is very sensitive to measurement noise, which limits its performance.

因此，综上采用传统的PI控制算法对三电平NPC变换器进行控制的方法，存在抗干扰性能较差，系统稳态性能及动态响应性能差的缺点。而采用基于观测器的滑模控制，存在抖振大，对测量噪声十分敏感的问题。因此，以上问题亟需解决。Therefore, in summary, the traditional PI control algorithm to control the three-level NPC converter has the disadvantages of poor anti-interference performance, poor system steady-state performance and poor dynamic response performance. The use of observer-based sliding mode control has the problem of large chattering and sensitivity to measurement noise. Therefore, the above problems need to be solved urgently.

发明内容SUMMARY OF THE INVENTION

本发明目的是为了解决三电平NPC变换器采用现有基于观测器的滑模控制存在抖振大，对测量噪声十分敏感的问题，提供了一种三电平NPC变换器的滑模控制方法。The purpose of the present invention is to solve the problem that the three-level NPC converter adopts the existing observer-based sliding mode control, which has large chattering and is very sensitive to measurement noise, and provides a sliding mode control method for the three-level NPC converter. .

本发明所述一种三电平NPC变换器的滑模控制方法，该方法为:The sliding mode control method of a kind of three-level NPC converter of the present invention, the method is:

采用直流电压调节环，利用三电平NPC变换器直流侧电压实际值vdc和直流侧电压参考值

Figure BDA0003126557740000011
通过自适应滑模控制器，获取当前采样点时刻直流侧的有功功率参考值p*；The DC voltage regulation loop is adopted, and the actual value of the DC side voltage v dc and the reference value of the DC side voltage of the three-level NPC converter are used.
Figure BDA0003126557740000011
Obtain the active power reference value p * of the DC side at the current sampling point through the adaptive sliding mode controller;

采用瞬时功率跟踪环，利用当前采样点时刻直流侧的有功功率参考值p*、有功功率实际值p、无功功率实际值q和预设的无功功率参考值q*，通过二阶滑模控制器，获取三电平NPC变换器平均占空比号δαβ；Using the instantaneous power tracking loop, using the active power reference value p * , the actual active power value p, the actual reactive power value q and the preset reactive power reference value q * of the DC side at the current sampling point, through the second-order sliding mode The controller obtains the average duty cycle number δ αβ of the three-level NPC converter;

对三电平NPC变换器平均占空比δαβ进行αβ/abc坐标变换，获得abc坐标系下的三电平NPC变换器平均占空比δa'bc；Perform αβ /abc coordinate transformation on the average duty cycle δαβ of the three-level NPC converter to obtain the average duty cycle δa'bc of the three-level NPC converter under the abc coordinate system;

采用电压平衡环，对直流侧不平衡电压实际值edc与直流侧不平衡电压参考值

Figure BDA0003126557740000021
作差，并将二者的差值进行PI调节，获得平衡占空比δba；The voltage balance loop is used to compare the actual value of the unbalanced voltage on the DC side, e dc , and the reference value of the unbalanced voltage on the DC side.
Figure BDA0003126557740000021
Make a difference, and perform PI adjustment on the difference between the two to obtain a balanced duty cycle δ ba ;

对平衡占空比δba和电平NPC变换器平均占空比δa'bc相加后，通过脉冲宽度调制器，获得三电平NPC变换器开关管控制信号，实现对三电平NPC变换器控制。After adding the balance duty ratio δ ba and the average duty ratio δ a ' bc of the level NPC converter, the control signal of the switch tube of the three-level NPC converter is obtained through the pulse width modulator, and the three-level NPC converter is realized. controller control.

进一步地，本发明中，采用直流电压调节环，利用三电平NPC变换器直流侧电压实际值vdc和直流侧电压参考值

Figure BDA0003126557740000022
通过自适应滑模控制器ASMC和非线性高增益观测器NHGO，获取当前采样点时刻负载的有功功率参考值p*的具体方法为：Further, in the present invention, a DC voltage regulation loop is adopted, and the actual DC side voltage v dc and the DC side voltage reference value of the three-level NPC converter are used.
Figure BDA0003126557740000022
Through the adaptive sliding mode controller ASMC and the nonlinear high gain observer NHGO, the specific method to obtain the active power reference value p * of the load at the current sampling point is as follows:

步骤A1、利用三电平NPC变换器直流侧电压实际值vdc和直流侧电压参考值

Figure BDA0003126557740000023
计算直流电压调节环跟踪误差sv；其中，
Figure BDA0003126557740000024
Step A1, using the actual value of the DC side voltage v dc and the reference value of the DC side voltage of the three-level NPC converter
Figure BDA0003126557740000023
Calculate the DC voltage regulation loop tracking error s v ; where,
Figure BDA0003126557740000024

步骤A2、通过自适应滑模控制器ASMC，对直流电压调节环跟踪误差sv进行纠正，并输出纠偏后的跟踪误差

Figure BDA0003126557740000025
Step A2: Correct the tracking error s v of the DC voltage regulation loop through the adaptive sliding mode controller ASMC, and output the tracking error after the correction
Figure BDA0003126557740000025

步骤A3、采用非线性高增益观测器NHGO对步骤A1中的x1和上一采样点时刻有功功率参考值p*对直流负载的有功功率x2进行观测，获得直流侧负载功率的估计值

Figure BDA0003126557740000026
Step A3: Use the nonlinear high-gain observer NHGO to observe the active power x 2 of the DC load at x 1 in step A1 and the active power reference value p * at the last sampling point to obtain an estimated value of the DC side load power
Figure BDA0003126557740000026

步骤A4、对所述纠偏后的跟踪误差

Figure BDA0003126557740000027
和直流侧负载功率的估计值
Figure BDA0003126557740000028
相加，获得当前采样点时刻有功功率参考值p*。Step A4, to the tracking error after the deviation correction
Figure BDA0003126557740000027
and the estimated value of the load power on the DC side
Figure BDA0003126557740000028
Add up to obtain the active power reference value p * at the current sampling point.

进一步地，本发明中，步骤A2中，自适应滑模控制器ASMC的动态方程为：Further, in the present invention, in step A2, the dynamic equation of the adaptive sliding mode controller ASMC is:

Figure BDA0003126557740000029
Figure BDA0003126557740000029

其中，Kv为自适应滑模控制器ASMC的增益，为一个时变量，自适应滑模控制器的自适应率

Figure BDA00031265577400000210
为：Among them, K v is the gain of the adaptive sliding mode controller ASMC, which is a time variable, and the adaptive rate of the adaptive sliding mode controller
Figure BDA00031265577400000210
for:

Figure BDA00031265577400000211
Figure BDA00031265577400000211

其中，Kl是自适应滑模控制器ASMC的增益变化率，Km是自适应滑模控制器ASMC的增益判定参数，ba是自适应滑模控制器ASMC的增益梯度判定参数。Among them, K l is the gain change rate of the adaptive sliding mode controller ASMC, K m is the gain determination parameter of the adaptive sliding mode controller ASMC, and b a is the gain gradient determination parameter of the adaptive sliding mode controller ASMC.

进一步地，本发明中，步骤A3中，非线性高增益观测器NHGO的动态方程为：Further, in the present invention, in step A3, the dynamic equation of the nonlinear high-gain observer NHGO is:

Figure BDA0003126557740000031
Figure BDA0003126557740000031

Figure BDA0003126557740000032
Figure BDA0003126557740000032

其中，C是直流侧电容的电容值，

Figure BDA0003126557740000033
是变量x1的估计值，
Figure BDA0003126557740000034
是变量x1的导数的估计值，
Figure BDA0003126557740000035
是直流负载的有功功率x2的导数的估计值，α1是非线性高增益观测器NHGO的前级增益参数，α2是非线性高增益观测器NHGO的后级增益参数，ε1和ε2是非线性高增益观测器NHGO的两个增益参数，且ε2大于ε1,bs是非线性高增益观测器NHGO的增益判定参数，sat(·)为饱和函数。where C is the capacitance value of the DC side capacitor,
Figure BDA0003126557740000033
is the estimated value of the variable x 1 ,
Figure BDA0003126557740000034
is an estimate of the derivative of the variable x 1 ,
Figure BDA0003126557740000035
is the estimated value of the derivative of the active power x 2 of the DC load, α 1 is the pre-stage gain parameter of the nonlinear high-gain observer NHGO, α 2 is the post-stage gain parameter of the nonlinear high-gain observer NHGO, ε 1 and ε 2 are non-linear high-gain observers. The two gain parameters of the linear high-gain observer NHGO, and ε 2 is greater than ε 1 , b s is the gain decision parameter of the nonlinear high-gain observer NHGO, and sat(·) is a saturation function.

进一步地，本发明中，采用瞬时功率跟踪环，利用当前采样点时刻直流侧的有功功率参考值p*、有功功率实际值p、无功功率实际值q和预设的无功功率参考值q*，通过二阶滑模控制器，获取三电平NPC变换器平均占空比δαβ的具体方法为：Further, in the present invention, an instantaneous power tracking loop is adopted, and the active power reference value p * , the actual active power value p, the actual reactive power value q and the preset reactive power reference value q of the DC side at the current sampling point are used. * , through the second-order sliding mode controller, the specific method to obtain the average duty cycle δ αβ of the three-level NPC converter is:

步骤B1、将当前采样点时刻有功功率参考值p*和有功功率实际值p比较，获得有功功率跟踪误差sp，同时，将当前采样点时刻无功功率实际值q和预设的无功功率参考值q*相比较，获得无功功率跟踪误差sq；Step B1: Compare the active power reference value p * at the current sampling point with the actual active power value p to obtain the active power tracking error sp, and at the same time, compare the actual reactive power value q at the current sampling point with the preset reactive power Compared with the reference value q * , the reactive power tracking error s q is obtained;

步骤B2、通过二阶滑模控制器SOSM对有功功率跟踪误差sp和无功功率跟踪误差sq进行纠正，获得纠偏后的有功功率跟踪误差

Figure BDA0003126557740000036
和纠偏后的无功功率跟踪误差
Figure BDA0003126557740000037
Step B2, correct the active power tracking error s p and the reactive power tracking error s q through the second-order sliding mode controller SOSM, and obtain the active power tracking error after the correction
Figure BDA0003126557740000036
and reactive power tracking error after offset correction
Figure BDA0003126557740000037

步骤B3、根据有功功率跟踪误差sp、无功功率跟踪误差sq、上一采样点时刻有功功率纠正量up和上一采样点时刻无功功率纠正量uq，利用高增益观测器NHGO对由系统不确定参数而引起的内部扰动lp和lq进行观测，获得内部扰动的估计值

Figure BDA0003126557740000038
和
Figure BDA0003126557740000039
其中，有功功率纠正量up和无功功率纠正量uq的初始值为0；Step B3: According to the active power tracking error sp, the reactive power tracking error s q , the active power correction amount up at the last sampling point, and the reactive power correction amount u q at the last sampling point, use the high-gain observer NHGO Observe the internal disturbances l p and l q caused by the uncertain parameters of the system, and obtain the estimated value of the internal disturbance
Figure BDA0003126557740000038
and
Figure BDA0003126557740000039
Among them, the initial values of the active power correction amount u p and the reactive power correction amount u q are 0;

步骤B4、根据纠偏后的跟踪误差

Figure BDA00031265577400000310
和
Figure BDA00031265577400000311
与内部扰动的估计值
Figure BDA00031265577400000312
和
Figure BDA00031265577400000313
更新当前采样点时刻有功功率纠正量up和无功功率纠正量uq；Step B4, according to the tracking error after correction
Figure BDA00031265577400000310
and
Figure BDA00031265577400000311
with estimates of internal disturbances
Figure BDA00031265577400000312
and
Figure BDA00031265577400000313
Update the active power correction amount u p and the reactive power correction amount u q at the current sampling point;

步骤B5、对有功功率实际值p和无功功率实际值q求导，分别获得有功功率导数

Figure BDA00031265577400000314
和无功功率导数
Figure BDA00031265577400000315
令
Figure BDA00031265577400000316
获得等效点的平均占空比
Figure BDA00031265577400000317
Step B5, derive the actual value p of the active power and the actual value q of the reactive power, and obtain the derivatives of the active power respectively
Figure BDA00031265577400000314
and reactive power derivative
Figure BDA00031265577400000315
make
Figure BDA00031265577400000316
Obtain the average duty cycle of the equivalent point
Figure BDA00031265577400000317

步骤B6、根据有功功率纠正量up、无功功率纠正量uq和等效点的平均占空比

Figure BDA00031265577400000318
获得平均占空比δαβ。Step B6, according to the active power correction amount u p , the reactive power correction amount u q and the average duty cycle of the equivalent point
Figure BDA00031265577400000318
The average duty cycle δαβ is obtained.

进一步地，本发明中，步骤B2中，二阶滑模控制器SOSM的动态方程为：Further, in the present invention, in step B2, the dynamic equation of the second-order sliding mode controller SOSM is:

Figure BDA0003126557740000041
Figure BDA0003126557740000041

其中，ki1和ki2是二阶滑模控制器SOSM的增益,t为时间。Among them, k i1 and k i2 are the gains of the second-order sliding mode controller SOSM, and t is the time.

进一步地，本发明中，步骤B3中，获得内部扰动的估计值

Figure BDA0003126557740000042
和
Figure BDA0003126557740000043
通过公式：Further, in the present invention, in step B3, the estimated value of the internal disturbance is obtained
Figure BDA0003126557740000042
and
Figure BDA0003126557740000043
Via the formula:

Figure BDA0003126557740000044
Figure BDA0003126557740000044

Figure BDA0003126557740000045
Figure BDA0003126557740000045

和and

Figure BDA0003126557740000046
Figure BDA0003126557740000046

Figure BDA0003126557740000047
Figure BDA0003126557740000047

计算实现，其中，

Figure BDA0003126557740000048
其中，vα和vβ为αβ坐标系下的变换器交流侧电压的α分量和β分量，L为交流侧线电感，α3是有功功率环路高增益观测器的前级增益参数，α4是有功功率环路高增益观测器的后级增益参数，α5是无功功率环路高增益观测器的前级增益参数，α6是无功功率环路高增益观测器的后级增益参数，εp是有功功率环路高增益观测器的增益，εq是无功功率环路高增益观测器的增益。Computational realization, where,
Figure BDA0003126557740000048
Among them, v α and v β are the α and β components of the AC side voltage of the converter in the αβ coordinate system, L is the AC side line inductance, α 3 is the front stage gain parameter of the active power loop high gain observer, α 4 is the post-stage gain parameter of the active power loop high-gain observer, α 5 is the pre-stage gain parameter of the reactive power loop high-gain observer, and α 6 is the post-stage gain parameter of the reactive power loop high-gain observer , ε p is the gain of the active power loop high gain observer, ε q is the gain of the reactive power loop high gain observer.

进一步地，本发明中，步骤B4中，当前采样点时刻有功功率纠正量up和无功功率纠正量uq为：Further, in the present invention, in step B4, the active power correction amount u p and the reactive power correction amount u q at the current sampling point are:

Figure BDA0003126557740000049
Figure BDA0003126557740000049

其中，

Figure BDA00031265577400000410
in,
Figure BDA00031265577400000410

进一步地，本发明中，步骤B5中，等效点的平均占空比

Figure BDA00031265577400000411
为：Further, in the present invention, in step B5, the average duty cycle of the equivalent point
Figure BDA00031265577400000411
for:

Figure BDA00031265577400000412
Figure BDA00031265577400000412

其中，vαβ为αβ坐标系下的变换器交流侧电压；J为矩阵，且

Figure BDA00031265577400000413
ω为电网电压的角频率；L为交流侧线电感。Among them, v αβ is the AC side voltage of the converter in the αβ coordinate system; J is a matrix, and
Figure BDA00031265577400000413
ω is the angular frequency of the grid voltage; L is the inductance of the AC side line.

进一步地，本发明中，采用电压平衡环，对直流侧不平衡电压实际值edc与直流侧不平衡电压参考值

Figure BDA0003126557740000051
作差，并将二者的差值进行PI调节，获得平衡占空比δba为:Further, in the present invention, a voltage balance loop is used to compare the actual value edc of the unbalanced voltage on the DC side and the reference value of the unbalanced voltage on the DC side
Figure BDA0003126557740000051
Make a difference, and adjust the difference between the two by PI to obtain a balanced duty cycle δba as:

Figure BDA0003126557740000052
Figure BDA0003126557740000052

其中，kpb为PI控制器的比例环节增益；kib为PI控制器的积分环节增益；t为时间。Among them, k pb is the proportional link gain of the PI controller; k ib is the integral link gain of the PI controller; t is the time.

本发明方法提高了三相NPC变换器的动态和稳态性能以及抗干扰能力。本发明所述的一种三电平NPC变换器的滑模控制方法，该控制方法是基于直流电压调节环、瞬时功率跟踪环和电压平衡环实现。通过直流电压调节环对直流侧电压实际值vdc进行控制，使直流侧电容器电压总和vdc调节到对应的期望值

Figure BDA0003126557740000053
通过瞬时功率跟踪环对有功功率实际值p和无功功率实际值q进行控制，使有功功率p和无功功率q准确的跟踪各自的参考值p*和q*、以及通过电压平衡环对直流侧不平衡电压实际值edc进行控制，确保直流侧两个电容的不平衡电压接近于0，结合三个环的共同作用生成控制信号对三电平NPC变换器进行控制，控制过程简单，通过直流电压调节环来调节直流母线电压，以确保电压阶跃阶段的快速动态响应，以及不确定干扰的直流侧负载接入电路时会引起直流侧电压的波动都能有效抑制。实现直流侧电容器的电压平衡，提高了控制稳定性。The method of the invention improves the dynamic and steady-state performance and the anti-interference ability of the three-phase NPC converter. The sliding mode control method of a three-level NPC converter according to the present invention is realized based on a DC voltage regulation loop, an instantaneous power tracking loop and a voltage balance loop. The actual value v dc of the DC side voltage is controlled by the DC voltage regulating loop, so that the sum of the voltages of the DC side capacitors v dc is adjusted to the corresponding desired value
Figure BDA0003126557740000053
The actual value p and the actual value q of the active power are controlled by the instantaneous power tracking loop, so that the active power p and the reactive power q can accurately track their respective reference values p * and q * , and the DC voltage is controlled by the voltage balance loop. The actual value of the side unbalanced voltage e dc is controlled to ensure that the unbalanced voltage of the two capacitors on the DC side is close to 0. Combined with the joint action of the three loops, a control signal is generated to control the three-level NPC converter. The control process is simple. The DC voltage regulation loop is used to adjust the DC bus voltage to ensure the fast dynamic response in the voltage step stage, and the fluctuation of the DC side voltage caused by the DC side load of uncertain interference can be effectively suppressed when connected to the circuit. The voltage balance of the DC side capacitors is realized, and the control stability is improved.

在直流电压调节环中，采用了自适应滑模控制器(ASMC)来快速地调节直流母线电压，不仅减小了抖振还确保了电压阶跃阶段的快速动态响应。同时，由于外部不确定干扰的存在，因此基于自适应滑模控制器(ASMC)(直流电压调节环)加入了一种非线性高增益观测器(NHGO)来抑制它(外部不确定干扰的存在)，该观测器对噪声不敏感，且具备强大的抗扰动能力；在瞬时功率跟踪环中，采用了一种简单有效的直接功率控制(DPC)策略来实现功率跟踪的目标，从而简化了内部环路的控制过程。另外，为了获得具有低谐波失真的交流电流以及对系统参数摄动的鲁棒性，采用了基于高增益观测器(HGO)的二阶滑模控制器(SOSM)以确保有功功率和无功功率可以迅速收敛到稳定状态；最后，在电压平衡环路中，使用PI调节器来确保直流侧电容器的电压平衡。通过实验测试，将本发明所提出的NPC功率变换器控制策略与其他控制方案进行了比较，证明了本方案的有效性和优越性。In the DC voltage regulation loop, an adaptive sliding mode controller (ASMC) is used to quickly regulate the DC bus voltage, which not only reduces chattering but also ensures fast dynamic response in the voltage step phase. At the same time, due to the existence of external uncertain interference, a nonlinear high-gain observer (NHGO) is added based on adaptive sliding mode controller (ASMC) (DC voltage regulation loop) to suppress it (existence of external uncertain interference). ), the observer is insensitive to noise and has strong anti-disturbance capability; in the instantaneous power tracking loop, a simple and effective direct power control (DPC) strategy is adopted to achieve the goal of power tracking, which simplifies the internal loop control process. In addition, in order to obtain AC current with low harmonic distortion and robustness to system parameter perturbation, a high-gain observer (HGO) based second-order sliding mode controller (SOSM) is employed to ensure active and reactive power The power can quickly converge to a steady state; finally, in the voltage balancing loop, a PI regulator is used to ensure the voltage balance of the DC link capacitors. Through experimental tests, the control strategy of the NPC power converter proposed by the present invention is compared with other control schemes, which proves the effectiveness and superiority of the scheme.

附图说明Description of drawings

图1为本发明所述的三电平NPC变换器与交流电网和直流微网之间连接的电路原理示意图；1 is a schematic diagram of the circuit principle of the connection between the three-level NPC converter according to the present invention, an alternating current grid and a direct current microgrid;

图2为本发明所述的生成控制信号的原理示意；2 is a schematic diagram of the principle of generating a control signal according to the present invention;

图3为电压参考值从750V调整至690V时，三电平NPC变换器的直流侧电压波形图；其中，Figure 3 is a waveform diagram of the DC side voltage of the three-level NPC converter when the voltage reference value is adjusted from 750V to 690V; among them,

图3a为PI控制器下的直流侧电压波形图；Figure 3a is a waveform diagram of the DC side voltage under the PI controller;

图3b为基于扩张状态观测器的滑模控制策略(LESO-SMC)下的直流侧电压波形图；Figure 3b is the DC side voltage waveform diagram under the sliding mode control strategy (LESO-SMC) based on the extended state observer;

图3c为基于非线性高增益观测器的滑模控制策略(NHGO-SMC)下的直流侧电压波形图；Figure 3c is the DC side voltage waveform diagram under the sliding mode control strategy (NHGO-SMC) based on the nonlinear high gain observer;

图3d为采用本发明控制方法下的直流侧电压波形图；Fig. 3d is a DC side voltage waveform diagram under the control method of the present invention;

图4为电压参考值从690V调整至750V时，三电平NPC变换器的直流侧电压波形图；其中，Figure 4 is a waveform diagram of the DC side voltage of the three-level NPC converter when the voltage reference value is adjusted from 690V to 750V; among them,

图4a为PI控制器下的直流侧电压波形图；Figure 4a is a waveform diagram of the DC side voltage under the PI controller;

图4b为基于扩张状态观测器的滑模控制策略(LESO-SMC)下的直流侧电压波形图；Fig. 4b is the DC side voltage waveform diagram under the sliding mode control strategy (LESO-SMC) based on the extended state observer;

图4c为基于非线性高增益观测器的滑模控制策略(NHGO-SMC)下的直流侧电压波形图；Figure 4c is the DC side voltage waveform diagram under the sliding mode control strategy (NHGO-SMC) based on the nonlinear high gain observer;

图4d为采用本发明控制方法下的直流侧电压波形图；Fig. 4d is a DC side voltage waveform diagram under the control method of the present invention;

图5a为接入负载R1和R2稳定运行时，PI控制器下的三相交流电流谐波频谱图；Figure 5a is the harmonic spectrum diagram of the three-phase AC current under the PI controller when the loads R 1 and R 2 are connected to run stably;

图5b为接入负载R1和R2稳定运行时，基于扩张状态观测器的滑模控制策略(LESO-SMC)下的三相交流电流谐波频谱图；Figure 5b is a three-phase AC current harmonic spectrum diagram under the sliding mode control strategy (LESO-SMC) based on the extended state observer when the connected loads R 1 and R 2 are running stably;

图5c为接入负载R1和R2稳定运行时，基于非线性高增益观测器的滑模控制策略(NHGO-SMC)下的三相交流电流谐波频谱图；Figure 5c shows the harmonic spectrum of the three-phase AC current under the sliding mode control strategy (NHGO-SMC) based on the nonlinear high-gain observer when the loads R 1 and R 2 are connected to operate stably;

图5d为接入负载R1和R2稳定运行时，采用本发明控制方法下的三相交流电流谐波频谱图；Fig. 5d is a three-phase AC current harmonic spectrum diagram under the control method of the present invention when the connected loads R 1 and R 2 operate stably;

图6a为PI控制器下的直流母线电压和a相电流的瞬态响应；Figure 6a shows the transient response of the DC bus voltage and a-phase current under the PI controller;

图6b为基于扩张状态观测器的滑模控制策略(LESO-SMC)下的直流母线电压和a相电流的瞬态响应；Figure 6b shows the transient responses of the DC bus voltage and a-phase current under the sliding mode control strategy based on the extended state observer (LESO-SMC);

图6c为基于非线性高增益观测器的滑模控制策略(NHGO-SMC)下的直流母线电压和a相电流的瞬态响应；Figure 6c shows the transient responses of the DC bus voltage and a-phase current under the sliding mode control strategy (NHGO-SMC) based on a nonlinear high-gain observer;

图6d为采用本发明控制方法下的直流母线电压和a相电流的瞬态响应。FIG. 6d is the transient response of the DC bus voltage and the a-phase current under the control method of the present invention.

具体实施方式Detailed ways

下面将结合本发明实施例中的附图，对本发明实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例仅仅是本发明一部分实施例，而不是全部的实施例。基于本发明中的实施例，本领域普通技术人员在没有作出创造性劳动的前提下所获得的所有其他实施例，都属于本发明保护的范围。The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, but not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those of ordinary skill in the art without creative work fall within the protection scope of the present invention.

需要说明的是，在不冲突的情况下，本发明中的实施例及实施例中的特征可以相互组合。It should be noted that the embodiments of the present invention and the features of the embodiments may be combined with each other under the condition of no conflict.

具体实施方式一：下面结合图1说明本实施方式，本实施方式所述一种三电平NPC变换器的滑模控制方法，该方法为:Specific embodiment one: the present embodiment is described below in conjunction with Fig. 1, the sliding mode control method of a kind of three-level NPC converter described in the present embodiment, the method is:

采用直流电压调节环，利用三电平NPC变换器直流侧电压实际值vdc和直流侧电压参考值

Figure BDA0003126557740000071
通过自适应滑模控制器，获取当前采样点时刻直流侧的有功功率参考值p*；The DC voltage regulation loop is adopted, and the actual value of the DC side voltage v dc and the reference value of the DC side voltage of the three-level NPC converter are used.
Figure BDA0003126557740000071
Obtain the active power reference value p * of the DC side at the current sampling point through the adaptive sliding mode controller;

采用瞬时功率跟踪环，利用当前采样点时刻直流侧的有功功率参考值p*、有功功率实际值p、无功功率实际值q和预设的无功功率参考值q*，通过二阶滑模控制器，获取三电平NPC变换器平均占空比号δαβ；Using the instantaneous power tracking loop, using the active power reference value p * , the actual active power value p, the actual reactive power value q and the preset reactive power reference value q * of the DC side at the current sampling point, through the second-order sliding mode The controller obtains the average duty cycle number δ αβ of the three-level NPC converter;

对三电平NPC变换器平均占空比δαβ进行αβ/abc坐标变换，获得abc坐标系下的三电平NPC变换器平均占空比δa'bc；Perform αβ /abc coordinate transformation on the average duty cycle δαβ of the three-level NPC converter to obtain the average duty cycle δa'bc of the three-level NPC converter under the abc coordinate system;

采用电压平衡环，对直流侧不平衡电压实际值edc与直流侧不平衡电压参考值

Figure BDA0003126557740000072
作差，并将二者的差值进行PI调节，获得平衡占空比δba；The voltage balance loop is used to compare the actual value of the unbalanced voltage on the DC side, e dc , and the reference value of the unbalanced voltage on the DC side.
Figure BDA0003126557740000072
Make a difference, and perform PI adjustment on the difference between the two to obtain a balanced duty cycle δ ba ;

对平衡占空比δba和电平NPC变换器平均占空比δa'bc相加后，通过脉冲宽度调制器，获得三电平NPC变换器开关管控制信号，实现对三电平NPC变换器控制。After adding the balance duty ratio δ ba and the average duty ratio δ a ' bc of the level NPC converter, the control signal of the switch tube of the three-level NPC converter is obtained through the pulse width modulator, and the three-level NPC converter is realized. controller control.

具体应用之前，先根据三电平NPC变换器的运行原理，建立三电平NPC变换器的状态空间平均模型；根据三电平NPC变换器的状态空间平均模型，确定三电平NPC变换器的控制目标；所述控制目标包括：使直流侧两个电容器电压总和vdc调节到直流侧电压参考值期望值

Figure BDA0003126557740000073
使有功功率p和无功功率q始终跟踪各自的参考值p*和q*、以及确保直流侧两个电容器的不平衡电压趋近于0，来生成相应控制信号对三电平NPC变换器进行控制。Before the specific application, according to the operation principle of the three-level NPC converter, establish the state space average model of the three-level NPC converter; according to the state space average model of the three-level NPC converter, determine the three-level NPC converter A control objective; the control objective includes: adjusting the sum of the voltages of the two capacitors on the DC side v dc to a desired value of the DC side voltage reference value
Figure BDA0003126557740000073
Make the active power p and reactive power q always track their respective reference values p * and q * , and ensure that the unbalanced voltage of the two capacitors on the DC side is close to 0, so as to generate corresponding control signals for the three-level NPC converter. control.

实际上，采用一种高效的控制法是十分有必要的。它可以确保系统达到稳定状态时，有功功率和无功功率保持在等效点附近，并且保证较高的电流质量。本发明不仅可以实现不同的控制目标，而且还可以改善三电平NPC变换器的动态、稳态性能以及抗干扰能力。In fact, it is necessary to adopt an efficient control method. It can ensure that when the system reaches a steady state, the active power and reactive power are kept near the equivalent point, and high current quality is guaranteed. The invention can not only achieve different control objectives, but also improve the dynamic, steady-state performance and anti-interference ability of the three-level NPC converter.

图1中，三相交流电源和电感器与三电平NPC变换器的交流侧相连，提供功率传输。在直流侧，三电平NPC变换器连接了两个电容器，以存储能量并稳定直流电压。此处，直流侧可以看作是直流微网，它主要由直流负载，其他变换器和各种可再生能源等组成。In Figure 1, a three-phase AC source and an inductor are connected to the AC side of a three-level NPC converter to provide power transfer. On the DC side, two capacitors are connected to the three-level NPC converter to store energy and stabilize the DC voltage. Here, the DC side can be regarded as a DC microgrid, which is mainly composed of DC loads, other converters and various renewable energy sources.

本实施方式所述的本发明方法提高了三相NPC变换器的动态和稳态性能以及抗干扰能力。本发明所述的一种三电平NPC变换器的滑模控制方法，该控制方法是基于直流电压调节环、瞬时功率跟踪环和电压平衡环实现，通过直流电压调节环对直流侧电压实际值vdc进行控制，使直流侧电容器电压总和vdc调节到对应的期望值

Figure BDA0003126557740000081
通过瞬时功率跟踪环对有功功率实际值p和无功功率实际值q进行跟踪，使有功功率p和无功功率q准确的跟踪各自的参考值p*和q*、以及通过电压平衡环对直流侧不平衡电压实际值edc进行控制，确保直流侧两个电容的不平衡电压接近于0，结合三个环的共同作用生成控制信号对三电平NPC变换器进行控制，控制过程简单，通过直流电压调节环来调节直流母线电压，以确保电压阶跃阶段的快速动态响应，以及不确定干扰的直流侧负载接入电路时会引起直流侧电压的波动都能有效抑制。实现直流侧电容器的电压平衡，提高了控制稳定性。The method of the present invention described in this embodiment improves the dynamic and steady-state performance and anti-interference ability of the three-phase NPC converter. The sliding mode control method of a three-level NPC converter according to the present invention is realized based on a DC voltage regulation loop, an instantaneous power tracking loop and a voltage balance loop. v dc is controlled so that the sum of the DC side capacitor voltage v dc is adjusted to the corresponding desired value
Figure BDA0003126557740000081
The actual value p and the actual value q of the active power are tracked through the instantaneous power tracking loop, so that the active power p and the reactive power q can accurately track their respective reference values p * and q * , and the DC voltage is adjusted through the voltage balance loop. The actual value of the side unbalanced voltage e dc is controlled to ensure that the unbalanced voltage of the two capacitors on the DC side is close to 0. Combined with the joint action of the three loops, a control signal is generated to control the three-level NPC converter. The control process is simple. The DC voltage regulation loop is used to adjust the DC bus voltage to ensure the fast dynamic response in the voltage step stage, and the fluctuation of the DC side voltage caused by the DC side load of uncertain interference can be effectively suppressed when connected to the circuit. The voltage balance of the DC side capacitors is realized, and the control stability is improved.

进一步地，本实施方式中，Further, in this embodiment,

采用直流电压调节环，利用三电平NPC变换器直流侧电压实际值vdc和直流侧电压参考值

Figure BDA0003126557740000082
通过自适应滑模控制器ASMC和非线性高增益观测器NHGO，获取当前采样点时刻直流侧的有功功率参考值p*的具体方法为：The DC voltage regulation loop is adopted, and the actual value of the DC side voltage v dc and the reference value of the DC side voltage of the three-level NPC converter are used.
Figure BDA0003126557740000082
Through the adaptive sliding mode controller ASMC and the nonlinear high gain observer NHGO, the specific method to obtain the active power reference value p * of the DC side at the current sampling point is as follows:

步骤A1、利用三电平NPC变换器直流侧电压实际值vdc和直流侧电压参考值

Figure BDA0003126557740000083
计算直流电压调节环跟踪误差sv；其中，
Figure BDA0003126557740000084
Step A1, using the actual value of the DC side voltage v dc and the reference value of the DC side voltage of the three-level NPC converter
Figure BDA0003126557740000083
Calculate the DC voltage regulation loop tracking error s v ; where,
Figure BDA0003126557740000084

步骤A2、通过自适应滑模控制器ASMC，对直流电压调节环跟踪误差sv进行纠正，并输出纠偏后的跟踪误差

Figure BDA0003126557740000085
Step A2: Correct the tracking error s v of the DC voltage regulation loop through the adaptive sliding mode controller ASMC, and output the tracking error after the correction
Figure BDA0003126557740000085

步骤A3、采用非线性高增益观测器NHGO对步骤A1中的x1和上一采样点时刻有功功率参考值p*对直流负载的有功功率x2进行观测，获得直流侧负载功率的估计值

Figure BDA0003126557740000086
Step A3: Use the nonlinear high-gain observer NHGO to observe the active power x 2 of the DC load at x 1 in step A1 and the active power reference value p * at the last sampling point to obtain an estimated value of the DC side load power
Figure BDA0003126557740000086

步骤A4、对所述纠偏后的跟踪误差

Figure BDA0003126557740000087
和直流侧负载功率的估计值
Figure BDA0003126557740000088
相加，获得当前采样点时刻有功功率参考值p*。Step A4, to the tracking error after the deviation correction
Figure BDA0003126557740000087
and the estimated value of the load power on the DC side
Figure BDA0003126557740000088
Add up to obtain the active power reference value p * at the current sampling point.

进一步地，本实施方式中，Further, in this embodiment,

步骤A2中，自适应滑模控制器ASMC的动态方程为：

Figure BDA0003126557740000089
In step A2, the dynamic equation of the adaptive sliding mode controller ASMC is:
Figure BDA0003126557740000089

其中，Kv为自适应滑模控制器ASMC的增益，为一个时变量，自适应滑模控制器的自适应率

Figure BDA00031265577400000810
为：Among them, K v is the gain of the adaptive sliding mode controller ASMC, which is a time variable, and the adaptive rate of the adaptive sliding mode controller
Figure BDA00031265577400000810
for:

Figure BDA0003126557740000091
Figure BDA0003126557740000091

其中，Kl是自适应滑模控制器ASMC的增益变化率，Km是自适应滑模控制器ASMC的增益判定参数，ba是自适应滑模控制器ASMC的增益梯度判定参数。进一步地，本发明中，Among them, K l is the gain change rate of the adaptive sliding mode controller ASMC, K m is the gain determination parameter of the adaptive sliding mode controller ASMC, and b a is the gain gradient determination parameter of the adaptive sliding mode controller ASMC. Further, in the present invention,

步骤A3中，非线性高增益观测器NHGO的动态方程为：In step A3, the dynamic equation of the nonlinear high-gain observer NHGO is:

Figure BDA0003126557740000092
Figure BDA0003126557740000092

Figure BDA0003126557740000093
Figure BDA0003126557740000093

其中，C是直流侧电容的电容值，

Figure BDA0003126557740000094
是变量x1的估计值，
Figure BDA0003126557740000095
是变量x1的导数的估计值，
Figure BDA0003126557740000096
是直流负载的有功功率x2的导数的估计值，α1是非线性高增益观测器NHGO的前级增益参数，α2是非线性高增益观测器NHGO的后级增益参数，ε1和ε2是非线性高增益观测器NHGO的两个增益参数，且ε2大于ε1,bs是非线性高增益观测器NHGO的增益判定参数，sat(·)为饱和函数。where C is the capacitance value of the DC side capacitor,
Figure BDA0003126557740000094
is the estimated value of the variable x 1 ,
Figure BDA0003126557740000095
is an estimate of the derivative of the variable x 1 ,
Figure BDA0003126557740000096
is the estimated value of the derivative of the active power x 2 of the DC load, α 1 is the pre-stage gain parameter of the nonlinear high-gain observer NHGO, α 2 is the post-stage gain parameter of the nonlinear high-gain observer NHGO, ε 1 and ε 2 are non-linear high-gain observers. The two gain parameters of the linear high-gain observer NHGO, and ε 2 is greater than ε 1 , b s is the gain decision parameter of the nonlinear high-gain observer NHGO, and sat(·) is a saturation function.

实施方式中，直流电压调节环基本的控制目标是将直流母线电压调节到指定值。为了获得快速的瞬态响应和对外部干扰不敏感等性能，针对电压调节环提出了基于NHGO的ASMC控制策略。ASMC的英文全称为Adaptive sliding mode control，自适应滑模；NHGO的英文全称为Nonlinear high-gain observer，非线性高增益观测器。In the embodiment, the basic control objective of the DC voltage regulation loop is to regulate the DC bus voltage to a specified value. In order to obtain fast transient response and insensitivity to external disturbances, an NHGO-based ASMC control strategy is proposed for the voltage regulation loop. The English full name of ASMC is Adaptive sliding mode control, adaptive sliding mode; the English full name of NHGO is Nonlinear high-gain observer, nonlinear high gain observer.

自适应滑模控制器ASMC的控制策略，克服了传统滑模控制(SMC)存在的抖振和动态性能的折衷问题，在保留了传统滑模控制的快速动态性能的同时，还减弱了抖振现象，因此将其应用于电力电子领域可以进一步提升变换器的性能。另一方面，尽管ASMC可以提高了系统的鲁棒性，但由于缺少干扰信息，其实现干扰消除的能力还不够，这意味着无法立即将干扰补偿给控制器。作为一项观测状态和扰动的技术，观测器适合于弥补系统的这一缺点，例如滑模观测器(SMO)和线性扩展状态观测器(LESO)。但是传统的观测器都存在着对噪声十分敏感的缺点，因此在实际应用中性能受限。非线性高增益观测器(NHGO)作为高增益观测器的改进版本，不仅对噪声不敏感，而且具备强大的抗扰动能力，十分适合应用于电力电子领域。故本实施方式中采用非线性高增益观测器NHGO实现消除干扰。The control strategy of the adaptive sliding mode controller (ASMC) overcomes the trade-off between chattering and dynamic performance of traditional sliding mode control (SMC), and reduces chattering while retaining the fast dynamic performance of traditional sliding mode control. Therefore, its application in the field of power electronics can further improve the performance of the converter. On the other hand, although ASMC can improve the robustness of the system, its ability to achieve interference cancellation is not enough due to the lack of interference information, which means that the interference cannot be compensated to the controller immediately. As a technique for observing states and disturbances, observers are suitable to compensate for this shortcoming of the system, such as sliding mode observer (SMO) and linear extended state observer (LESO). However, traditional observers all have the disadvantage of being very sensitive to noise, so their performance is limited in practical applications. As an improved version of high gain observer, nonlinear high gain observer (NHGO) is not only insensitive to noise, but also has strong anti-disturbance ability, which is very suitable for application in the field of power electronics. Therefore, in this embodiment, a nonlinear high-gain observer NHGO is used to eliminate interference.

进一步地，本实施方式中，采用瞬时功率跟踪环，利用当前采样点时刻直流侧的有功功率参考值p*、有功功率实际值p、无功功率实际值q和预设的无功功率参考值q*，通过二阶滑模控制器，获取三电平NPC变换器平均占空比号δαβ的具体方法为:Further, in this embodiment, an instantaneous power tracking loop is adopted, and the active power reference value p * , the actual active power value p, the actual reactive power value q and the preset reactive power reference value of the DC side at the current sampling point are used. q * , through the second-order sliding mode controller, the specific method to obtain the average duty cycle number δ αβ of the three-level NPC converter is:

步骤B1、将当前采样点时刻有功功率参考值p*和有功功率实际值p比较，获得有功功率跟踪误差sp，同时，将当前采样点时刻无功功率实际值q和预设的无功功率参考值q*相比较，获得无功功率跟踪误差sq；Step B1: Compare the active power reference value p * at the current sampling point with the actual active power value p to obtain the active power tracking error sp, and at the same time, compare the actual reactive power value q at the current sampling point with the preset reactive power Compared with the reference value q * , the reactive power tracking error s q is obtained;

步骤B2、通过二阶滑模控制器SOSM对有功功率跟踪误差sp和无功功率跟踪误差sq进行纠正，获得纠偏后的有功功率跟踪误差

Figure BDA0003126557740000101
和纠偏后的无功功率跟踪误差
Figure BDA0003126557740000102
Step B2, correct the active power tracking error s p and the reactive power tracking error s q through the second-order sliding mode controller SOSM, and obtain the active power tracking error after the correction
Figure BDA0003126557740000101
and reactive power tracking error after offset correction
Figure BDA0003126557740000102

步骤B3、根据有功功率跟踪误差sp、无功功率跟踪误差sq、上一采样点时刻有功功率纠正量up和上一采样点时刻无功功率纠正量uq，利用高增益观测器NHGO对由系统不确定参数而引起的内部扰动lp和lq进行观测，获得内部扰动的估计值

Figure BDA0003126557740000103
和
Figure BDA0003126557740000104
其中，有功功率纠正量up和无功功率纠正量uq的初始值为0；Step B3: According to the active power tracking error sp, the reactive power tracking error s q , the active power correction amount up at the last sampling point, and the reactive power correction amount u q at the last sampling point, use the high-gain observer NHGO Observe the internal disturbances l p and l q caused by the uncertain parameters of the system, and obtain the estimated value of the internal disturbance
Figure BDA0003126557740000103
and
Figure BDA0003126557740000104
Among them, the initial values of the active power correction amount u p and the reactive power correction amount u q are 0;

步骤B4、根据纠偏后的跟踪误差

Figure BDA0003126557740000105
和
Figure BDA0003126557740000106
与内部扰动的估计值
Figure BDA0003126557740000107
和
Figure BDA0003126557740000108
更新当前采样点时刻有功功率纠正量up和无功功率纠正量uq；Step B4, according to the tracking error after correction
Figure BDA0003126557740000105
and
Figure BDA0003126557740000106
with estimates of internal disturbances
Figure BDA0003126557740000107
and
Figure BDA0003126557740000108
Update the active power correction amount u p and the reactive power correction amount u q at the current sampling point;

步骤B5、对有功功率实际值p和无功功率实际值q求导，分别获得有功功率导数

Figure BDA0003126557740000109
和无功功率导数
Figure BDA00031265577400001010
令
Figure BDA00031265577400001011
获得等效点的平均占空比
Figure BDA00031265577400001012
Step B5, derive the actual value p of the active power and the actual value q of the reactive power, and obtain the derivatives of the active power respectively
Figure BDA0003126557740000109
and reactive power derivative
Figure BDA00031265577400001010
make
Figure BDA00031265577400001011
Obtain the average duty cycle of the equivalent point
Figure BDA00031265577400001012

步骤B6、根据有功功率纠正量up、无功功率纠正量uq和等效点的平均占空比

Figure BDA00031265577400001013
获得平均占空比δαβ。Step B6, according to the active power correction amount u p , the reactive power correction amount u q and the average duty cycle of the equivalent point
Figure BDA00031265577400001013
The average duty cycle δαβ is obtained.

进一步地，本实施方式中，Further, in this embodiment,

步骤B2中，二阶滑模控制器SOSM的动态方程为：In step B2, the dynamic equation of the second-order sliding mode controller SOSM is:

Figure BDA00031265577400001014
Figure BDA00031265577400001014

其中，ki1和ki2是二阶滑模控制器SOSM的增益，t为时间。Among them, k i1 and k i2 are the gains of the second-order sliding mode controller SOSM, and t is the time.

进一步地，本实施方式中，Further, in this embodiment,

步骤B3中，获得内部扰动的估计值

Figure BDA00031265577400001015
和
Figure BDA00031265577400001016
通过以下公式计算实现：In step B3, the estimated value of the internal disturbance is obtained
Figure BDA00031265577400001015
and
Figure BDA00031265577400001016
It is calculated by the following formula:

Figure BDA00031265577400001017
Figure BDA00031265577400001017

Figure BDA00031265577400001018
Figure BDA00031265577400001018

和and

Figure BDA0003126557740000111
Figure BDA0003126557740000111

Figure BDA0003126557740000112
Figure BDA0003126557740000112

其中，

Figure BDA0003126557740000113
vα和vβ为αβ坐标系下的变换器交流侧电压的α分量和β分量，L为交流侧线电感，α3是有功功率环路高增益观测器的前级增益参数，α4是有功功率环路高增益观测器的后级增益参数，α5是无功功率环路高增益观测器的前级增益参数，α6是无功功率环路高增益观测器的后级增益参数，εp是有功功率环路高增益观测器的增益，εq是无功功率环路高增益观测器的增益。in,
Figure BDA0003126557740000113
v α and v β are the α component and β component of the AC side voltage of the converter in the αβ coordinate system, L is the AC side line inductance, α 3 is the front stage gain parameter of the active power loop high gain observer, and α 4 is the active power loop. The post-stage gain parameter of the power loop high-gain observer, α 5 is the pre-stage gain parameter of the reactive power loop high-gain observer, α 6 is the post-stage gain parameter of the reactive power loop high-gain observer, ε p is the gain of the active power loop high gain observer, and ε q is the gain of the reactive power loop high gain observer.

本实施方式中，需要实现的控制目标是对有功功率实际值p和无功功率实际值q进行控制，使他们分别跟踪其参考值。此处采用SOSM控制策略来实现该目标以保证系统的快速响应和稳态性能。此外，为了提升系统对参数摄动的鲁棒性，还设计了一个HGO来抑制不确定参数对系统的影响。SOSM的英文全称为Second-order sliding mode，二阶滑模；HGO的英文全称为High-gain observer，高增益观测器。In this embodiment, the control objective that needs to be realized is to control the actual value p of the active power and the actual value q of the reactive power, so that they track their reference values respectively. Here, the SOSM control strategy is adopted to achieve this goal to ensure the fast response and steady-state performance of the system. In addition, in order to improve the robustness of the system to parameter perturbation, an HGO is also designed to suppress the influence of uncertain parameters on the system. The English full name of SOSM is Second-order sliding mode, the second-order sliding mode; the English full name of HGO is High-gain observer, high gain observer.

进一步地，本实施方式中，Further, in this embodiment,

步骤B4中，当前采样点时刻有功功率纠正量up和无功功率纠正量uq为：In step B4, the active power correction amount u p and the reactive power correction amount u q at the current sampling point are:

Figure BDA0003126557740000114
Figure BDA0003126557740000114

其中，

Figure BDA0003126557740000115
in,
Figure BDA0003126557740000115

进一步地，本实施方式中，Further, in this embodiment,

步骤B5中，等效点的平均占空比

Figure BDA0003126557740000116
为：In step B5, the average duty cycle of the equivalent point
Figure BDA0003126557740000116
for:

Figure BDA0003126557740000117
Figure BDA0003126557740000117

其中，vαβ为αβ坐标系下的变换器交流侧电压；J为矩阵，且

Figure BDA0003126557740000118
ω为电网电压的角频率；L为交流侧线电感。Among them, v αβ is the AC side voltage of the converter in the αβ coordinate system; J is a matrix, and
Figure BDA0003126557740000118
ω is the angular frequency of the grid voltage; L is the inductance of the AC side line.

进一步地，本实施方式中，Further, in this embodiment,

采用电压平衡环，对直流侧不平衡电压实际值edc与直流侧不平衡电压参考值

Figure BDA0003126557740000119
作差，并将二者的差值进行PI调节，获得平衡占空比δba为:The voltage balance loop is used to compare the actual value of the unbalanced voltage on the DC side, e dc , and the reference value of the unbalanced voltage on the DC side.
Figure BDA0003126557740000119
Make a difference, and adjust the difference between the two by PI to obtain a balanced duty cycle δba as:

Figure BDA0003126557740000121
Figure BDA0003126557740000121

其中，kpb为PI控制器的比例环节增益；kib为PI控制器的积分环节增益；t为时间。Among them, k pb is the proportional link gain of the PI controller; k ib is the integral link gain of the PI controller; t is the time.

为了验证本申请提出的控制策略的优越性，本发明所述的一种三电平NPC变换器的滑模控制方法与传统的PI控制策略，基于扩张状态观测器的滑模控制策略(LESO-SMC)以及基于非线性高增益观测器的滑模控制策略(NHGO-SMC)进行了实验对比。三电平NPC变换器的参数如表I所示。In order to verify the superiority of the control strategy proposed in this application, the sliding mode control method of a three-level NPC converter described in the present invention and the traditional PI control strategy, the sliding mode control strategy based on the extended state observer (LESO- SMC) and a sliding mode control strategy based on a nonlinear high-gain observer (NHGO-SMC) were compared experimentally. The parameters of the three-level NPC converter are shown in Table I.

表I实验平台参数Table I experimental platform parameters

Figure BDA0003126557740000122
Figure BDA0003126557740000122

首先进行的是动态性能的测试。图3a、图3b、图3c和图3d分别是采用PI控制器、LESO-SMC控制器、NHGO-SMC控制器以及本申请提出的NHGO-ASMC控制器(即：本发明所述一种三电平NPC变换器的滑模控制方法)下电压指令从750V变为690V时的动态响应图，图4a、图4b、图4c和图4d则为690V变为750V时的动态响应图。从这两次的实验比较结果均可看出本申请具备更好的动态性能，动态响应时间短，超调电压小。The first step is to test the dynamic performance. Figure 3a, Figure 3b, Figure 3c and Figure 3d respectively use the PI controller, the LESO-SMC controller, the NHGO-SMC controller and the NHGO-ASMC controller proposed in this application (ie: a three-electrical Figure 4a, Figure 4b, Figure 4c and Figure 4d are the dynamic response diagrams when 690V is changed to 750V. From the comparison results of the two experiments, it can be seen that the present application has better dynamic performance, short dynamic response time and small overshoot voltage.

接下来进行的是稳态性能的测试。连接直流负载R1和R2，当NPC稳定运行时，观察不同控制策略下的三相交流电流质量。图5a、图5b、图5c和图5d分别是采用PI控制器、LESO-SMC控制器、NHGO-SMC控制器以及本申请提出的NHGO-ASMC控制器下的电流谐波频谱图，它们的总谐波失真(THD)分别为2.1％、2.2％、2.1％和2.0％。显然，本申请提出的NHGO-ASMC控制策略得到的电流THD更低，电流质量更好。The next step is to test the steady state performance. Connect the DC loads R 1 and R 2 , and observe the three-phase AC current quality under different control strategies when the NPC runs stably. Fig. 5a, Fig. 5b, Fig. 5c and Fig. 5d are the current harmonic spectrum diagrams using the PI controller, the LESO-SMC controller, the NHGO-SMC controller and the NHGO-ASMC controller proposed in this application, respectively. Their total The harmonic distortion (THD) was 2.1%, 2.2%, 2.1% and 2.0%, respectively. Obviously, the current THD obtained by the NHGO-ASMC control strategy proposed in this application is lower and the current quality is better.

最后进行的是抗扰动性能测试。在R1接入电路时的直流母线电压和a相电流的瞬态响应如图6所示，图6a、图6b、图6c和图6d分别是采用PI控制器、LESO-SMC控制器、NHGO-SMC控制器以及本申请提出的NHGO-ASMC控制器下的瞬态响应。可以观察到，本申请所提出的方法相比于其他控制器实现了更小的电压波动以及恢复时间。因此，根据上述实验结果，与其他的控制策略相比，本申请提出的NHGO-ASMC控制策略具有更好的抗干扰能力。Finally, the anti-disturbance performance test is carried out. The transient responses of DC bus voltage and phase a current when R1 is connected to the circuit are shown in Fig. 6. Fig. 6a, Fig. 6b, Fig. 6c and Fig. 6d are the PI controller, LESO-SMC controller, NHGO controller, respectively. - Transient response under the SMC controller and the NHGO-ASMC controller proposed in this application. It can be observed that the method proposed in this application achieves smaller voltage fluctuation and recovery time than other controllers. Therefore, according to the above experimental results, compared with other control strategies, the NHGO-ASMC control strategy proposed in this application has better anti-interference ability.

虽然在本文中参照了特定的实施方式来描述本发明，但是应该理解的是，这些实施例仅仅是本发明的原理和应用的示例。因此应该理解的是，可以对示例性的实施例进行许多修改，并且可以设计出其他的布置，只要不偏离所附权利要求所限定的本发明的精神和范围。应该理解的是，可以通过不同于原始权利要求所描述的方式来结合不同的从属权利要求和本文中所述的特征。还可以理解的是，结合单独实施例所描述的特征可以使用在其他所述实施例中。Although the invention has been described herein with reference to specific embodiments, it should be understood that these embodiments are merely illustrative of the principles and applications of the invention. It should therefore be understood that many modifications may be made to the exemplary embodiments and other arrangements can be devised without departing from the spirit and scope of the invention as defined by the appended claims. It should be understood that the features described in the various dependent claims and herein may be combined in different ways than are described in the original claims. It will also be appreciated that features described in connection with a single embodiment may be used in other described embodiments.

Claims (10)

1. A sliding mode control method of a three-level NPC converter is characterized by comprising the following steps:
the method adopts a direct-current voltage regulating ring and utilizes the actual value v of the direct-current side voltage of the three-level NPC converterdcAnd a DC side voltage reference value
Figure FDA0003126557730000011
Obtaining an active power reference value p of a direct current side at the current sampling point moment through an Adaptive Sliding Mode Controller (ASMC) and a non-linear high-gain observer (NHGO)*；
An instantaneous power tracking loop is adopted, and the active power reference value p of the current sampling point on the direct current side is utilized*An active power actual value p, a reactive power actual value q and a preset reactive power reference value q*Acquiring the average duty cycle number delta of the three-level NPC converter through a second-order sliding mode controllerαβ；
For three-level NPC converterMean duty cycle deltaαβPerforming alpha beta/abc coordinate transformation to obtain the average duty ratio delta 'of the three-level NPC converter in the abc coordinate system'abc；
Using a voltage balancing loop to measure the actual value e of the DC-side unbalanced voltagedcWith reference value of DC-side unbalanced voltage
Figure FDA0003126557730000012
Making difference, and performing PI regulation on the difference value of the two values to obtain a balance duty ratio deltaba；
For balanced duty cycle deltabaSum level NPC converter average duty cycle delta'abcAnd after addition, a control signal of a switching tube of the three-level NPC converter is obtained through a pulse width modulator, so that the three-level NPC converter is controlled.
2. The sliding-mode control method of the three-level NPC converter according to claim 1, characterized in that the actual value v of the DC-side voltage of the three-level NPC converter is utilized by using a DC voltage regulation loopdcAnd a DC side voltage reference value
Figure FDA0003126557730000013
Obtaining an active power reference value p of a direct current side at the current sampling point moment through an Adaptive Sliding Mode Controller (ASMC) and a non-linear high-gain observer (NHGO)*The specific method comprises the following steps:
step A1, utilizing the actual value v of the DC side voltage of the three-level NPC converterdcAnd a DC side voltage reference value
Figure FDA0003126557730000014
Calculating the tracking error s of DC voltage regulation loopv(ii) a Wherein,
Figure FDA0003126557730000015
step A2, tracking error s of DC voltage regulation loop through self-adaptive sliding mode controller ASMCvCorrecting and outputting the corrected tracking errorDifference (D)
Figure FDA0003126557730000016
Step A3, adopting a nonlinear high-gain observer NHGO to perform on x in step A11And the active power reference value p at the last sampling point moment*Active power x to DC load2Observing to obtain the estimated value of the load power at the DC side
Figure FDA0003126557730000017
Step A4, correcting the corrected tracking error
Figure FDA0003126557730000018
And estimated value of DC side load power
Figure FDA0003126557730000019
Adding to obtain the active power reference value p at the current sampling point moment*。
3. The sliding-mode control method of the three-level NPC converter according to claim 3, wherein in step a2, the dynamic equation of the adaptive sliding-mode controller ASMC is:
Figure FDA0003126557730000021
wherein, KvAdaptive rate of adaptive sliding mode controller for gain of adaptive sliding mode controller ASMC, for a time variable
Figure FDA0003126557730000022
Comprises the following steps:
Figure FDA0003126557730000023
wherein, KlIs the gain change rate of the adaptive sliding mode controller ASMC,Kmis a gain decision parameter of the adaptive sliding mode controller ASMC, baIs a gain gradient decision parameter of the adaptive sliding mode controller ASMC.
4. The sliding-mode control method of the three-level NPC converter according to claim 3, wherein in step A3, the dynamic equation of the nonlinear high-gain observer NHGO is:
Figure FDA0003126557730000024
Figure FDA0003126557730000025
wherein C is the capacitance of the DC side capacitor,
Figure FDA0003126557730000026
is a variable x1Is determined by the estimated value of (c),
Figure FDA0003126557730000027
is a variable x1An estimate of the derivative of (a) is,
Figure FDA0003126557730000028
active power x being a DC load2Is estimated from the derivative of (a)1Preceding stage gain parameter, alpha, of a non-linear high-gain observer NHGO2Post-stage gain parameter, epsilon, of a non-linear high-gain observer NHGO1And ε2Two gain parameters of a non-linear high-gain observer NHGO, and epsilon2Greater than epsilon1 bsAnd the gain judgment parameter of the non-linear high-gain observer NHGO is a saturation function sat (·).
5. The sliding-mode control method of the three-level NPC converter according to claim 1 or 4, characterized in that an instantaneous power tracking loop is adoptedUsing the active power reference value p of the DC side at the current sampling point moment*An active power actual value p, a reactive power actual value q and a preset reactive power reference value q*Acquiring the average duty cycle number delta of the three-level NPC converter through a second-order sliding mode controllerαβThe specific method comprises the following steps:
step B1, obtaining the active power reference value p at the current sampling point moment*Comparing with the actual value p of the active power to obtain the tracking error s of the active powerpSimultaneously, the real reactive power value q at the current sampling point moment and a preset reactive power reference value q are obtained*Comparing to obtain the tracking error s of reactive powerq；
Step B2, tracking error s of active power through second-order sliding mode controller SOSMpAnd reactive power tracking error sqCorrecting to obtain the corrected active power tracking error
Figure FDA0003126557730000029
And the corrected reactive power tracking error
Figure FDA00031265577300000210
Step B3, tracking error s according to active powerpReactive power tracking error sqLast sampling point moment active power correction upAnd the reactive power correction u at the moment of the last sampling pointqUsing a high gain observer NHGO to correct the internal disturbance l caused by the uncertain parameters of the systempAnd lqObserving to obtain the estimated value of internal disturbance
Figure FDA0003126557730000031
And
Figure FDA0003126557730000032
wherein the correction amount u of active powerpAnd a reactive power correction uqIs 0;
step B4, according to the corrected tracking error
Figure FDA0003126557730000033
And
Figure FDA0003126557730000034
estimation of internal disturbances
Figure FDA0003126557730000035
And
Figure FDA0003126557730000036
updating the active power correction u at the current sampling point momentpAnd a reactive power correction uq；
Step B5, the real value p of the active power and the real value q of the reactive power are differentiated to obtain the derivative of the active power respectively
Figure FDA0003126557730000037
And reactive power derivative
Figure FDA0003126557730000038
Order to
Figure FDA0003126557730000039
Obtaining an average duty cycle of the equivalent point
Figure FDA00031265577300000310
Step B6, correcting quantity u according to active powerpAnd a reactive power correction amount uqAnd average duty cycle of equivalent point
Figure FDA00031265577300000311
Obtaining the average duty cycle deltaαβ。
6. The sliding-mode control method of the three-level NPC converter according to claim 5, wherein in step B2, the dynamic equation of the second-order sliding-mode controller SOSM is:
Figure FDA00031265577300000312
wherein k isi1And ki2Is the gain of the second order sliding mode controller SOSM, t is time.
7. The sliding-mode control method for the three-level NPC converter according to claim 6, characterized in that in step B3, an estimate of the internal disturbance is obtained
Figure FDA00031265577300000313
And
Figure FDA00031265577300000314
the method is realized by the following formula:
Figure FDA00031265577300000315
Figure FDA00031265577300000316
and
Figure FDA00031265577300000317
Figure FDA00031265577300000318
wherein,
Figure FDA00031265577300000319
wherein v isαAnd vβIs alpha component and beta of converter AC side voltage in alpha beta coordinate systemComponent, L is AC side line inductance, α3Is a preceding-stage gain parameter, alpha, of a high-gain observer of an active power loop4Is a back-stage gain parameter, alpha, of the active power loop high-gain observer5Is the preceding-stage gain parameter, alpha, of the high-gain observer of the reactive power loop6Is a post-stage gain parameter, epsilon, of a reactive power loop high-gain observerpIs the gain, ε, of a high-gain observer of an active power loopqIs the gain of the reactive power loop high-gain observer.
8. The sliding-mode control method for the three-level NPC converter according to claim 7, wherein in step B4, the correction amount u of the active power at the current sampling point momentpAnd a reactive power correction uqComprises the following steps:
Figure FDA0003126557730000041
wherein,
Figure FDA0003126557730000042
9. the sliding-mode control method of the three-level NPC converter as claimed in claim 8, wherein in step B5, the average duty cycle of the equivalent point
Figure FDA0003126557730000043
Comprises the following steps:
Figure FDA0003126557730000044
a computational implementation wherein vαβThe voltage of the AC side of the converter under an alpha beta coordinate system; j is a matrix, and
Figure FDA0003126557730000045
omega is the voltage of the power gridThe angular frequency of (d); l is an alternating current side wire inductor.
10. The sliding-mode control method of the three-level NPC converter according to claim 1 or 5, characterized in that the actual value e of the DC-side unbalanced voltage is adjusted by using a voltage balancing loopdcWith reference value of DC-side unbalanced voltage
Figure FDA0003126557730000046
Making difference, and performing PI regulation on the difference value of the two values to obtain a balance duty ratio deltabaComprises the following steps:
Figure FDA0003126557730000047
wherein k ispbThe proportional link gain of the PI controller is obtained; k is a radical ofibThe integral link gain of the PI controller is obtained; t is time.
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