CN113346770A - Sliding mode control method of three-level NPC converter - Google Patents

Sliding mode control method of three-level NPC converter Download PDF

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CN113346770A
CN113346770A CN202110692178.7A CN202110692178A CN113346770A CN 113346770 A CN113346770 A CN 113346770A CN 202110692178 A CN202110692178 A CN 202110692178A CN 113346770 A CN113346770 A CN 113346770A
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gain
active power
level npc
voltage
npc converter
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CN113346770B (en
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吴立刚
刘健行
孙光辉
沈肖宁
殷允飞
高亚斌
姚蔚然
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Harbin Institute of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/145—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means
    • H02M7/155—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a thyratron or thyristor type requiring extinguishing means using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/01—Arrangements for reducing harmonics or ripples
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/36—Arrangements for transfer of electric power between ac networks via a high-tension dc link
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
    • Y02E40/40—Arrangements for reducing harmonics
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E60/00—Enabling technologies; Technologies with a potential or indirect contribution to GHG emissions mitigation
    • Y02E60/60—Arrangements for transfer of electric power between AC networks or generators via a high voltage DC link [HVCD]

Abstract

A sliding mode control method of a three-level NPC converter belongs to the technical field of power electronic control and solves the problems that the existing sliding mode control based on an observer of the three-level NPC converter is large in buffeting and sensitive to measurement noise. The invention adopts a direct-current voltage adjusting ring to obtain the active power reference value p of the load at the current sampling point moment*(ii) a Acquiring average duty cycle number delta of three-level NPC converter by adopting instantaneous power tracking loopαβ(ii) a Using voltage-balancing ringsTo obtain a balanced duty cycle deltaba(ii) a For balanced duty cycle deltabaSum level NPC converter average duty cycle delta'abcAnd after addition, a control signal of a switching tube of the three-level NPC converter is obtained through a pulse width modulator, so that the three-level NPC converter is controlled. The invention is suitable for controlling the three-level NPC converter.

Description

Sliding mode control method of three-level NPC converter
Technical Field
The invention belongs to the technical field of power electronic control, and particularly relates to a sliding mode control method of a three-level NPC converter.
Background
Over the past few decades, a wide variety of multilevel power converter topologies have been proposed to meet the needs of medium and high voltage applications. A three-level Neutral Point Clamped (NPC) power converter was first proposed in 1979 as a high performance and low loss multi-level power converter. Compared with a traditional two-level converter, the NPC power converter has the advantages of higher voltage level, better output voltage waveform and the like. At present, the high-voltage power supply is mature and applied to active front end or variable speed drive and other medium and high voltage industrial applications. Such as direct current micro-grids (MGs), photovoltaic power generation, wind turbines, motor drives and energy storage systems, etc.
Currently, there are many control methods used by NPC converters in these applications, including traditional PI control and recently proposed observer-based sliding mode control, among others. Although they all achieve basic control objectives, there are still some drawbacks that can be summarized as:
(1) by using PI control, the normal operation of NPC can be ensured, but the dynamic and steady-state performance of the NPC cannot be ensured; in addition, when external interference occurs, for example, in the loading condition, the PI controller cannot quickly suppress disturbance, so that a large overshoot of the dc bus voltage occurs, the entire system is damaged, and the anti-interference performance is poor.
(2) By using sliding mode control based on the observer, although the dynamic performance can be improved, a large buffeting phenomenon is generated, and adverse effects are caused on the system. Furthermore, the disturbance observer, although improving the immunity of the system, is very sensitive to measurement noise, limiting its performance.
Therefore, in summary, the method for controlling the three-level NPC converter by using the conventional PI control algorithm has the disadvantages of poor anti-interference performance and poor system steady-state performance and dynamic response performance. And the sliding mode control based on the observer has the problems of large buffeting and high sensitivity to measurement noise. Therefore, the above problems need to be solved.
Disclosure of Invention
The invention aims to solve the problems that the existing sliding mode control based on an observer of a three-level NPC converter has large buffeting and is very sensitive to measurement noise, and provides a sliding mode control method of the three-level NPC converter.
The invention discloses a sliding mode control method of a three-level NPC converter, which comprises the following steps:
the method adopts a direct-current voltage regulating ring and utilizes the actual value v of the direct-current side voltage of the three-level NPC converterdcAnd a DC side voltage reference value
Figure BDA0003126557740000011
Obtaining an active power reference value p of a current sampling point moment direct current side through a self-adaptive sliding mode controller*；
An instantaneous power tracking loop is adopted, and the active power reference value p of the current sampling point on the direct current side is utilized*An active power actual value p, a reactive power actual value q and a preset reactive power reference value q*Acquiring the average duty cycle number delta of the three-level NPC converter through a second-order sliding mode controllerαβ；
Average duty ratio delta for three-level NPC converterαβCarrying out alpha beta/abc coordinate transformation to obtain the average duty ratio delta of the three-level NPC converter under the abc coordinate systema'bc；
Using a voltage balancing loop to measure the actual value e of the DC-side unbalanced voltagedcWith reference value of DC-side unbalanced voltage
Figure BDA0003126557740000021
Making difference, and performing PI regulation on the difference value of the two values to obtain a balance duty ratio deltaba；
For balanced duty cycle deltabaAverage duty cycle delta of sum level NPC convertera'bcAnd after addition, a control signal of a switching tube of the three-level NPC converter is obtained through a pulse width modulator, so that the three-level NPC converter is controlled.
Further, in the invention, a direct-current voltage regulating ring is adopted, and the actual value v of the direct-current side voltage of the three-level NPC converter is utilizeddcAnd a DC side voltage reference value
Figure BDA0003126557740000022
Obtaining an active power reference value p of a load at the moment of a current sampling point through an Adaptive Sliding Mode Controller (ASMC) and a non-linear high-gain observer (NHGO)*The specific method comprises the following steps:
step A1, utilizing the actual value v of the DC side voltage of the three-level NPC converterdcAnd a DC side voltage reference value
Figure BDA0003126557740000023
Calculating the tracking error s of DC voltage regulation loopv(ii) a Wherein the content of the first and second substances,
Figure BDA0003126557740000024
step A2, tracking error s of DC voltage regulation loop through self-adaptive sliding mode controller ASMCvCorrecting and outputting the corrected tracking error
Figure BDA0003126557740000025
Step A3, adopting a nonlinear high-gain observer NHGO to perform on x in step A11And the active power reference value p at the last sampling point moment*Active power x to DC load2Observing to obtain the estimated value of the load power at the DC side
Figure BDA0003126557740000026
Step A4, correcting the corrected tracking error
Figure BDA0003126557740000027
And estimated value of DC side load power
Figure BDA0003126557740000028
Adding to obtain the active power reference value p at the current sampling point moment*。
Further, in the present invention, in step a2, the dynamic equation of the adaptive sliding mode controller ASMC is:
Figure BDA0003126557740000029
wherein, KvAdaptive rate of adaptive sliding mode controller for gain of adaptive sliding mode controller ASMC, for a time variable
Figure BDA00031265577400000210
Comprises the following steps:
Figure BDA00031265577400000211
wherein, KlIs the gain change rate, K, of the adaptive sliding mode controller ASMCmIs a gain decision parameter of the adaptive sliding mode controller ASMC, baIs a gain gradient decision parameter of the adaptive sliding mode controller ASMC.
Further, in the present invention, in step a3, the dynamic equation of the nonlinear high-gain observer NHGO is:
Figure BDA0003126557740000031
Figure BDA0003126557740000032
wherein C is the capacitance of the DC side capacitor,
Figure BDA0003126557740000033
is a variable x1Is determined by the estimated value of (c),
Figure BDA0003126557740000034
is a variable x1An estimate of the derivative of (a) is,
Figure BDA0003126557740000035
active power x being a DC load2Is estimated from the derivative of (a)1Preceding stage gain parameter, alpha, of a non-linear high-gain observer NHGO2Post-stage gain parameter, epsilon, of a non-linear high-gain observer NHGO1And ε2Two gain parameters of a non-linear high-gain observer NHGO, and epsilon2Greater than epsilon1,bsAnd the gain judgment parameter of the non-linear high-gain observer NHGO is a saturation function sat (·).
Furthermore, in the invention, an instantaneous power tracking loop is adopted, and the active power reference value p of the current sampling point moment direct current side is utilized*An active power actual value p, a reactive power actual value q and a preset reactive power reference value q*Obtaining the average duty ratio delta of the three-level NPC converter through a second-order sliding mode controllerαβThe specific method comprises the following steps:
step B1, obtaining the active power reference value p at the current sampling point moment*Comparing with the actual value p of the active power to obtain the tracking error s of the active powerpSimultaneously, the real reactive power value q at the current sampling point moment and a preset reactive power reference value q are obtained*Comparing to obtain the tracking error s of reactive powerq；
Step B2, tracking error s of active power through second-order sliding mode controller SOSMpAnd reactive power tracking error sqCorrecting to obtain the corrected active power tracking error
Figure BDA0003126557740000036
And the corrected reactive power tracking error
Figure BDA0003126557740000037
Step B3, tracking error s according to active powerpReactive power tracking error sqLast sampling point moment active power correction upAnd the reactive power correction u at the moment of the last sampling pointqUsing a high gain observer NHGO to correct the internal disturbance l caused by the uncertain parameters of the systempAnd lqObserving to obtain the estimated value of internal disturbance
Figure BDA0003126557740000038
And
Figure BDA0003126557740000039
wherein the correction amount u of active powerpAnd a reactive power correction uqIs 0;
step B4, according to the corrected tracking error
Figure BDA00031265577400000310
And
Figure BDA00031265577400000311
estimation of internal disturbances
Figure BDA00031265577400000312
And
Figure BDA00031265577400000313
updating the active power correction u at the current sampling point momentpAnd a reactive power correction uq；
Step B5, the real value p of the active power and the real value q of the reactive power are differentiated to obtain the derivative of the active power respectively
Figure BDA00031265577400000314
And reactive power derivative
Figure BDA00031265577400000315
Order to
Figure BDA00031265577400000316
Obtaining an average duty cycle of the equivalent point
Figure BDA00031265577400000317
Step B6, correcting quantity u according to active powerpAnd a reactive power correction amount uqAnd average duty cycle of equivalent point
Figure BDA00031265577400000318
Obtaining the average duty cycle deltaαβ。
Further, in the present invention, in step B2, the dynamic equation of the second-order sliding mode controller SOSM is:
Figure BDA0003126557740000041
wherein k isi1And ki2Is the gain of the second order sliding mode controller SOSM, t is time.
Further, in the present invention, in step B3, an estimated value of the internal disturbance is obtained
Figure BDA0003126557740000042
And
Figure BDA0003126557740000043
by the formula:
Figure BDA0003126557740000044
Figure BDA0003126557740000045
and
Figure BDA0003126557740000046
Figure BDA0003126557740000047
a computational implementation in which, among other things,
Figure BDA0003126557740000048
wherein v isαAnd vβIs alpha component and beta component of converter AC side voltage in alpha beta coordinate system, L is AC side line inductance, alpha3Is a preceding-stage gain parameter, alpha, of a high-gain observer of an active power loop4Is a back-stage gain parameter, alpha, of the active power loop high-gain observer5Is the preceding-stage gain parameter, alpha, of the high-gain observer of the reactive power loop6Is a post-stage gain parameter, epsilon, of a reactive power loop high-gain observerpIs the gain, ε, of a high-gain observer of an active power loopqIs the gain of the reactive power loop high-gain observer.
Further, in the present invention, in step B4, the current sampling point time active power correction upAnd a reactive power correction uqComprises the following steps:
Figure BDA0003126557740000049
wherein the content of the first and second substances,
Figure BDA00031265577400000410
further, in the present invention, in step B5, the average duty ratio of the equivalent point
Figure BDA00031265577400000411
Comprises the following steps:
Figure BDA00031265577400000412
wherein v isαβThe voltage of the AC side of the converter under an alpha beta coordinate system; j is a matrix, and
Figure BDA00031265577400000413
omega is the angular frequency of the grid voltage; l is an alternating current side wire inductor.
Furthermore, in the invention, a voltage balance ring is adopted to compare the actual value e of the unbalanced voltage on the direct current sidedcWith reference value of DC-side unbalanced voltage
Figure BDA0003126557740000051
Making difference, and performing PI regulation on the difference value of the two values to obtain a balance duty ratio deltabaComprises the following steps:
Figure BDA0003126557740000052
wherein k ispbThe proportional link gain of the PI controller is obtained; k is a radical ofibThe integral link gain of the PI controller is obtained; t is time.
The method improves the dynamic and steady-state performance and the anti-interference capability of the three-phase NPC converter. The sliding mode control method of the three-level NPC converter is realized based on a direct-current voltage adjusting ring, an instantaneous power tracking ring and a voltage balancing ring. The actual value v of the DC side voltage is adjusted by a DC voltage adjusting ringdcControl is performed so that the sum v of the DC-side capacitor voltagesdcAdjusted to corresponding desired values
Figure BDA0003126557740000053
The real value p of the active power and the real value q of the reactive power are controlled by an instantaneous power tracking loop, so that the active power p and the reactive power q accurately track respective reference values p*And q is*And the actual value e of the unbalanced voltage on the DC side is calculated by the voltage balance ringdcControl is carried out to ensure that the unbalanced voltage of two capacitors on the direct current side is close to 0 and the combined action of three ringsThe control signal is generated to control the three-level NPC converter, the control process is simple, the direct-current bus voltage is regulated through the direct-current voltage regulating ring, so that the rapid dynamic response of a voltage step stage is ensured, and the fluctuation of the direct-current side voltage caused when an uncertain interfered direct-current side load is connected into the circuit can be effectively inhibited. The voltage balance of the direct-current side capacitor is realized, and the control stability is improved.
In the direct-current voltage regulating loop, a self-Adaptive Sliding Mode Controller (ASMC) is adopted to quickly regulate the direct-current bus voltage, so that buffeting is reduced, and quick dynamic response of a voltage step stage is ensured. Meanwhile, due to the existence of external uncertain disturbance, a nonlinear high-gain observer (NHGO) is added on the basis of an Adaptive Sliding Mode Controller (ASMC) (direct current voltage regulating loop) to inhibit the external uncertain disturbance (the existence of the external uncertain disturbance), and the observer is insensitive to noise and has strong disturbance rejection capability; in the instantaneous power tracking loop, a simple and effective Direct Power Control (DPC) strategy is adopted to realize the aim of power tracking, thereby simplifying the control process of an inner loop. In addition, in order to obtain alternating current with low harmonic distortion and robustness to system parameter perturbation, a second-order sliding mode controller (SOSM) based on a High Gain Observer (HGO) is adopted to ensure that active power and reactive power can rapidly converge to a stable state; finally, in the voltage balancing loop, a PI regulator is used to ensure voltage balancing of the dc link capacitor. Through experimental tests, the NPC power converter control strategy provided by the invention is compared with other control schemes, and the effectiveness and superiority of the scheme are proved.
Drawings
Fig. 1 is a schematic diagram of a circuit principle of connection between a three-level NPC converter and an ac power grid and a dc microgrid according to the present invention;
FIG. 2 is a schematic illustration of the principle of generating control signals according to the present invention;
FIG. 3 is a diagram of DC side voltage waveforms of a three-level NPC converter when the voltage reference is adjusted from 750V to 690V; wherein the content of the first and second substances,
FIG. 3a is a diagram of DC side voltage waveforms under the PI controller;
FIG. 3b is a diagram of DC-side voltage waveforms under a sliding mode control strategy (LESO-SMC) based on an extended state observer;
FIG. 3c is a graph of DC side voltage waveforms under a sliding mode control strategy (NHGO-SMC) based on a non-linear high-gain observer;
FIG. 3d is a diagram of DC side voltage waveforms under the control method of the present invention;
FIG. 4 is a diagram of the DC side voltage waveform of a three-level NPC converter with the voltage reference adjusted from 690V to 750V; wherein the content of the first and second substances,
FIG. 4a is a diagram of DC side voltage waveforms under the PI controller;
FIG. 4b is a diagram of DC-side voltage waveforms under a sliding mode control strategy (LESO-SMC) based on an extended state observer;
FIG. 4c is a graph of DC side voltage waveforms under a sliding mode control strategy (NHGO-SMC) based on a non-linear high-gain observer;
FIG. 4d is a diagram of DC side voltage waveforms under the control method of the present invention;
FIG. 5a shows an access load R1And R2When the device runs stably, a three-phase alternating current harmonic frequency spectrogram under a PI controller;
FIG. 5b shows an access load R1And R2When the device runs stably, a three-phase alternating current harmonic frequency spectrogram under a sliding mode control strategy (LESO-SMC) based on an extended state observer;
FIG. 5c shows an access load R1And R2When the frequency spectrum is stably operated, a three-phase alternating current harmonic frequency spectrum diagram is based on a sliding mode control strategy (NHGO-SMC) of a nonlinear high-gain observer;
FIG. 5d shows an access load R1And R2When the three-phase alternating current harmonic spectrum is stably operated, the three-phase alternating current harmonic spectrum chart under the control method is adopted;
FIG. 6a is the transient response of the DC bus voltage and phase a current under the PI controller;
FIG. 6b is the transient response of the DC bus voltage and the a-phase current under the sliding mode control strategy (LESO-SMC) based on the extended state observer;
FIG. 6c is the transient response of DC bus voltage and a-phase current under the sliding mode control strategy (NHGO-SMC) based on a non-linear high-gain observer;
fig. 6d is the transient response of the dc bus voltage and the a-phase current under the control method of the present invention.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
It should be noted that the embodiments and features of the embodiments may be combined with each other without conflict.
The first embodiment is as follows: the present embodiment is described below with reference to fig. 1, and the sliding mode control method of the three-level NPC converter according to the present embodiment includes:
the method adopts a direct-current voltage regulating ring and utilizes the actual value v of the direct-current side voltage of the three-level NPC converterdcAnd a DC side voltage reference value
Figure BDA0003126557740000071
Obtaining an active power reference value p of a current sampling point moment direct current side through a self-adaptive sliding mode controller*；
An instantaneous power tracking loop is adopted, and the active power reference value p of the current sampling point on the direct current side is utilized*An active power actual value p, a reactive power actual value q and a preset reactive power reference value q*Acquiring the average duty cycle number delta of the three-level NPC converter through a second-order sliding mode controllerαβ；
Average duty ratio delta for three-level NPC converterαβCarrying out alpha beta/abc coordinate transformation to obtain the average duty ratio delta of the three-level NPC converter under the abc coordinate systema'bc；
Using voltage balancingLoop, versus actual value e of DC side unbalance voltagedcWith reference value of DC-side unbalanced voltage
Figure BDA0003126557740000072
Making difference, and performing PI regulation on the difference value of the two values to obtain a balance duty ratio deltaba；
For balanced duty cycle deltabaAverage duty cycle delta of sum level NPC convertera'bcAnd after addition, a control signal of a switching tube of the three-level NPC converter is obtained through a pulse width modulator, so that the three-level NPC converter is controlled.
Before the method is applied specifically, a state space average model of the three-level NPC converter is established according to the operation principle of the three-level NPC converter; determining a control target of the three-level NPC converter according to a state space average model of the three-level NPC converter; the control targets include: make the sum v of the voltages of two capacitors on the DC sidedcAdjusting to desired value of DC side voltage reference value
Figure BDA0003126557740000073
Making active power p and reactive power q always track respective reference value p*And q is*And ensuring that the unbalanced voltage of the two capacitors on the direct current side approaches 0 to generate a corresponding control signal to control the three-level NPC converter.
In fact, it is necessary to adopt an efficient control method. The method can ensure that active power and reactive power are kept near an equivalent point when a system reaches a stable state, and ensures higher current quality. The three-level NPC converter can not only realize different control targets, but also improve the dynamic and steady-state performance and the anti-interference capability of the three-level NPC converter.
In fig. 1, a three-phase ac power source and an inductor are connected to the ac side of a three-level NPC converter to provide power transfer. On the dc side, a three-level NPC converter connects two capacitors to store energy and stabilize the dc voltage. Here, the dc side may be regarded as a dc microgrid, which is mainly composed of a dc load, other converters, various renewable energy sources, and the like.
The method of the invention improves the dynamic and steady-state performance and the anti-interference capability of the three-phase NPC converter. The sliding mode control method of the three-level NPC converter is realized based on a direct-current voltage adjusting ring, an instantaneous power tracking ring and a voltage balancing ring, and the actual value v of the direct-current side voltage is adjusted by the direct-current voltage adjusting ringdcControl is performed so that the sum v of the DC-side capacitor voltagesdcAdjusted to corresponding desired values
Figure BDA0003126557740000081
Tracking the actual value p of the active power and the actual value q of the reactive power through an instantaneous power tracking loop, so that the active power p and the reactive power q accurately track respective reference values p*And q is*And the actual value e of the unbalanced voltage on the DC side is calculated by the voltage balance ringdcThe control is carried out, unbalanced voltages of two capacitors on the direct current side are enabled to be close to 0, a control signal is generated by combining the combined action of three rings to control the three-level NPC converter, the control process is simple, the direct current bus voltage is adjusted through the direct current voltage adjusting ring, the rapid dynamic response of the voltage step stage is ensured, and the fluctuation of the direct current side voltage caused when an uncertain interfered direct current side load is connected into the circuit can be effectively inhibited. The voltage balance of the direct-current side capacitor is realized, and the control stability is improved.
Further, in the present embodiment, the first and second substrates,
the method adopts a direct-current voltage regulating ring and utilizes the actual value v of the direct-current side voltage of the three-level NPC converterdcAnd a DC side voltage reference value
Figure BDA0003126557740000082
Obtaining an active power reference value p of a direct current side at the current sampling point moment through an Adaptive Sliding Mode Controller (ASMC) and a non-linear high-gain observer (NHGO)*The specific method comprises the following steps:
step A1, utilizing the actual value v of the DC side voltage of the three-level NPC converterdcAnd a DC side voltage reference value
Figure BDA0003126557740000083
Calculating the tracking error s of DC voltage regulation loopv(ii) a Wherein the content of the first and second substances,
Figure BDA0003126557740000084
step A2, tracking error s of DC voltage regulation loop through self-adaptive sliding mode controller ASMCvCorrecting and outputting the corrected tracking error
Figure BDA0003126557740000085
Step A3, adopting a nonlinear high-gain observer NHGO to perform on x in step A11And the active power reference value p at the last sampling point moment*Active power x to DC load2Observing to obtain the estimated value of the load power at the DC side
Figure BDA0003126557740000086
Step A4, correcting the corrected tracking error
Figure BDA0003126557740000087
And estimated value of DC side load power
Figure BDA0003126557740000088
Adding to obtain the active power reference value p at the current sampling point moment*。
Further, in the present embodiment, the first and second substrates,
in step a2, the dynamic equation of the adaptive sliding mode controller ASMC is:
Figure BDA0003126557740000089
wherein, KvAdaptive rate of adaptive sliding mode controller for gain of adaptive sliding mode controller ASMC, for a time variable
Figure BDA00031265577400000810
Comprises the following steps:
Figure BDA0003126557740000091
wherein, KlIs the gain change rate, K, of the adaptive sliding mode controller ASMCmIs a gain decision parameter of the adaptive sliding mode controller ASMC, baIs a gain gradient decision parameter of the adaptive sliding mode controller ASMC. Further, in the present invention, the first and second substrates,
in step a3, the dynamic equation of the nonlinear high-gain observer NHGO is:
Figure BDA0003126557740000092
Figure BDA0003126557740000093
wherein C is the capacitance of the DC side capacitor,
Figure BDA0003126557740000094
is a variable x1Is determined by the estimated value of (c),
Figure BDA0003126557740000095
is a variable x1An estimate of the derivative of (a) is,
Figure BDA0003126557740000096
active power x being a DC load2Is estimated from the derivative of (a)1Preceding stage gain parameter, alpha, of a non-linear high-gain observer NHGO2Post-stage gain parameter, epsilon, of a non-linear high-gain observer NHGO1And ε2Two gain parameters of a non-linear high-gain observer NHGO, and epsilon2Greater than epsilon1,bsAnd the gain judgment parameter of the non-linear high-gain observer NHGO is a saturation function sat (·).
In an embodiment, the basic control objective of the dc voltage regulation loop is to regulate the dc bus voltage to a specified value. In order to obtain the performances of quick transient response, insensitivity to external interference and the like, an ASMC control strategy based on NHGO is proposed for a voltage regulation loop. ASMC is called Adaptive sliding mode control in English, and Adaptive sliding mode control; the English language of NHGO is known as Nonlinear high-gain observer.
The control strategy of the adaptive sliding mode controller ASMC overcomes the compromise problem of buffeting and dynamic performance existing in the traditional Sliding Mode Control (SMC), reserves the quick dynamic performance of the traditional sliding mode control, and simultaneously weakens the buffeting phenomenon, so that the ASMC is applied to the field of power electronics and can further improve the performance of a converter. On the other hand, although ASMC can improve the robustness of the system, its ability to achieve interference cancellation is not sufficient due to the lack of interference information, which means that interference cannot be compensated for immediately to the controller. As a technique for observing states and disturbances, observers are suitable to compensate for this disadvantage of systems, such as Sliding Mode Observers (SMO) and Linear Extended State Observers (LESO). However, the conventional observer has a disadvantage of being very sensitive to noise, and thus has limited performance in practical applications. The nonlinear high-gain observer (NHGO) is an improved version of the high-gain observer, is not sensitive to noise, has strong disturbance rejection capability, and is very suitable for being applied to the field of power electronics. Therefore, in the embodiment, the interference elimination is realized by adopting a non-linear high-gain observer NHGO.
Further, in this embodiment, an instantaneous power tracking loop is adopted, and the active power reference value p on the dc side at the current sampling point is used*An active power actual value p, a reactive power actual value q and a preset reactive power reference value q*Acquiring the average duty cycle number delta of the three-level NPC converter through a second-order sliding mode controllerαβThe specific method comprises the following steps:
step B1, obtaining the active power reference value p at the current sampling point moment*Comparing with the actual value p of the active power to obtain the tracking error s of the active powerpSimultaneously, the actual reactive power value q at the current sampling point moment and a preset reactive power reference are obtainedValue q*Comparing to obtain the tracking error s of reactive powerq；
Step B2, tracking error s of active power through second-order sliding mode controller SOSMpAnd reactive power tracking error sqCorrecting to obtain the corrected active power tracking error
Figure BDA0003126557740000101
And the corrected reactive power tracking error
Figure BDA0003126557740000102
Step B3, tracking error s according to active powerpReactive power tracking error sqLast sampling point moment active power correction upAnd the reactive power correction u at the moment of the last sampling pointqUsing a high gain observer NHGO to correct the internal disturbance l caused by the uncertain parameters of the systempAnd lqObserving to obtain the estimated value of internal disturbance
Figure BDA0003126557740000103
And
Figure BDA0003126557740000104
wherein the correction amount u of active powerpAnd a reactive power correction uqIs 0;
step B4, according to the corrected tracking error
Figure BDA0003126557740000105
And
Figure BDA0003126557740000106
estimation of internal disturbances
Figure BDA0003126557740000107
And
Figure BDA0003126557740000108
updating the active power correction u at the current sampling point momentpAnd a reactive power correction uq；
Step B5, the real value p of the active power and the real value q of the reactive power are differentiated to obtain the derivative of the active power respectively
Figure BDA0003126557740000109
And reactive power derivative
Figure BDA00031265577400001010
Order to
Figure BDA00031265577400001011
Obtaining an average duty cycle of the equivalent point
Figure BDA00031265577400001012
Step B6, correcting quantity u according to active powerpAnd a reactive power correction amount uqAnd average duty cycle of equivalent point
Figure BDA00031265577400001013
Obtaining the average duty cycle deltaαβ。
Further, in the present embodiment, the first and second substrates,
in step B2, the dynamic equation of the second-order sliding mode controller SOSM is:
Figure BDA00031265577400001014
wherein k isi1And ki2Is the gain of the second order sliding mode controller SOSM, t is time.
Further, in the present embodiment, the first and second substrates,
in step B3, an estimate of the internal disturbance is obtained
Figure BDA00031265577400001015
And
Figure BDA00031265577400001016
the method is realized by the following formula:
Figure BDA00031265577400001017
Figure BDA00031265577400001018
and
Figure BDA0003126557740000111
Figure BDA0003126557740000112
wherein the content of the first and second substances,
Figure BDA0003126557740000113
vαand vβIs alpha component and beta component of converter AC side voltage in alpha beta coordinate system, L is AC side line inductance, alpha3Is a preceding-stage gain parameter, alpha, of a high-gain observer of an active power loop4Is a back-stage gain parameter, alpha, of the active power loop high-gain observer5Is the preceding-stage gain parameter, alpha, of the high-gain observer of the reactive power loop6Is a post-stage gain parameter, epsilon, of a reactive power loop high-gain observerpIs the gain, ε, of a high-gain observer of an active power loopqIs the gain of the reactive power loop high-gain observer.
In this embodiment, the control target to be realized is to control the actual active power value p and the actual reactive power value q so that they respectively track their reference values. The goal is achieved here using the SOSM control strategy to ensure fast response and steady state performance of the system. In addition, in order to improve the robustness of the system to the parameter perturbation, an HGO is designed to inhibit the influence of uncertain parameters on the system. The SOSM is called a Second-order sliding mode in English; the HGO is generally known in english as a High-gain observer.
Further, in the present embodiment, the first and second substrates,
in step B4, the real power correction u at the current sampling point timepAnd a reactive power correction uqComprises the following steps:
Figure BDA0003126557740000114
wherein the content of the first and second substances,
Figure BDA0003126557740000115
further, in the present embodiment, the first and second substrates,
in step B5, average duty ratio of equivalent point
Figure BDA0003126557740000116
Comprises the following steps:
Figure BDA0003126557740000117
wherein v isαβThe voltage of the AC side of the converter under an alpha beta coordinate system; j is a matrix, and
Figure BDA0003126557740000118
omega is the angular frequency of the grid voltage; l is an alternating current side wire inductor.
Further, in the present embodiment, the first and second substrates,
using a voltage balancing loop to measure the actual value e of the DC-side unbalanced voltagedcWith reference value of DC-side unbalanced voltage
Figure BDA0003126557740000119
Making difference, and performing PI regulation on the difference value of the two values to obtain a balance duty ratio deltabaComprises the following steps:
Figure BDA0003126557740000121
wherein k ispbThe proportional link gain of the PI controller is obtained; k is a radical ofibThe integral link gain of the PI controller is obtained; t is time.
In order to verify the superiority of the control strategy provided by the application, the sliding mode control method of the three-level NPC converter is compared with the traditional PI control strategy through experiments, wherein the sliding mode control strategy (LESO-SMC) is based on an extended state observer, and the sliding mode control strategy (NHGO-SMC) is based on a nonlinear high-gain observer. The parameters of the three-level NPC converter are shown in table I.
Table I experimental platform parameters
Figure BDA0003126557740000122
First, a test of dynamic performance was performed. Fig. 3a, 3b, 3c and 3d are dynamic response graphs when a voltage command is changed from 750V to 690V by using a PI controller, a LESO-SMC controller, a NHGO-SMC controller and a NHGO-ASMC controller proposed in the present application (i.e., a sliding mode control method of a three-level NPC converter according to the present invention), and fig. 4a, 4b, 4c and 4d are dynamic response graphs when a voltage command is changed from 690V to 750V, respectively. The results of the two experimental comparisons show that the dynamic response time is short, and the overshoot voltage is small.
Next, a test for steady state performance was performed. Connecting a DC load R1And R2And when the NPC operates stably, observing the quality of the three-phase alternating current under different control strategies. Fig. 5a, 5b, 5c and 5d are graphs of current harmonic spectra using a PI controller, a LESO-SMC controller, a NHGO-SMC controller, and a NHGO-ASMC controller as proposed in the present application, respectively, with Total Harmonic Distortion (THD) of 2.1%, 2.2%, 2.1% and 2.0%, respectively. Obviously, the current THD obtained by the NHGO-ASMC control strategy provided by the application is lower, and the current quality is better.
Finally, the disturbance resistance performance test is carried out. At R1The transient response of the DC bus voltage and phase a current when switched into the circuit is shown in FIG. 6, FIGS. 6a, 6b, 6c andFIG. 6d is a transient response using a PI controller, a LESO-SMC controller, a NHGO-SMC controller, and the NHGO-ASMC controller proposed herein, respectively. It can be observed that the method proposed by the present application achieves smaller voltage fluctuations and recovery times than other controllers. Therefore, according to the experimental results, compared with other control strategies, the NHGO-ASMC control strategy provided by the application has better anti-interference capability.
Although the invention herein has been described with reference to particular embodiments, it is to be understood that these embodiments are merely illustrative of the principles and applications of the present invention. It is therefore to be understood that numerous modifications may be made to the illustrative embodiments and that other arrangements may be devised without departing from the spirit and scope of the present invention as defined by the appended claims. It should be understood that features described in different dependent claims and herein may be combined in ways different from those described in the original claims. It is also to be understood that features described in connection with individual embodiments may be used in other described embodiments.

Claims (10)

1. A sliding mode control method of a three-level NPC converter is characterized by comprising the following steps:
the method adopts a direct-current voltage regulating ring and utilizes the actual value v of the direct-current side voltage of the three-level NPC converterdcAnd a DC side voltage reference value
Figure FDA0003126557730000011
Obtaining an active power reference value p of a direct current side at the current sampling point moment through an Adaptive Sliding Mode Controller (ASMC) and a non-linear high-gain observer (NHGO)*；
An instantaneous power tracking loop is adopted, and the active power reference value p of the current sampling point on the direct current side is utilized*An active power actual value p, a reactive power actual value q and a preset reactive power reference value q*Acquiring the average duty cycle number delta of the three-level NPC converter through a second-order sliding mode controllerαβ；
For three-level NPC converterMean duty cycle deltaαβPerforming alpha beta/abc coordinate transformation to obtain the average duty ratio delta 'of the three-level NPC converter in the abc coordinate system'abc；
Using a voltage balancing loop to measure the actual value e of the DC-side unbalanced voltagedcWith reference value of DC-side unbalanced voltage
Figure FDA0003126557730000012
Making difference, and performing PI regulation on the difference value of the two values to obtain a balance duty ratio deltaba；
For balanced duty cycle deltabaSum level NPC converter average duty cycle delta'abcAnd after addition, a control signal of a switching tube of the three-level NPC converter is obtained through a pulse width modulator, so that the three-level NPC converter is controlled.
2. The sliding-mode control method of the three-level NPC converter according to claim 1, characterized in that the actual value v of the DC-side voltage of the three-level NPC converter is utilized by using a DC voltage regulation loopdcAnd a DC side voltage reference value
Figure FDA0003126557730000013
Obtaining an active power reference value p of a direct current side at the current sampling point moment through an Adaptive Sliding Mode Controller (ASMC) and a non-linear high-gain observer (NHGO)*The specific method comprises the following steps:
step A1, utilizing the actual value v of the DC side voltage of the three-level NPC converterdcAnd a DC side voltage reference value
Figure FDA0003126557730000014
Calculating the tracking error s of DC voltage regulation loopv(ii) a Wherein the content of the first and second substances,
Figure FDA0003126557730000015
step A2, tracking error s of DC voltage regulation loop through self-adaptive sliding mode controller ASMCvCorrecting and outputting the corrected tracking errorDifference (D)
Figure FDA0003126557730000016
Step A3, adopting a nonlinear high-gain observer NHGO to perform on x in step A11And the active power reference value p at the last sampling point moment*Active power x to DC load2Observing to obtain the estimated value of the load power at the DC side
Figure FDA0003126557730000017
Step A4, correcting the corrected tracking error
Figure FDA0003126557730000018
And estimated value of DC side load power
Figure FDA0003126557730000019
Adding to obtain the active power reference value p at the current sampling point moment*。
3. The sliding-mode control method of the three-level NPC converter according to claim 3, wherein in step a2, the dynamic equation of the adaptive sliding-mode controller ASMC is:
Figure FDA0003126557730000021
wherein, KvAdaptive rate of adaptive sliding mode controller for gain of adaptive sliding mode controller ASMC, for a time variable
Figure FDA0003126557730000022
Comprises the following steps:
Figure FDA0003126557730000023
wherein, KlIs the gain change rate of the adaptive sliding mode controller ASMC,Kmis a gain decision parameter of the adaptive sliding mode controller ASMC, baIs a gain gradient decision parameter of the adaptive sliding mode controller ASMC.
4. The sliding-mode control method of the three-level NPC converter according to claim 3, wherein in step A3, the dynamic equation of the nonlinear high-gain observer NHGO is:
Figure FDA0003126557730000024
Figure FDA0003126557730000025
wherein C is the capacitance of the DC side capacitor,
Figure FDA0003126557730000026
is a variable x1Is determined by the estimated value of (c),
Figure FDA0003126557730000027
is a variable x1An estimate of the derivative of (a) is,
Figure FDA0003126557730000028
active power x being a DC load2Is estimated from the derivative of (a)1Preceding stage gain parameter, alpha, of a non-linear high-gain observer NHGO2Post-stage gain parameter, epsilon, of a non-linear high-gain observer NHGO1And ε2Two gain parameters of a non-linear high-gain observer NHGO, and epsilon2Greater than epsilon1 bsAnd the gain judgment parameter of the non-linear high-gain observer NHGO is a saturation function sat (·).
5. The sliding-mode control method of the three-level NPC converter according to claim 1 or 4, characterized in that an instantaneous power tracking loop is adoptedUsing the active power reference value p of the DC side at the current sampling point moment*An active power actual value p, a reactive power actual value q and a preset reactive power reference value q*Acquiring the average duty cycle number delta of the three-level NPC converter through a second-order sliding mode controllerαβThe specific method comprises the following steps:
step B1, obtaining the active power reference value p at the current sampling point moment*Comparing with the actual value p of the active power to obtain the tracking error s of the active powerpSimultaneously, the real reactive power value q at the current sampling point moment and a preset reactive power reference value q are obtained*Comparing to obtain the tracking error s of reactive powerq；
Step B2, tracking error s of active power through second-order sliding mode controller SOSMpAnd reactive power tracking error sqCorrecting to obtain the corrected active power tracking error
Figure FDA0003126557730000029
And the corrected reactive power tracking error
Figure FDA00031265577300000210
Step B3, tracking error s according to active powerpReactive power tracking error sqLast sampling point moment active power correction upAnd the reactive power correction u at the moment of the last sampling pointqUsing a high gain observer NHGO to correct the internal disturbance l caused by the uncertain parameters of the systempAnd lqObserving to obtain the estimated value of internal disturbance
Figure FDA0003126557730000031
And
Figure FDA0003126557730000032
wherein the correction amount u of active powerpAnd a reactive power correction uqIs 0;
step B4, according to the corrected tracking error
Figure FDA0003126557730000033
And
Figure FDA0003126557730000034
estimation of internal disturbances
Figure FDA0003126557730000035
And
Figure FDA0003126557730000036
updating the active power correction u at the current sampling point momentpAnd a reactive power correction uq；
Step B5, the real value p of the active power and the real value q of the reactive power are differentiated to obtain the derivative of the active power respectively
Figure FDA0003126557730000037
And reactive power derivative
Figure FDA0003126557730000038
Order to
Figure FDA0003126557730000039
Obtaining an average duty cycle of the equivalent point
Figure FDA00031265577300000310
Step B6, correcting quantity u according to active powerpAnd a reactive power correction amount uqAnd average duty cycle of equivalent point
Figure FDA00031265577300000311
Obtaining the average duty cycle deltaαβ。
6. The sliding-mode control method of the three-level NPC converter according to claim 5, wherein in step B2, the dynamic equation of the second-order sliding-mode controller SOSM is:
Figure FDA00031265577300000312
wherein k isi1And ki2Is the gain of the second order sliding mode controller SOSM, t is time.
7. The sliding-mode control method for the three-level NPC converter according to claim 6, characterized in that in step B3, an estimate of the internal disturbance is obtained
Figure FDA00031265577300000313
And
Figure FDA00031265577300000314
the method is realized by the following formula:
Figure FDA00031265577300000315
Figure FDA00031265577300000316
and
Figure FDA00031265577300000317
Figure FDA00031265577300000318
wherein the content of the first and second substances,
Figure FDA00031265577300000319
wherein v isαAnd vβIs alpha component and beta of converter AC side voltage in alpha beta coordinate systemComponent, L is AC side line inductance, α3Is a preceding-stage gain parameter, alpha, of a high-gain observer of an active power loop4Is a back-stage gain parameter, alpha, of the active power loop high-gain observer5Is the preceding-stage gain parameter, alpha, of the high-gain observer of the reactive power loop6Is a post-stage gain parameter, epsilon, of a reactive power loop high-gain observerpIs the gain, ε, of a high-gain observer of an active power loopqIs the gain of the reactive power loop high-gain observer.
8. The sliding-mode control method for the three-level NPC converter according to claim 7, wherein in step B4, the correction amount u of the active power at the current sampling point momentpAnd a reactive power correction uqComprises the following steps:
Figure FDA0003126557730000041
wherein the content of the first and second substances,
Figure FDA0003126557730000042
9. the sliding-mode control method of the three-level NPC converter as claimed in claim 8, wherein in step B5, the average duty cycle of the equivalent point
Figure FDA0003126557730000043
Comprises the following steps:
Figure FDA0003126557730000044
a computational implementation wherein vαβThe voltage of the AC side of the converter under an alpha beta coordinate system; j is a matrix, and
Figure FDA0003126557730000045
omega is the voltage of the power gridThe angular frequency of (d); l is an alternating current side wire inductor.
10. The sliding-mode control method of the three-level NPC converter according to claim 1 or 5, characterized in that the actual value e of the DC-side unbalanced voltage is adjusted by using a voltage balancing loopdcWith reference value of DC-side unbalanced voltage
Figure FDA0003126557730000046
Making difference, and performing PI regulation on the difference value of the two values to obtain a balance duty ratio deltabaComprises the following steps:
Figure FDA0003126557730000047
wherein k ispbThe proportional link gain of the PI controller is obtained; k is a radical ofibThe integral link gain of the PI controller is obtained; t is time.
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