CN112564566B - Method for expanding high-speed operation range of IMC-SPMSM system - Google Patents

Method for expanding high-speed operation range of IMC-SPMSM system Download PDF

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CN112564566B
CN112564566B CN202011522679.2A CN202011522679A CN112564566B CN 112564566 B CN112564566 B CN 112564566B CN 202011522679 A CN202011522679 A CN 202011522679A CN 112564566 B CN112564566 B CN 112564566B
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modulation
overmodulation
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CN112564566A (en
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阎彦
邹诺凡
史婷娜
张振
宋鹏
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Zhejiang University ZJU
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
    • H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H02P25/024—Synchronous motors controlled by supply frequency
    • H02P25/026—Synchronous motors controlled by supply frequency thereby detecting the rotor position
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2201/00—Indexing scheme relating to controlling arrangements characterised by the converter used
    • H02P2201/01—AC-AC converter stage controlled to provide a defined AC voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2205/00—Indexing scheme relating to controlling arrangements characterised by the control loops
    • H02P2205/01—Current loop, i.e. comparison of the motor current with a current reference
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2205/00—Indexing scheme relating to controlling arrangements characterised by the control loops
    • H02P2205/07—Speed loop, i.e. comparison of the motor speed with a speed reference
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

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Abstract

The invention discloses a method for expanding the high-speed operation range of an IMC-SPMSM system, which designs a rectification-level modulation strategy to ensure that the average value V of intermediate direct-current voltage output by IMCpnIn that
Figure DDA0002849680670000011
Figure DDA0002849680670000012
Within a range. The inversion stage adopts space vector modulation and minimum phase difference overmodulation in a linear modulation region and an overmodulation region respectively, and finally the IMC maximum voltage transmission ratio reaches 1 through proper combination of rectification and inversion two-stage modulation strategies; in the aspect of control strategy design, a voltage error feedback weak magnetic control method is combined, a rectification-stage overmodulation depth controller is designed, and rectification-stage voltage is increased when weak magnetic operation reaches a power limit.

Description

Method for expanding high-speed operation range of IMC-SPMSM system
Technical Field
The invention belongs to the field of control of motors, and relates to a method for expanding a high-speed operation range of an indirect matrix converter-permanent magnet synchronous motor (IMC-SPMSM) system, which aims to solve the problem of limited speed regulation range of the IMC-SPMSM system caused by the fact that the maximum voltage transmission ratio of IMC is only 0.866.
Background
Surface-Mounted Permanent Magnet Synchronous motors (spmmss) are widely used in the fields of white appliances, industrial automation, aerospace and the like because of their high power density. These applications typically require a wide range of system speeds. Therefore, when the power supply voltage is constant, field weakening control and overmodulation control are generally used to obtain the highest possible speed control range by making full use of the power supply voltage.
At present, flux weakening control strategies based on vector control can be roughly classified into three categories, namely, feedforward strategies, feedback strategies and hybrid strategies. The feedforward strategy usually uses a motor model to calculate the reference current value of the weak magnetic region as a feedforward compensation quantity, so that the response speed of the method is high, but the control effect is influenced by motor parameters. To improve system robustness, a variety of feedback strategies are proposed. Common feedback strategies are: voltage amplitude feedback, voltage error feedback before and after overmodulation, and effective vector action time feedback. The last two strategies can effectively utilize the voltage vector of the overmodulation region when the motor is in weak magnetic operation, so that the voltage utilization rate is high, and larger torque and higher rotating speed output can be realized. But the effective vector action time feedback strategy cannot reach six-pulse wave operation. The hybrid strategy results in a more complex control structure while achieving complementary advantages by adding a feedback loop to the feedforward path.
In recent years, an Indirect Matrix Converter (IMC) is used as a direct AC-AC power Converter, and has the advantages of no need of an intermediate energy storage link, sinusoidal input and output waveforms, adjustable input power factor and the like, so that the application research in the field of driving of motors, particularly SPMSM, is increasingly widespread. However, the IMC maximum linear Voltage Transfer Ratio (VTR) is only 0.866, and the utilization rate of the power supply Voltage is low, which makes it inferior in competition with the common ac-dc-ac converter. The overmodulation technology can improve VTR of IMC, but the overmodulation of a rectification stage and an inversion stage needs to be considered at the same time, and the overmodulation technology is relatively complex. Moreover, no energy storage device is arranged between the rectifying stage and the inverter stage of the IMC, which means that the amplitude of the input direct-current voltage of the inverter stage is always changed, and the difficulty of the SPMSM flux weakening control is increased.
Disclosure of Invention
The invention aims at expanding the high-speed operation range of an IMC-SPMSM system (namely expanding an indirect matrix converter-permanent magnet synchronous motor, the same below), and provides a wide-speed-domain weak magnetic control strategy suitable for the IMC-SPMSM system, which comprises the following aspects: firstly, applying an overmodulation strategy in a rectification stage and an inversion stage to promote VTR to 1; secondly, a voltage error feedback flux weakening control method is adopted, so that a voltage vector output by the system is positioned at a voltage limit boundary, and direct-current voltage output by a rectifier stage is fully utilized; and finally, the difference between the current set value and the current limit value is used as input, the modulation depth angle alpha of the rectifier stage is used as output, an overmodulation depth PI controller is designed, and the range of the speed regulating system is further widened by effectively controlling the overmodulation depth.
The technical scheme adopted by the invention is as follows:
a method for expanding the high-speed operation range of an IMC-SPMSM system, which comprises the following steps:
s1, constructing an IMC rectification level modulation strategy to enable the IMC intermediate direct current voltage average value VpnIn that
Figure BDA0002849680650000021
A linear change;
s2, adopting a space vector modulation strategy in a linear modulation region and adopting a minimum phase difference overmodulation strategy in an overmodulation region by inverter modulation;
s3, designing a flux-weakening controller based on a voltage error feedback method, constructing a value function by adopting a voltage error, and calculating optimal reference values of stator currents of a d axis and a q axis by using a gradient descent method to realize full utilization of IMC rectification-level output voltage in a flux-weakening operation area;
s4, designing a rectification-stage overmodulation depth PI controller according to a stator current reference vector IrefAmplitude of (I)refAnd current limit value IlimAnd adjusting the modulation depth angle alpha of the IMC rectification stage and increasing the voltage output of the rectification stage by the deviation, so that the speed regulation range of the IMC-SPMSM system is further widened and the transient performance is improved when the IMC-SPMSM system reaches the power limit value.
Preferably, the specific steps of S1 are as follows:
the three-phase input voltage of the IMC is balanced and symmetrical, and the expression is as follows:
Figure BDA0002849680650000022
in the formula, va、vb、vcFor IMC three-phase input voltage, VimAnd ωiThe amplitude and angular frequency of the input phase voltage are shown, and t represents time;
defining an instantaneous maximum v in a three-phase input voltageimaxMiddle value vimidAnd a minimum value viminComprises the following steps:
Figure BDA0002849680650000031
in the formula, max represents taking the maximum value, mid represents taking the middle value, and min represents taking the minimum value;
will omegait ∈ [0,2 π)) into 12 input voltage partitions of width π/6, i.e., [0, π/6 ], [0, π/3), … …, [11 π/6,2 π), each partition defining line voltage variables as follows:
Δvimax＝vimax-vimin (3)
Figure BDA0002849680650000032
defining IMC intermediate DC voltage instantaneous value as vpnV is then controlled by the on-off of the rectifier stage switching devicepn∈{Δvimax,Δvimid}; setting the switching period TsInner, vpn＝ΔvimaxAnd vpn＝ΔvimidRespectively at a time of TmaxAnd TmidWherein 0 is less than or equal to Tmax，Tmid≤Ts，Tmax+Tmid＝TsThen, the following two modulation modes are defined:
the modulation method 1 is as follows:
Figure BDA0002849680650000033
Figure BDA0002849680650000034
the modulation method 2 is as follows:
Tmax＝Ts,Tmid＝0 (7)
in the formula, k i1,2, …,12 is the input voltage partition number;
under the control of input side current as unit power factor, the direct current output v corresponding to the modulation mode 1 and the modulation mode 2pnAt TsThe average values in each case are
Figure BDA0002849680650000035
And
Figure BDA0002849680650000036
the calculation formula is as follows:
Figure BDA0002849680650000037
Figure BDA0002849680650000041
the method controls the time of the independent action of the modulation mode 1 and the modulation mode 2 in one IMC input voltage partition so as to enable the IMC intermediate direct-current voltage average value VpnIn that
Figure BDA0002849680650000042
Linearly changing.
Preferably, the specific steps of S2 are as follows:
in the linear modulation region, the IMC adopts a space vector modulation strategy; under the modulation strategy, for any one inputVoltage division with reference to output line voltage vectors
Figure BDA0002849680650000043
Is expressed by the vector composition of
Figure BDA0002849680650000044
In the formula (I), the compound is shown in the specification,
Figure BDA0002849680650000045
for two equivalent basic voltage vectors constituting an output voltage sector,
Figure BDA0002849680650000046
is a zero vector, dα、 dβ、d0Respectively represent
Figure BDA0002849680650000047
And
Figure BDA0002849680650000048
duty cycle of the action; according to the volt-second equilibrium principle, duty cycle dα、 dβThe expression of (a) is:
dα＝mvsin(π/3-θv),dβ＝mvsin(θv) (11)
in the formula, thetavThe included angle value between the voltage vector of the reference output line and the sector boundary where the vector is located; m isvIs the output voltage space vector modulation ratio, and
Figure BDA0002849680650000049
outputting line voltage vectors as references
Figure BDA00028496806500000410
An amplitude value;
Figure BDA00028496806500000420
outputting the average value of the intermediate direct current voltage for the IMC rectification stage;
when the IMC inverter stage overmodulation is operated,
Figure BDA00028496806500000411
endpoint exceedance
Figure BDA00028496806500000412
The formed regular hexagon adopts a minimum phase difference overmodulation strategy; under the modulation strategy, output line voltage vector
Figure BDA00028496806500000413
And vector
Figure BDA00028496806500000414
In phase and
Figure BDA00028496806500000415
the terminal is located with a base vector
Figure BDA00028496806500000416
The duty ratio d at the boundary of regular hexagon with vertex as terminalα、dβThe expression is as follows:
Figure BDA00028496806500000417
preferably, the specific steps of S3 are as follows:
the cost function is defined as follows:
Figure BDA00028496806500000418
in the formula,. DELTA.vd、ΔvqRespectively the d-axis voltage difference and the q-axis voltage difference before and after overmodulation, and the expression is
Figure BDA00028496806500000419
In the formula, vd *、vq *D and q axis voltage reference values output by the current regulator respectively; v. ofd、vqRespectively outputting d-axis voltage reference values and q-axis voltage reference values after overmodulation;
Figure BDA0002849680650000051
d and q axis current reference values respectively; lambda is the permanent magnetic flux linkage amplitude;
obtaining d and q axis current reference values introducing weak magnetic components by applying a gradient descent method aiming at the formula (14)
Figure BDA0002849680650000052
Figure BDA0002849680650000053
The expression is as follows:
Figure BDA0002849680650000054
in the formula, LsIs stator inductance, ωrIs the angular speed of the rotor, npIs the number of pole pairs of the motor, s is Laplace operator, omegacIs the cut-off frequency, omega, of a first-order low-pass filtercSetting as a current regulator bandwidth; beta is a coefficient influencing the transient performance and the speed regulation range of the system, and is set as follows:
Figure BDA0002849680650000055
in the formula, ωrmaxThe maximum operation speed which can be reached by the motor;
a voltage error feedback-flux weakening controller is constructed based on the formula (15), so that the output voltage vector of the system is positioned on a voltage limit boundary as much as possible, and the maximum fundamental voltage output by the over-modulation of the inverter level is fully utilized to widen the speed regulation range of the IMC-SPMSM system.
Preferably, in S4, in S4, the transfer function of the rectifier stage overmodulation depth PI controller is:
α(s)＝(Iref(s)-Ilim(s))(kpi+kii/s) (17)
in the formula, kpi、kiiProportional coefficient and integral coefficient of the rectification level overmodulation depth controller respectively; wherein IlimThe selection of (a) needs to satisfy two conditions: firstly, guarantee Ilim＜Imax(ii) a Secondly, IlimAnd ImaxThe difference between should be greater than the current harmonic peak caused by overmodulation.
Compared with the prior art, the technical scheme of the invention has the following beneficial effects:
the invention improves the fundamental voltage transmission ratio of IMC, and IMC inverter-level modulation fully utilizes the direct-current voltage output by the rectifier level, thereby realizing reasonable and smooth switching of the rectifier level under the over-modulation and non-over-modulation working conditions, widening the speed regulation range of the system and improving the transient performance of the IMC-SPMSM speed regulation system in a weak magnetic region.
Drawings
FIG. 1 is a schematic diagram of a wide-speed-domain operation strategy of an IMC-SPMSM system;
FIG. 2a is a schematic diagram of a rectified stage modulated input voltage division;
FIG. 2b shows the output of the rectifier stage
Figure BDA0002849680650000056
A waveform schematic diagram;
FIG. 3a is a composite diagram of an inverse spatial vector diagram and a reference voltage vector;
FIG. 3b is a vector trace diagram of output voltage of an inverter level overmodulation region (
Figure BDA0002849680650000061
And k ism＝2)；
FIG. 4a is a trace of the maximum output voltage vector endpoint;
FIG. 4b is a voltage current constraint and flux weakening zone current vector end point trace;
FIG. 5 is a simulation waveform of the ramp-up process from 1200r/min to 1500r/min, wherein: FIG. 5(a) is a rotation speed waveform, FIG. 5(b) is a torque waveform, and FIG. 5(c) is id、iqWave form, FIG. 5(d) Is an alpha-angle waveform, and v is shown in FIG. 5(e)pnWaveform, FIG. 5(f) is input current iaA waveform;
FIG. 6 is a comparison of performance with and without over-modulation depth control, where: FIG. 6a is a speed comparison, FIG. 6b is a torque comparison, and FIG. 6c is a stator current comparison;
FIG. 7 shows the simulation results of torque-rotation speed characteristic curves of the IMC-SPMSM system under different control strategies.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention clearer, the following describes the technical solutions of a preferred embodiment of the present invention in further detail with reference to the accompanying drawings:
1. wide-speed-range operation strategy structure for expanding indirect matrix converter-permanent magnet synchronous motor (IMC-SPMSM) system
The schematic structure of the wide-speed-range control strategy of the IMC-SPMSM system in this embodiment is shown in fig. 1. When the speed is below the base speed, the mode of Maximum Torque current ratio control (MTPA) is adopted, and the SPMSM is id0 vector control mode; the IMC rectification stage adopts a space vector modulation strategy without zero vectors, and the inverter modulation is the same as the two-level VSI space vector modulation. When the SPMSM operates at the basic speed, a weak magnetic control mode is adopted, a voltage error feedback-weak magnetic control strategy of the SPMSM system driven by two-level VSI is referred, and a voltage error feedback-weak magnetic control scheme under the IMC-SPMSM system is given by PART _1 in attached figure 1.
In order to expand the speed operation range of the IMC-SPMSM system as much as possible, the SPMSM is subjected to flux weakening control, and simultaneously, the IMC inverter stage and the IMC rectifier stage work in a linear modulation region or an overmodulation region according to operation requirements (rotating speed reference value and load torque condition). Specifically, when the output of the voltage error feedback-field weakening controller (PART _1) is less than 0, the SPMSM enters a field weakening state, and the modulation strategy of the inverter stage is switched to the minimum phase difference overmodulation strategy. Further, if the output of the rectification stage overmodulation depth controller (PART _2) is greater than 0, the rectification stage of IMC will also transition from linear modulation to overmodulation.
IMC-SPMSM system modulation method
2.1. Modulation of rectifier stage
The three-phase input voltage of IMC is balanced and symmetrical, and the IMC has the following components:
Figure BDA0002849680650000071
in the formula, va、vb、vcFor IMC three-phase input voltage, VimAnd ωiTo input the phase voltage amplitude and angular frequency, t represents time.
Defining an instantaneous maximum v in a three-phase input voltageimaxMiddle value vimidAnd a minimum value viminComprises the following steps:
Figure BDA0002849680650000072
in the formula, max represents a maximum value, mid represents a median value, and min represents a minimum value.
For ease of analysis, let ω beit ∈ [0,2 π)) into 12 input voltage partitions of width π/6, i.e., [0, π/6 ], [0, π/3 ]), … …, [11 π/6,2 π). According to v in the present embodimentimax、vimidAnd viminThe value changes, and the IMC input voltage is divided into 12 intervals, as shown in fig. 2 a.
The line voltage variables are defined in each partition as:
Δvimax＝vimax-vimin (3)
Figure BDA0002849680650000073
intermediate DC voltage v of IMC by on-off control of rectifying stage switching devicepn∈{Δvimax,Δvimid}. Setting the switching period TsInner, vpn＝ΔvimaxAnd vpn＝ΔviminRespectively at a time of TmaxAnd Tmid(0≤Tmax,Tmid≤Ts， Tmax+Tmid＝Ts) Then, the following two modulation modes are defined:
modulation system 1 (hereinafter referred to as CASE 1):
Figure BDA0002849680650000074
Figure BDA0002849680650000081
modulation system 2 (hereinafter referred to as CASE 2):
Tmax＝Ts,Tmid＝0 (7)
in the formula, k i1,2, …,12 are input voltage division numbers corresponding to (r), (c), … … in fig. 2
Figure BDA0002849680650000082
Under the control of input side current as unit power factor, the direct current output v corresponding to the modulation mode 1 and the modulation mode 2pnAt TsThe average values in each case are
Figure BDA0002849680650000083
And
Figure BDA0002849680650000084
the calculation formula is as follows:
Figure BDA0002849680650000085
Figure BDA0002849680650000086
the corresponding waveform is shown in fig. 2b, from which it can be seen that,
Figure BDA0002849680650000087
and
Figure BDA0002849680650000088
has periodicity. Therefore, an IMC rectification-level modulation strategy can be constructed by utilizing two modulation modes of CASE 1 and CASE2 and controlling the time of independent action of CASE 1 and CASE in an IMC input voltage partition, so that the average value V of the IMC intermediate direct-current voltage is enabled to be VpnIn that
Figure BDA0002849680650000089
Linear variation, in particular:
taking the interval (r) as an example, as shown in FIG. 2b, when ω isiWhen t belongs to [0, pi/6-alpha), CASE 1 is adopted for rectification level modulation, and at the moment
Figure BDA00028496806500000810
And
Figure BDA00028496806500000811
overlapping; when ω isiWhen t belongs to [ pi/6-alpha, pi/6), CASE2 is adopted for rectification level modulation, and at the moment
Figure BDA00028496806500000812
And
Figure BDA00028496806500000813
and (4) overlapping. According to the area equivalence principle, the method comprises the following steps:
Figure BDA00028496806500000814
substituting formula (8) and formula (9) for formula (10) to obtain:
Figure BDA00028496806500000815
from the formula (11), when α ∈ [0, π/6 ]]Time, average value of DC voltage output by rectifier stage
Figure BDA00028496806500000816
In particular, when α is 0rad, only CASE 1 modulation is adopted; when alpha is pi/6 rad, only CASE2 modulation mode is adopted.
2.2. Inverse level modulation
For the IMC inverter stage, the output line voltages corresponding to the allowed 8 switch states are written into a space vector form, and 6 effective vectors V with fixed space positions and variable amplitude can be obtained1～V6And 2 zero vectors (collectively denoted as V)0). The active vector divides the complex plane into 6 sectors as shown in figure 3 a. In the figure, the position of the upper end of the main shaft,
Figure BDA0002849680650000091
for the initial phase angle, omega, of the output voltageoFor outputting fundamental voltage angular frequency, input angular frequency ratio omegaiI.e. km＝ωo/ωi。
Due to vpnThe value of (A) varies with the modulation process of the rectifier stage, so that the effective vector VlThe amplitude of (c) also changes. Let when vpnAre respectively Δ vimax、ΔvimidWhen, VlAre each Vl_max、Vl_midDefining the equivalent base voltage vector as:
Figure BDA0002849680650000092
in the formula (I), the compound is shown in the specification,
Figure BDA0002849680650000093
as vectors
Figure BDA0002849680650000094
Of a current amplitude of
Figure BDA0002849680650000095
And l is 1,2, … and 6 is the sector number of the output voltage.
In the linear modulation region, the IMC adopts a space vector modulation strategy. Under the modulation strategy, vectorThe vector is located in the interval (r) in FIG. 2a, for example, when referring to the output line voltage vector
Figure BDA0002849680650000096
When in the interval (r), the two equivalent basic voltage vectors forming the sector
Figure BDA0002849680650000097
And zero vector
Figure BDA0002849680650000098
Synthesis in the synthetic relationship
Figure BDA0002849680650000099
In the formula (d)α、dβAnd d0Respectively represent vectors
Figure BDA00028496806500000910
And zero vector
Figure BDA00028496806500000911
Duty cycle of the action. For the interval (i) of the first time,
Figure BDA00028496806500000912
the two equivalent basic voltage vectors with phase angles of-pi/6 and pi/6 in the complex plane respectively, and the rest intervals can be adjusted correspondingly.
According to the volt-second balance principle, the expression of the obtained duty ratio is as follows
dα＝mvsin(π/3-θv),dβ＝mvsin(θv) (14)
In the formula, thetavIs the angle value between the voltage vector of the reference output line and the sector boundary where the vector is located, and theta is the angle value of the interval phivE [0, pi/3), as shown in FIG. 3 a; m isvIs the output voltage space vector modulation ratio, and
Figure BDA00028496806500000913
Figure BDA00028496806500000914
outputting line voltage vectors as references
Figure BDA00028496806500000915
An amplitude value;
Figure BDA00028496806500000916
the average value of the intermediate dc voltage is output for the IMC rectification stage.
For any one input voltage partition, it can calculate the reference output line voltage vector by equation (13) above
Figure BDA00028496806500000917
Reference line voltage vector when IMC inverter level overmodulation operation
Figure BDA0002849680650000101
Beyond the base vector
Figure BDA0002849680650000102
The terminal is a regular hexagon with vertexes, and at the moment, the IMC inverse modulation is carried out by adopting a minimum phase difference overmodulation strategy. In this modulation mode, voltage vector is output
Figure BDA0002849680650000103
And vector
Figure BDA0002849680650000104
In phase and
Figure BDA0002849680650000105
the terminal is located with a base vector
Figure BDA0002849680650000106
On the boundary of regular hexagon with vertex as terminal (note: hexagon radius)
Figure BDA0002849680650000107
Time-variant) as shown in figure 3 b. The duty ratio expression under the minimum phase difference overmodulation strategy is as follows:
Figure BDA0002849680650000108
will be in the complex plane
Figure BDA0002849680650000109
Is converted into a time domain waveform, as shown in the right side v of fig. 3bABShown in the waveform, the expression is as follows:
Figure BDA00028496806500001010
due to the fact that
Figure BDA00028496806500001011
The amplitude is in periodic variation, so that Fourier transformation is carried out on the output of the inverter stage, and the difficulty in obtaining the amplitude of the fundamental wave of the output voltage is high. Therefore, it is used here
Figure BDA00028496806500001012
Instead of the former
Figure BDA00028496806500001013
To find vABThe fundamental voltage amplitude approximation of (a) is expressed as:
Figure BDA00028496806500001014
to be provided with
Figure BDA00028496806500001015
A regular hexagon terminating in vertices is shown in dashed lines in fig. 3 b. Let vector
Figure BDA00028496806500001016
And
Figure BDA00028496806500001017
in the same phase, the phase of the signal is changed,
Figure BDA00028496806500001018
the terminals being located on imaginary lines, to be in the complex plane
Figure BDA00028496806500001019
Is converted into a time-domain waveform, as shown in the figure
Figure BDA00028496806500001020
Shown in the waveform, the expression is as follows:
Figure BDA00028496806500001021
the above equation is expanded into a fourier series form,
Figure BDA00028496806500001022
the fundamental amplitude approximation can be expressed as:
Figure BDA00028496806500001023
defining the voltage transmission ratio q as the fundamental amplitude V of the output phase voltageomAmplitude V of voltage of input phaseimThe ratio of (a) to (b), namely:
q＝Vom/Vim (20)
by combining the equations (8), (9), (19) and (20), the inverter stage adopts minimum phase difference modulation, and the rectifier stage adopts CASE 1 and CASE2 modulation modes, so that the voltage transmission ratio can reach 0.954 and 1, respectively.
Based on the above analysis, the IMC modulation region can be divided into three parts according to the voltage transfer ratio, namely, a linear modulation region, an overmodulation I region and an overmodulation II region, specifically:
(1) when q is more than or equal to 0 and less than or equal to 0.866, the IMC is in a linear modulation region, at the moment, the rectification stage only adopts a CASE 1 modulation mode, and the inverter stage adopts a traditional space vector modulation method;
(2) when q is more than 0.866 and less than or equal to 0.954, the IMC is in an overmodulation I region, at the moment, the rectification stage only adopts a CASE 1 modulation mode, but the inversion stage adopts a minimum phase difference overmodulation method;
(3) when q is more than 0.954 and less than or equal to 1, the IMC is in an overmodulation II region, at the moment, a CASE 1 and CASE2 combined modulation mode is adopted in a rectification stage, and a minimum phase difference overmodulation method is adopted in an inversion stage.
IMC-SPMSM system weak magnetic control strategy
In a rotating coordinate system with the rotor magnetic field oriented, the SPMSM stator voltage equation and the electromagnetic torque equation can be expressed as:
Figure BDA0002849680650000111
Figure BDA0002849680650000112
in the formula, vdAnd vqD and q axis stator voltages, respectively; i.e. idAnd iqD and q axis stator currents respectively; rsIs a stator resistor; l issIs a stator inductance; omegarIs the rotor angular velocity; p is a differential operator; lambda is the permanent magnetic flux linkage amplitude; n ispThe number of pole pairs of the motor is; t iseIs an electromagnetic torque.
Considering the influences of factors such as IMC capacity, input voltage, modulation mode and motor thermal rating, the SPMSM quadrature-direct axis voltage and current need to meet the following constraint conditions:
Figure BDA0002849680650000113
in the formula, VmaxIs the maximum voltage applied to the stator windings of the motor; i ismaxThe maximum current passed to the winding.
IMC under linear modulation, Vmax＝0.866Vim(ii) a After entering the overmodulation region, VmaxInstantaneous valueAs a function of output voltage phase angle. According to the space vector modulation principle, when the zero vector action time is 0, the maximum output voltage vector V under the phase position can be obtainedout_maxThe moving locus of the vector end point in the complex plane forms an IMC voltage boundary, and the amplitude expression is as follows:
Figure BDA0002849680650000121
v is shown by the combination of formula (8), formula (9) and formula (24)out_maxAngle with input power factor
Figure BDA0002849680650000122
Initial phase angle of output voltage
Figure BDA0002849680650000123
Output-to-input angular frequency ratio kmAnd the rectification stage modulation mode. FIG. 4a shows
Figure BDA0002849680650000124
And is
Figure BDA0002849680650000125
In three different CASEs (CASE 1 modulation and k)m1.5, CASE 1 modulation and km1.5, CASE2 modulation and km2), maximum output voltage vector Vout_max1、Vout_max2And Vout_max3The travel path of the end point in the complex plane.
Neglecting the stator resistance voltage drop, the equation (21) to (24) is used in id-iqThe voltage and current limit constraints are plotted in the coordinate system as shown in fig. 4 b. The current limit constraint is represented by a current limit circle centered at the origin; the voltage limit constraint is expressed as a point (-lambda/L)s0) voltage limit boundary with center, the radius of the inscribed circle of the voltage limit boundary is smaller as the rotating speed of the motor is higher, and the shapes of the voltage boundary are different under different rotating speeds. In FIG. 4b,. omega.0＜ω1Taking the CASE of the rectifier stage adopting CASE 1 modulation mode, when the rotor angular velocity is ω0Time pairThe voltage limit boundary shape is the (1) case of FIG. 4 a; angular speed of rotor is omega1The corresponding voltage limit boundary shape is the (2) case in fig. 4 a.
Under the weak magnetic control of the IMC-SPMSM system, the voltage limit boundary is very complex, and the boundary shape changes along with the output frequency (namely the motor rotating speed), thereby bringing difficulty to the design of a weak magnetic control loop.
The field weakening controller of the strategy is constructed based on a voltage error feedback-field weakening controller, the structure of the controller is shown as PART _1 in figure 1, and an input signal is the difference value of a q-axis voltage reference value before and after overmodulation; the output signal is used to regulate the d-axis current of the motor.
The motor is restrained by voltage limit after entering flux weakening operation, and the IMC inverter stage is in a minimum phase difference overmodulation state at the moment. Neglecting stator resistance voltage drop, the d and q axis voltage difference before and after overmodulation can be expressed as:
Figure BDA0002849680650000126
in the formula (I), the compound is shown in the specification,
Figure BDA0002849680650000131
d and q axis voltage reference values output by the current regulator respectively; v. ofd、vqRespectively outputting d-axis voltage reference values and q-axis voltage reference values after overmodulation;
Figure BDA0002849680650000132
d and q axis current reference values are respectively, and lambda is a permanent magnetic flux linkage amplitude.
To minimize voltage error, a cost function is defined:
Figure BDA0002849680650000133
substituting formula (25) for formula (26) and deriving i by gradient descent methodd、iqIs determined. For SPMSM, the electricity required for operational control is satisfiedUnder the condition that the output torque of the motor is not changed,
Figure BDA0002849680650000134
the variation should be 0. The method comprises the following specific steps:
Figure BDA0002849680650000135
in the formula (I), the compound is shown in the specification,
Figure BDA0002849680650000136
is an improved current reference value; beta is a constant greater than zero.
Integrating the two sides of equation (27) yields:
Figure BDA0002849680650000137
note: s in parentheses all represent laplacian, the same below.
As can be seen from the formula (28), if Δ vqAs soon as it is greater than 0, due to the integration effect,
Figure BDA0002849680650000138
cannot be recovered to
Figure BDA0002849680650000139
(i.e., the system cannot exit the flux weakening operating region), and for this purpose, a low-pass filter is used instead of the integrator, the modified current reference value expression is:
Figure BDA00028496806500001310
in the formula, ωcThe cut-off frequency of a first order low pass filter.
In general, ω iscSetting as current regulator bandwidth, i.e. ωc＝ki/kpWherein k isi＝Rsωc、kp＝Lsωc. The coefficient beta affects the transient performance and the speed regulation range of the system: the larger beta is, the stronger the magnetic flux weakening capability is, and the transient current regulation performance is better; however, the greater β, the lower the system maximum output torque or maximum operating speed. To make the dimensions of the current expression the same before and after the improvement of equation (29), β may be set as:
Figure BDA00028496806500001311
in the formula, ωrmaxThe maximum operating speed achievable by the motor.
Based on the voltage error feedback-flux weakening controller, the flux weakening operation working points of the motor are analyzed as follows: as shown in FIG. 4b, assume that the load torque is T0The motor is accelerated from below the basic speed to omega0The motor operating point enters a weak magnetic region from the point A and follows the T0The curve moves until point B.
For point B, it is the intersection of the voltage limit boundary and the constant torque line, but at id-iqThe voltage limit boundary in the coordinate system is defined by npωrThe rotation is performed for the angular velocity, resulting in point B being a moving point. From the derivation process, the voltage error feedback-flux weakening controller constructed based on the formula (29) can enable the system output voltage vector to be on the voltage limit boundary as much as possible, so that the maximum fundamental voltage output by the inverter stage overmodulation is effectively utilized to widen the speed regulation range of the IMC-SPMSM system.
IMC rectification level overmodulation depth controller design
During operation of the motor, when the stator current vector end point is located at the intersection of the current limit circle and the voltage limit boundary, as shown at point C in fig. 4b, the system will reach the power limit under the IMC output voltage and current constraints. In this state, if the current-level overmodulation depth is further adjusted on the basis of the flux-weakening control, the system speed-adjusting range is expected to be further widened, and the transient performance is favorably improved. Around the above thought, the invention designs the IMC rectification stage overmodulation depth controller. The design is explained as follows.
The IMC rectification stage overmodulation depth controller structure is shown as PART _2 in FIG. 1. The input signal to the controller is a stator current reference vector IrefIs expressed as:
Figure BDA0002849680650000141
and a current limit value Ilim。IrefAnd IlimThe difference value of (2) is regulated by PI, and an IMC rectification-stage modulation depth angle alpha is output, namely the transfer function of the rectification-stage overmodulation depth PI controller is as follows:
α(s)＝(Iref(s)-Ilim(s))(kpi+kii/s) (32)
in the formula, kpi、kiiProportional coefficient and integral coefficient of the rectification level overmodulation depth controller are respectively.
A key variable in overmodulation depth controller design is the limiting current IlimIs selected to satisfy two conditions, firstly, I is ensuredlim＜Imax(ii) a Secondly, IlimAnd ImaxThe difference between should be greater than the current harmonic peak caused by overmodulation. For the second point, the correlation analysis is explained as follows:
when the IMC is in the overmodulation II region, the voltage transmission ratio q can reach 1, and the current harmonic output by the IMC is most abundant. The rectification stage adopts a CASE2 modulation mode to carry out v of IMCpnPerforming fourier expansion, one can obtain:
Figure BDA0002849680650000151
wherein n is 1,2,3 ….
Considering only the fundamental wave, then:
Figure BDA0002849680650000152
the inverter stage being in a minimum phase difference over-modulation stateWhile the inverted output voltage vector, i.e. the stator voltage vector, moves along the voltage boundary, assuming vpnBeing constant, the stator voltage vector magnitude can be expressed as:
Figure BDA0002849680650000153
in the formula, a is a fluctuation coefficient of the stator voltage in the overmodulation region.
Substitution of formula (34) for formula (35) can give:
Figure BDA0002849680650000154
from equation (36), the harmonic frequency of the stator current caused by overmodulation is mainly 6 ωo、6ωi、6(ωo-ωi) And 6(ω)o+ωi)。
Neglecting the stator resistance, the relationship between stator current and stator voltage in steady state can be obtained from equation (21):
Figure BDA0002849680650000155
at a frequency of 6 omegaiFor example, neglecting the stator inductance variation, the stator current harmonic is calculated as:
Figure BDA0002849680650000161
in the formula (I), the compound is shown in the specification,
Figure BDA0002849680650000162
is the included angle between the stator voltage vector and the d-axis,
Figure BDA0002849680650000167
at a frequency of 6 omegaiThe resulting current harmonics.
In the same way, 6 omega can be obtainedo、6(ωo-ωi) And 6(ω)o+ωi) Generated stator current harmonics
Figure BDA0002849680650000163
And
Figure BDA0002849680650000164
thus, it is possible to provide
Figure BDA0002849680650000165
After the rectifying-stage overmodulation depth control is put into operation, the current vector is controlled on a current limiting circle; if IlimAnd ImaxThe difference between the two is smaller than the peak value of the current harmonic, the current harmonic easily causes the angle alpha to reach the amplitude limiting value, namely the regulator is saturated, so that the regulation capacity is lost, and the alpha is always equal to pi/6.
5. Simulation verification
The effect of the invention has been verified by digital simulation. The simulation parameters are set as follows: the three-phase input voltage is 380V/50 Hz; the IMC output current limit value is 15A; the inductance, resistance and capacitance of the input filter are respectively 0.8mH, 20 omega and 20 muF; IMC switching frequency is 10 kHz; the SPMSM parameters are shown in the table below. The parameters of the current regulator and the speed regulator are set according to an internal model principle, wherein the bandwidth of a current loop is 3000 rad/s.
Figure BDA0002849680650000166
FIG. 5 shows the simulation result of the system in the process of accelerating the load torque to 10 N.m and the motor speed from 1200r/min to 1500 r/min. It can be seen from the figure that when the speed is increased to the weak magnetic region, the IMC rectification stage works in a CASE I modulation mode, the inverter stage works in a minimum phase difference overmodulation mode, the maximum fundamental voltage amplitude which can be output is increased when linear modulation is carried out, and the current loop regulation capacity is increased, so that the actual rotating speed average value can well follow a given value, and the overmodulation strategy adopted by the scheme of the invention is verified to be effective.
FIG. 6 shows that only voltage error feedback control and voltage error inverse control are adoptedComparing simulation results of the feedback control and the overmodulation depth control, wherein the operation working conditions of the motor are as follows: when the load torque is 10N m, the acceleration is from 1500r/min to 1750 r/min. Wherein n is1、Te1And Is1Respectively a rotating speed waveform, a torque waveform and a stator current waveform when only voltage error feedback control exists; in the same way, n2、Te2And Is2The voltage error feedback control and the overmodulation depth control represent a rotating speed waveform, a torque waveform and a stator current waveform respectively. The waveform in the figure shows that n is n in the process of accelerating the motor from 1500r/min to 1750r/min2Ratio n1The stability is reached more quickly; after stabilization, Te2Ratio Te2The amplitude of the vibration is greater. In this process, the stator current value is greater than the current limit value IlimThe overmodulation depth control is always in working state and the stator current Is2Is regulated at IlimLeft and right. Therefore, when the overmodulation depth control is adopted, the voltage margin of the system is increased, so that the dynamic process is accelerated. But correspondingly the system harmonic currents will increase.
Fig. 7 shows the maximum torque output by the system at different speeds. In the figure, the curves a, b and c respectively represent "idThe torque-rotation speed characteristic curves of the control system are 0, only the voltage error feedback control and the control system adopting the voltage error feedback control and the overmodulation depth control. The curve in the figure shows that under the same load torque, the rotating speed value corresponding to the curve b is about 30% greater than that corresponding to the curve a, and on the basis of the rotating speed value corresponding to the curve c, the rotating speed value corresponding to the curve b is increased by about 5%. Therefore, the method effectively widens the speed regulation range of the IMC-SPMSM system at high speed.
The above-described embodiments are merely preferred embodiments of the present invention, which should not be construed as limiting the invention. Various changes and modifications may be made by one of ordinary skill in the pertinent art without departing from the spirit and scope of the present invention. Therefore, the technical scheme obtained by adopting the mode of equivalent replacement or equivalent transformation is within the protection scope of the invention.

Claims (4)

1. A method for expanding the high-speed operation range of an IMC-SPMSM system is characterized by comprising the following steps:
s1, constructing an IMC rectification level modulation strategy to enable the average value V of the IMC intermediate direct current voltage to be VpnIn that
Figure FDA0003496857540000011
A linear change;
s2, adopting a space vector modulation strategy in a linear modulation region and adopting a minimum phase difference overmodulation strategy in an overmodulation region by inverter modulation;
s3, designing a flux-weakening controller based on a voltage error feedback method, constructing a value function by adopting a voltage error, and calculating optimal reference values of stator currents of a d axis and a q axis by using a gradient descent method to realize full utilization of IMC rectification-level output voltage in a flux-weakening operation area;
s4, designing a rectification-stage overmodulation depth PI controller according to a stator current reference vector IrefAmplitude of (I)refAnd current limit value IlimThe IMC-SPMSM system further widens the speed regulation range of the system and improves the transient performance when the IMC-SPMSM system reaches a power limit value by adjusting the modulation depth angle alpha of the IMC rectification stage and increasing the voltage output of the rectification stage;
the specific steps of S1 are as follows:
the three-phase input voltage of the IMC is balanced and symmetrical, and the expression is as follows:
Figure FDA0003496857540000012
in the formula, va、vb、vcFor IMC three-phase input voltage, VimAnd ωiThe amplitude and angular frequency of the input phase voltage are shown, and t represents time;
defining an instantaneous maximum v in a three-phase input voltageimaxMiddle value vimidAnd a minimum value viminComprises the following steps:
Figure FDA0003496857540000013
in the formula, max represents taking the maximum value, mid represents taking the middle value, and min represents taking the minimum value;
will omegait ∈ [0,2 π)) into 12 input voltage partitions of width π/6, i.e., [0, π/6 ], [0, π/3), … …, [11 π/6,2 π), each partition defining line voltage variables as follows:
Δvimax＝vimax-vimin (3)
Figure FDA0003496857540000021
defining IMC intermediate DC voltage instantaneous value as vpnV is then controlled by the on-off of the rectifier stage switching devicepn∈{Δvimax,Δvimid}; setting the switching period TsInner, vpn＝ΔvimaxAnd vpn＝ΔvimidRespectively at a time of TmaxAnd TmidWherein 0 is less than or equal to Tmax，Tmid≤Ts，Tmax+Tmid＝TsThen, the following two modulation modes are defined:
the modulation method 1 is as follows:
Figure FDA0003496857540000022
Figure FDA0003496857540000023
the modulation method 2 is as follows:
Tmax＝Ts,Tmid＝0 (7)
in the formula, ki1,2, …,12 is the input voltage partition number;
modulation mode 1 and modulation mode under the control of input side current as unit power factor2 corresponding to the DC output vpnAt TsThe average values in each case are
Figure FDA0003496857540000024
And
Figure FDA0003496857540000025
the calculation formula is as follows:
Figure FDA0003496857540000026
Figure FDA0003496857540000027
the method controls the time of the independent action of the modulation mode 1 and the modulation mode 2 in one IMC input voltage partition so as to enable the IMC intermediate direct-current voltage average value VpnIn that
Figure FDA0003496857540000028
Linearly changing.
2. The method for expanding the high-speed operation range of the IMC-SPMSM system according to claim 1, wherein the specific steps of S2 are as follows:
in the linear modulation region, the IMC adopts a space vector modulation strategy; under the modulation strategy, for any one input voltage partition, the vector of the voltage of the reference output line is
Figure FDA0003496857540000031
Is expressed by the vector composition of
Figure FDA0003496857540000032
In the formula (I), the compound is shown in the specification,
Figure FDA0003496857540000033
for two equivalent basic voltage vectors constituting an output voltage sector,
Figure FDA0003496857540000034
is a zero vector, dα、dβ、d0Respectively represent
Figure FDA0003496857540000035
And
Figure FDA0003496857540000036
duty cycle of the action; according to the volt-second equilibrium principle, duty cycle dα、dβThe expression of (a) is:
dα＝mvsin(π/3-θv),dβ＝mvsin(θv) (11)
in the formula, thetavThe included angle value between the voltage vector of the reference output line and the sector boundary where the vector is located; m isvIs the output voltage space vector modulation ratio, and
Figure FDA0003496857540000037
Figure FDA0003496857540000038
outputting line voltage vectors as references
Figure FDA0003496857540000039
An amplitude value;
Figure FDA00034968575400000310
outputting the average value of the intermediate direct current voltage for the IMC rectification stage;
when the IMC inverter stage overmodulation is operated,
Figure FDA00034968575400000311
endpoint exceedance
Figure FDA00034968575400000312
The formed regular hexagon adopts a minimum phase difference overmodulation strategy; under the modulation strategy, output line voltage vector
Figure FDA00034968575400000313
And vector
Figure FDA00034968575400000314
In phase and
Figure FDA00034968575400000315
the terminal is located with a base vector
Figure FDA00034968575400000316
The duty ratio d at the boundary of regular hexagon with vertex as terminalα、dβThe expression is as follows:
Figure FDA00034968575400000317
3. the method for expanding the high-speed operation range of the IMC-SPMSM system as claimed in claim 2, wherein the specific steps of S3 are as follows:
the cost function is defined as follows:
Figure FDA00034968575400000318
in the formula,. DELTA.vd、ΔvqRespectively the d-axis voltage difference and the q-axis voltage difference before and after overmodulation, and the expression is
Figure FDA00034968575400000319
In the formula, vd *、vq *D and q axis voltage reference values output by the current regulator respectively; v. ofd、vqRespectively outputting d-axis voltage reference values and q-axis voltage reference values after overmodulation;
Figure FDA00034968575400000320
d and q axis current reference values respectively; lambda is the permanent magnetic flux linkage amplitude;
obtaining d and q axis current reference values introducing weak magnetic components by applying a gradient descent method aiming at the formula (14)
Figure FDA0003496857540000041
Figure FDA0003496857540000042
The expression is as follows:
Figure FDA0003496857540000043
in the formula, LsIs stator inductance, ωrIs the angular speed of the rotor, npIs the number of pole pairs of the motor, s is Laplace operator, omegacIs the cut-off frequency, omega, of a first-order low-pass filtercSetting as a current regulator bandwidth; beta is a coefficient influencing the transient performance and the speed regulation range of the system, and is set as follows:
Figure FDA0003496857540000044
in the formula, ωrmaxThe maximum operation speed which can be reached by the motor;
a voltage error feedback-flux weakening controller is constructed based on the formula (15), so that the output voltage vector of the system is positioned on a voltage limit boundary as much as possible, and the maximum fundamental voltage output by the over-modulation of the inverter level is fully utilized to widen the speed regulation range of the IMC-SPMSM system.
4. The method for extending the high-speed operation range of an IMC-SPMSM system of claim 3 where in S4 the transfer function of the rectifier stage overmodulation depth PI controller is:
α(s)＝(Iref(s)-Ilim(s))(kpi+kii/s) (17)
in the formula, kpi、kiiProportional coefficient and integral coefficient of the rectification level overmodulation depth controller respectively; wherein IlimThe selection of (a) needs to satisfy two conditions: firstly, guarantee Ilim<Imax(ii) a Secondly, IlimAnd ImaxThe difference between should be greater than the current harmonic peak caused by overmodulation.
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Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN102064760A (en) * 2009-11-18 2011-05-18 上海三菱电梯有限公司 Method for controlling commutation according to voltage type commutation strategy for matrix converter
JP2011109886A (en) * 2009-11-20 2011-06-02 Diamond Electric Mfg Co Ltd Indirect matrix converter
CN102594242A (en) * 2012-02-10 2012-07-18 北方工业大学 Vector control method based on indirect matrix converter multi-machine transmission system
JPWO2010116865A1 (en) * 2009-04-06 2012-10-18 ナブテスコ株式会社 Indirect matrix converter, swivel control device for work machine, and work machine
CN106330034A (en) * 2016-06-14 2017-01-11 北方工业大学 Model predictive control method for simplified sampling of indirect matrix converter

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPWO2010116865A1 (en) * 2009-04-06 2012-10-18 ナブテスコ株式会社 Indirect matrix converter, swivel control device for work machine, and work machine
CN102064760A (en) * 2009-11-18 2011-05-18 上海三菱电梯有限公司 Method for controlling commutation according to voltage type commutation strategy for matrix converter
JP2011109886A (en) * 2009-11-20 2011-06-02 Diamond Electric Mfg Co Ltd Indirect matrix converter
CN102594242A (en) * 2012-02-10 2012-07-18 北方工业大学 Vector control method based on indirect matrix converter multi-machine transmission system
CN106330034A (en) * 2016-06-14 2017-01-11 北方工业大学 Model predictive control method for simplified sampling of indirect matrix converter

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
基于双级矩阵变换器的永磁同步电机调速系统的研究;胥鹏飞;《中国优秀博硕士学位论文全文数据库(硕士)工程科技Ⅱ辑》;20200715;全文 *

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