CN112519877A - Method for realizing motor drive controller of electric steering wheel - Google Patents
Method for realizing motor drive controller of electric steering wheel Download PDFInfo
- Publication number
- CN112519877A CN112519877A CN202011318469.1A CN202011318469A CN112519877A CN 112519877 A CN112519877 A CN 112519877A CN 202011318469 A CN202011318469 A CN 202011318469A CN 112519877 A CN112519877 A CN 112519877A
- Authority
- CN
- China
- Prior art keywords
- current
- calculating
- speed
- initializing
- circuit
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Pending
Links
Images
Classifications
-
- B—PERFORMING OPERATIONS; TRANSPORTING
- B62—LAND VEHICLES FOR TRAVELLING OTHERWISE THAN ON RAILS
- B62D—MOTOR VEHICLES; TRAILERS
- B62D5/00—Power-assisted or power-driven steering
- B62D5/04—Power-assisted or power-driven steering electrical, e.g. using an electric servo-motor connected to, or forming part of, the steering gear
- B62D5/0457—Power-assisted or power-driven steering electrical, e.g. using an electric servo-motor connected to, or forming part of, the steering gear characterised by control features of the drive means as such
- B62D5/046—Controlling the motor
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
- H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Chemical & Material Sciences (AREA)
- Combustion & Propulsion (AREA)
- Transportation (AREA)
- Mechanical Engineering (AREA)
- Control Of Motors That Do Not Use Commutators (AREA)
- Control Of Ac Motors In General (AREA)
Abstract
The invention discloses a method for realizing a motor drive controller of an electric steering wheel, belonging to the field of automatic driving, aiming at solving the technical problem of controlling the high-precision action of the electric steering wheel by the operation of a drive motor so as to realize automatic driving, and adopting the technical scheme that: the method comprises the steps that a main control chip DSP sends an instruction to a frameless motor control driver through a digital communication bus, a control circuit in the frameless motor control driver receives the instruction, a servo motor magnetic field orientation vector control system is adopted to calculate three-phase voltage of a servo motor according to current, speed and position information of the servo motor, and then PWM signals are output; a grid driving power supply in the frameless motor control driver supplies power to the driving circuit, and the driving circuit amplifies the PWM signal to drive a power tube MOSFET in the frameless motor control driver, so that low-noise and high-efficiency energy conversion is realized.
Description
Technical Field
The invention relates to the technical field of automatic driving, in particular to a method for realizing a motor drive controller of an electric steering wheel.
Background
With the vigorous development of the unmanned driving technology of vehicles and the development of advanced servo control systems, people have higher and higher requirements on the overall working performance of the steering engine, and the steering engine is promoted to develop towards the direction that the volume and the mass are continuously reduced, the bearing capacity is continuously enhanced, and the control performance is continuously improved. The steering engine is an important actuating mechanism of a vehicle unmanned steering wheel control system, the quality of the performance of the steering engine directly influences the dynamic quality of the vehicle control system, and the steering engine has the advantages of simplicity, reliability, good manufacturability, convenience in use and maintenance, single energy source, low cost, easiness in control and the like, is widely concerned and deeply researched, and is widely applied to the field of vehicle control systems.
The automatic steering wheel that the existing market was used all adopts hydraulic pressure principle to drive and turns to the hydraulic pump, realizes the rotation of wheel, and the hydraulic pressure scheme installation can change the hydraulic circuit of whole vehicle, changes the structure of former car, causes very big difficulty to the after-sale warranty of later stage vehicle to it is very troublesome to install. Should market demand, electronic steering wheel has appeared, and on-vehicle former dress storage battery power supply need not to change former car structure, only need steering wheel below installation a driving motor can, but how to move the high accuracy action of control electronic steering wheel through driving motor operation, and then realize that autopilot is the present technical problem who awaits a urgent solution.
Disclosure of Invention
The technical task of the invention is to provide a method for realizing a motor drive controller of an electric steering wheel, which solves the problem of how to control the high-precision action of the electric steering wheel by the operation of a drive motor so as to realize automatic driving.
The technical task of the invention is realized according to the following mode, the method for realizing the electric steering wheel motor drive controller is characterized in that a main control chip DSP sends an instruction to a frameless motor control driver through a digital communication bus, a control circuit in the frameless motor control driver receives the instruction, a servo motor magnetic field orientation vector control system is adopted to calculate the three-phase voltage of a servo motor according to the current, speed and position information of the servo motor, and then the three-phase voltage is output through a PWM signal; a grid driving power supply in the frameless motor control driver supplies power to a driving circuit, and the driving circuit amplifies a PWM signal to drive a power tube MOSFET in the frameless motor control driver, so that low-noise and high-efficiency energy conversion is realized;
the frameless motor control driver (MCU) is in data transmission with the motion controller through the CAN communication circuit, and the battery pack supplies power to the frameless motor control driver.
Preferably, the servo motor field orientation vector control system is specifically as follows:
measuring stator current i output by an inverter through a current sensorA、iBConverted into digital quantity by A/D converter of main control chip DSP and calculated iCThe formula is as follows:
iC＝-(iA+iB)；
(II) converting the current i through Clarke conversion and Park conversionA、iB、iCConversion into a direct current component i in a rotating coordinate systemsq、isd，isq、isdAs the negative feedback quantity of the current loop;
(III) measuring the mechanical displacement theta of the motor by using a linear grating rulermAnd converts it into an electrical angle thetaeAnd a speed n; wherein the electrical angle thetaeFor participating in the computation of the Park transform and inverse transform; the speed n is used as the negative feedback quantity of the speed loop;
(IV) setting the speed nrefThe deviation from the speed feedback quantity n is passed through a speed PI regulator, the output of which is taken as a current q-axis reference component i for torque controlsqref；
(V), isqrefAnd isdref(equal to zero) and current feedback quantity isa、isdRespectively output Odq phase voltage components V of the rotating coordinate system through a current PI regulatorsqrefAnd Vsdref；
VI, VsqrefAnd VsdrefThen through Park inverseComponent V of stator phase voltage vector of transformation conversion type O alpha beta rectangular coordinate systemsarefAnd Vsbref；
(VII) component V of stator phase voltage vectorsares、VsbrefAnd when the number of the sectors is known, a PWM control signal is generated by utilizing a voltage space vector SVPWM technology to control the inverter.
Preferably, the frameless motor control driver is powered by a power conversion circuit, and the power conversion circuit comprises a bus voltage + 5V-12V power module and a 5V-3.3V power module; the bus voltage is converted into a +5V and 12V power module through a DC/DC conversion chip, and then is converted into 3.3V from 5V by a voltage conversion chip;
wherein, 3.3V is used for supplying power for the DSP of the main control chip and peripheral circuits thereof;
+5V is used for communication of the CAN communication circuit and power supply of the current adopting and processing circuit;
12V is used for power supply of the driving circuit.
Preferably, the current adoption and processing circuit adopts an ACS712 current sensor IC, the ACS712 current sensor IC detects the magnitude of the current through a Hall effect and outputs a voltage value with 2.5V as a reference, and the voltage value of Vout is divided by a high-precision resistor; after passing through the diode, the digital signal enters an AD acquisition pin of the DSP of the main control chip for AD conversion, and the diode plays a role in protection.
Preferably, a current protection circuit is arranged in the frameless motor control driver, and when the current peak value in the servo motor winding exceeds the rated current of the power tube MOSFET, namely the set value of the comparator LM339 is reached, the current protection circuit outputs a low level signal Fault signal to the Fault synthesis circuit, triggers to generate a high level signal to the tri-state output bus receiver, acts to output a turn-off signal, so that the power switch is turned off, and the power switch tube is protected from being damaged.
Preferably, the CAN communication circuit adopts a high-speed real-time bus, sends a given position and a given speed to the frameless motor control driver, controls the rotation of the servo motor and realizes the motion function of the joint; the method specifically comprises the following steps:
according to the motion required by one joint or a plurality of joints, the upper computer calculates the real-time running speed and position of each servo motor on the joint according to kinematics, the real-time requirement is high, and real-time data is transmitted to the corresponding servo motor through a high-speed real-time bus.
Preferably, the CAN communication circuit adopts a model SN65HVD230 produced by TI as a receiving and sending chip of a local area network, and the power supply of the SN65HVD230 is 3.3V;
the CAN communication circuit adopts a TVS (bidirectional transient voltage suppressor) as a CAN bus protector for protection and fault tolerance, so that a CAN transceiver in the CAN communication circuit is prevented from being influenced by EMI and ESD;
the diodes in the CAN communication circuit are in bidirectional configuration, so that clamping of a long cable system to normal data line signals due to common-mode voltage imbalance is prevented.
Preferably, the main control chip DSP is initialized specifically as follows:
(1) starting a main program of the main control chip DSP;
(2) initializing a system;
(3) clearing protection and error data;
(4) initializing hardware;
(5) initializing the functional module: the method comprises the steps of initializing a PWM module, initializing AD conversion, initializing an interrupt module, initializing an SCI module, initializing a CAN module and initializing an I/O port module;
(6) starting the linear grating ruler function;
(7) detecting and defining a phase current zero point;
(8) initializing the position of a motor rotor;
(9) initializing functional variables of the computer;
(10) enabling the interrupt;
(11) and judging whether to wait for interruption.
Preferably, the servo motor field orientation vector control system comprises a timer underflow interrupt submodule, which is specifically as follows:
s1, protecting the field;
s2, current sampling;
s3, sampling position difference;
s4, digital filtering;
s5, calculating the position of the rotor;
s6, refreshing SCI communication data;
s7, refreshing communication data of the CAN communication circuit:
s8, judging whether an outer closed loop exists:
if yes, go to step S9;
if not, jumping to step S11;
s9, position PID control;
s10, speed PI control;
s11, phase current processing;
s12, Clarke transformation;
s13, solving sim and cos values;
s14, Park conversion;
s15, adjusting q-axis current PI;
s16, adjusting the d-axis current PI;
s17, performing Park inverse transformation;
s18, obtaining a sector;
s19, obtaining T0、T1And T2；
And S20, updating data by the PWM comparator.
More preferably, the speed PI control in step S10 is specifically as follows:
s10-1, adjusting the speed PI;
s10-2, setting n for reading speed*；
S10-3, calculating the deviation e (kt) ═ n*-n；
S10-4, calculating np(kT)＝Kp[e(kt)-e(kT-T)]；
S10-5, calculating ni(kT)＝Ki e(kt)；
S10-6, calculating delta (kT) ═ np(kT)+ni(kT)；
S10-7, processing to prevent integral saturation;
s10-8, according to n (kT)Q-axis current give iq *；
The q-axis current PI adjustment in step S15 is specifically as follows:
s15-1, adjusting q-axis current PI;
s15-2, read Current given iq *；
S15-3, calculating the deviation e (kt) iq *-iq；
S15-4, calculating ip(kT)＝Kp[e(kt)-e(kT-T)]；
S15-5, calculating ii(kT)＝Ki e(kt)；
S15-6, calculating Δ (kT) ═ ip(kT)+ii(kT)；
S15-7, processing to prevent integral saturation;
s15-8, giving q-axis voltage given V according to i (kT)q *；
The d-axis current PI adjustment in step S16 is specifically as follows:
s16-1, adjusting d-axis current PI;
s16-2, read Current given id *；
S16-3, calculating the deviation e (kt) id *-id；
S16-4, calculating ip(kT)＝Kp[e(kt)-e(kT-T)]；
S16-5, calculating ii(kT)＝Ki e(kt)；
S16-6, calculating Δ (kT) ═ ip(kT)+ii(kT)；
S16-7, processing to prevent integral saturation;
s16-8, giving q-axis voltage given V according to i (kT)d *；
The sector is obtained in step S18 specifically as follows:
s18-1, calculating a sector;
s18-2, sector P ═ 0;
S18-3、P1＝sign(Vsβ)；
S18-4、P2＝2sign[sin60°Vsα-sin30°Vsβ]；
S18-5、P3＝4sign[-sin60°Vsα-sin30°Vsβ]；
S18-6、P＝P1+P2+P3；
s18-7, determining a SECTOR SECTOR by table lookup;
the PWM comparator update data in step S20 is specifically as follows:
s20-1, calculating SVPWM;
s20-2, reading a SECTOR number SECTOR;
s20-3, reading the inverse matrix table by table lookup, and calculating C0、C1And C2；
S20-4, calculating T1＝0.5C1；
S20-5, calculating T2＝0.5C2；
S20-6, calculating T0＝0.25C0；
S20-7, calculating a PWM comparator value;
and S20-8, updating the PWM comparator value.
The realization method of the electric steering wheel motor driving controller has the following advantages:
the invention adopts DSP produced by TI company as main control chip to realize motor control algorithm. A power MOSFET is selected as a power device, so that low-noise and high-efficiency energy conversion is realized;
the invention adopts advanced vector control algorithm to realize the accurate control of the controller on the torque and the rotating speed of the motor, perfect protection functions of undervoltage, overvoltage, overcurrent and the like, and improve the reliability of the system;
the CAN communication circuit is adopted to transmit data with other or motion controllers, so that the transmission rate, flexibility and reliability of the data are improved;
the invention adopts an absolute encoder to acquire the speed signal of the motor, transmits the information to a master control through an RS232 communication mode, processes data, is unique at each position of the absolute encoder, is not lost when power is down, and has strong anti-interference capability;
the servo motor is adopted as an actuating mechanism of the vehicle unmanned control system to improve the dynamic response performance and the control accuracy of the system; the servo motor adopts a magnetic encoder for feedback, a Hall element is not arranged, the space is saved, the problem of repair caused by damage of the Hall element is solved, and the driver has the function of automatically driving the motor to make a change position after being electrified;
the invention also has the following characteristics:
firstly, an inheritance principle is adopted, and a motor servo driver sufficiently inherits the research results of the conventional low-power driving key technology in design;
secondly, adopting a derating design according to a reliability principle;
selecting components, raw materials and standard components in an optimal catalog as much as possible, selecting civil components with good performance and low cost, and reducing the design cost;
fourthly, the principle of light weight and miniaturization reduces the weight and the volume of the product through the optimized design, improves the maintainability of the product and reduces the cost.
Drawings
The invention is further described below with reference to the accompanying drawings.
FIG. 1 is a schematic diagram of a servo motor control driver;
FIG. 2 is a schematic block diagram of a power conversion;
FIG. 3 is a schematic diagram of a voltage conversion circuit;
FIG. 4 is a schematic diagram of a current sampling and processing circuit;
FIG. 5 is a schematic diagram of a current protection circuit;
FIG. 6 is a schematic diagram of a CAN communication circuit;
FIG. 7 is a flow chart of an initialization process of the DSP system;
FIG. 8 is a schematic diagram of a servo motor field oriented vector control system;
FIG. 9 is a block flow diagram of a timer underflow interrupt submodule;
FIG. 10 is a block diagram of a process for sector acquisition;
FIG. 11 is a schematic block diagram of the PWM comparator update data;
FIG. 12 is a flow chart of speed PI control;
FIG. 13 is a block flow diagram of q-axis current regulation;
FIG. 14 is a block flow diagram of d-axis current regulation.
Detailed Description
The method for implementing the electric steering wheel motor drive controller of the present invention will be described in detail below with reference to the drawings and specific embodiments.
Example (b):
as shown in fig. 1, the method for implementing the electric steering wheel motor drive controller of the present invention includes that a main control chip DSP sends an instruction to a frameless motor control driver through a digital communication bus, a control circuit in the frameless motor control driver receives the instruction, calculates a three-phase voltage of a servo motor by using a servo motor magnetic field orientation vector control system according to current, speed and position information of the servo motor, and outputs the three-phase voltage through a PWM signal; a grid driving power supply in the frameless motor control driver supplies power to a driving circuit, and the driving circuit amplifies a PWM signal to drive a power tube MOSFET in the frameless motor control driver, so that low-noise and high-efficiency energy conversion is realized; the frameless motor control driver (MCU) is in data transmission with the motion controller through the CAN communication circuit, and the battery pack supplies power to the frameless motor control driver.
A typical steering engine servo motor control driver mainly comprises a power tube bridge circuit, a power tube driving circuit, a grid driving power supply, a control circuit, communication and the like.
The frameless motor control driver in the embodiment adopts a power conversion circuit for power supply, and the power conversion circuit comprises a bus voltage +5V and 12V power module and a 5V-3.3V power module; the bus voltage is converted into +5V and 12V power modules through a DC/DC conversion chip, and then is converted into 3.3V from 5V by the voltage conversion chip, as shown in figure 2; wherein, 3.3V is used for supplying power for the DSP of the main control chip and peripheral circuits thereof; +5V is used for communication of the CAN communication circuit and power supply of the current adopting and processing circuit; 12V is used for power supply of the driving circuit.
Wherein 3.3V provides the IO port and the core for the DSP28035, the invention selects the chip LD1117ADT33TR dedicated to UTC corporation to provide the required power voltage for 28035, such as the voltage conversion circuit shown in fig. 3.
The current employing and processing circuitry in this embodiment employs an ACS712 current sensor IC, which ACS712 current sensor IC is a precise, economical solution to ac-dc sensing in industrial, commercial and communication systems. Small packages are well suited for applications with narrow space, and also reduce cost due to the reduced area of the circuit board. Typical application areas include motor control, load detection and management, switched mode power supplies and over-current fault protection.
The device has a precise, low-offset linear hall sensor circuit and is provided with copper conductive paths near the wafer surface. The applied current through the copper conductive path can generate a magnetic field that can be induced by the integrated hall IC and converted to a proportional voltage. The accuracy of the device is optimized by the proximity of the magnetic field to the hall sensor. The precise proportional voltage is provided by a low-bias stable chopping BiCMOS hall IC, which comprises an Allegro patented digital temperature compensation device, and can realize ultra-precise performance based on temperature.
The output of the device has a positive slope as the current through the primary copper conductor path (from pins 1 and 2 to pins 3 and 4) increases, which path is used for current sensing. Typical values for this conductive path internal resistance are 0.65m Ω (packaged with LC SOIC 8) and 0.85m Ω (packaged with MA SOIC 16W), both of which have low power consumption characteristics.
The terminals of the conductive path are electrically isolated from the sensor leads. The ACS712 current sensor can thus be used in high-level current sensing applications without the use of high-side differential amplifiers or other expensive isolation techniques. This is particularly true for motor control.
The ACS712 current sensor IC detects the magnitude of current through Hall effect, outputs a voltage value with 2.5V as reference, and divides the voltage value of Vout through a high-precision resistor; after passing through the diode, the signal enters an AD acquisition pin of the DSP of the main control chip for AD conversion, and the diode plays a role in protection, as shown in figure 4.
In the frameless motor control driver in this embodiment, a current protection circuit is provided, which is designed to prevent an element from being damaged by an excessive current, and the implementation circuit is shown in fig. 5. When the current peak value in the servo motor winding exceeds the rated current of the power tube MOSFET, namely the set value of the comparator LM339 is reached, the current protection circuit outputs a low level signal Fault signal to the Fault comprehensive circuit, triggers to generate a high level signal to the tri-state output bus receiver, acts to output a turn-off signal, and turns off the power switch, so that the power switch tube is protected, and a power device is prevented from being damaged.
The CAN communication circuit in the embodiment adopts a high-speed real-time bus, sends a given position and a given speed to the frameless motor control driver, controls the rotation of the servo motor and realizes the motion function of the joint; the method specifically comprises the following steps:
according to the motion required by one joint or a plurality of joints, the upper computer calculates the real-time running speed and position of each servo motor on the joint according to kinematics, the real-time requirement is high, and real-time data is transmitted to the corresponding servo motor through a high-speed real-time bus.
Because the CAN port of the upper computer outputs differential signals, the DSP adopts TTL level in communication. Therefore, the problem of level conversion exists in the communication between the upper computer and the DSP. To solve this problem, the CAN communication circuit uses model SN65HVD230 produced by TI as a receiving and transmitting chip of the lan, which has the following characteristics:
(1) can have a transmission speed of 1 Mbaud;
(2) the bus has the function of protecting transient and can reduce the interference of frequency;
(3) strong capability of resisting EMI (electromagnetic interference);
(4) thermal protection, short circuit protection to battery and ground;
(5) multiple nodes may be connected 110.
The power supply of the SN65HVD230 is 3.3V, and the power supply of the SN65HVD230 is also 3.3V. The electrical connection between the main control chip and the CAN chip is ensured, and the stability and reliability of the whole circuit are ensured; as shown in fig. 6, the CAN communication circuit uses TVS (bidirectional transient voltage suppressor) as a CAN bus protector for protection and fault tolerance, so that the CAN transceiver in the CAN communication circuit is protected from EMI and ESD; the diodes in the CAN communication circuit are in bidirectional configuration, so that clamping of a long cable system to normal data line signals due to common-mode voltage imbalance is prevented.
The main control chip DSP initialization in this embodiment includes machine frequency, function module enable, watchdog setting, and the like. The power protection error is cleared because the hardware is designed with under-voltage protection and the like, the protection signal generates hardware interruption and enables the corresponding PWM output pin to be in a high resistance state, and therefore the power protection error signal is cleared before the program runs. The measured deviation current is used for calculating the true value of the measured current, because the A/D converter adopted by the system is unipolar, but the measured current conversion signal is signed, so that the signed current value is converted into the unsigned current value by the related circuit in the hardware part, and the unsigned current value is not required to be converted into the signed stator current output value in the program. Therefore, the offset current needs to be measured, namely, the A/D sampling value at the time of zero current is obtained, and then the A/D sampling value of the offset current is subtracted from the A/D sampling value every time, so that the signed real current value can be obtained. As shown in fig. 7, the initialization of the main control chip DSP is specifically as follows:
(1) starting a main program of the main control chip DSP;
(2) initializing a system;
(3) clearing protection and error data;
(4) initializing hardware;
(5) initializing the functional module: the method comprises the steps of initializing a PWM module, initializing AD conversion, initializing an interrupt module, initializing an SCI module, initializing a CAN module and initializing an I/O port module;
(6) starting the linear grating ruler function;
(7) detecting and defining a phase current zero point;
(8) initializing the position of a motor rotor;
(9) initializing functional variables of the computer;
(10) enabling the interrupt;
(11) and judging whether to wait for interruption.
As shown in fig. 8, the servo motor field orientation vector control system is embodied as follows:
measuring stator current i output by an inverter through a current sensorA、iBConverted into digital quantity by A/D converter of main control chip DSP and calculated iCThe formula is as follows:
iC＝-(iA+iB)；
(II) converting the current i through Clarke conversion and Park conversionA、iB、iCConversion into a direct current component i in a rotating coordinate systemsq、isd，isq、isdAs the negative feedback quantity of the current loop;
(III) measuring the mechanical displacement theta of the motor by using a linear grating rulermAnd converts it into an electrical angle thetaeAnd a speed n; wherein the electrical angle thetaeFor participating in the computation of the Park transform and inverse transform; the speed n is used as the negative feedback quantity of the speed loop;
(IV) setting the speed nrefThe deviation from the speed feedback quantity n is passed through a speed PI regulator, the output of which is taken as a current q-axis reference component i for torque controlsqref；
(V), isqrefAnd isdref(equal to zero) and current feedback quantity isa、isdRespectively output Odq phase voltage components V of the rotating coordinate system through a current PI regulatorsqrefAnd Vsdref；
VI, VsqrefAnd VsdrefAnd then, the component V of the stator phase voltage vector of the orthogonal coordinate system of the Park inverse transformation conversion formula O alpha beta is usedsarefAnd Vsbref；
(VII) component V of stator phase voltage vectorsares、VsbrefAnd when the number of the sectors is known, a PWM control signal is generated by utilizing a voltage space vector SVPWM technology to control the inverter.
As shown in fig. 9, the servo motor field orientation vector control system in this embodiment includes a timer underflow interrupt submodule, which is specifically as follows:
s1, protecting the field;
s2, current sampling;
s3, sampling position difference;
s4, digital filtering;
s5, calculating the position of the rotor;
s6, refreshing SCI communication data;
s7, refreshing communication data of the CAN communication circuit:
s8, judging whether an outer closed loop exists:
if yes, go to step S9;
if not, jumping to step S11;
s9, position PID control;
s10, speed PI control;
s11, phase current processing;
s12, Clarke transformation;
s13, solving sim and cos values;
s14, Park conversion;
s15, adjusting q-axis current PI;
s16, adjusting the d-axis current PI;
s17, performing Park inverse transformation;
s18, obtaining a sector;
s19, obtaining T0、T1And T2；
And S20, updating data by the PWM comparator.
As shown in fig. 12, the speed PI control in step S10 in the present embodiment is specifically as follows:
s10-1, adjusting the speed PI;
s10-2, setting n for reading speed*；
S10-3, calculating the deviation e (kt) ═ n*-n；
S10-4, calculating np(kT)＝Kp[e(kt)-e(kT-T)]；
S10-5, calculating ni(kT)＝Ki e(kt)；
S10-6, calculating delta (kT) ═ np(kT)+ni(kT)；
S10-7, processing to prevent integral saturation;
s10-8, giving q-axis current given i according to n (kT)q *；
As shown in fig. 13, the q-axis current PI adjustment in step S15 in the present embodiment is as follows:
s15-1, adjusting q-axis current PI;
s15-2, read Current given iq *；
S15-3, calculating the deviation e (kt) iq *-iq；
S15-4, calculating ip(kT)＝Kp[e(kt)-e(kT-T)]；
S15-5, calculating ii(kT)＝Ki e(kt)；
S15-6, calculating Δ (kT) ═ ip(kT)+ii(kT)；
S15-7, processing to prevent integral saturation;
s15-8, giving q-axis voltage given V according to i (kT)q *；
As shown in fig. 14, the d-axis current PI adjustment in step S16 in this embodiment is as follows:
s16-1, adjusting d-axis current PI;
s16-2, read Current given id *；
S16-3, calculating the deviation e (kt) id *-id；
S16-4, calculating ip(kT)＝Kp[e(kt)-e(kT-T)]；
S16-5, calculating ii(kT)＝Ki e(kt)；
S16-6, calculating Δ (kT) ═ ip(kT)+ii(kT)；
S16-7, processing to prevent integral saturation;
s16-8, giving q-axis voltage given V according to i (kT)d *；
As shown in fig. 10, the sector obtained in step S18 in this embodiment is specifically as follows:
s18-1, calculating a sector;
s18-2, sector P ═ 0;
S18-3、P1＝sign(Vsβ)；
S18-4、P2＝2sign[sin60°Vsα-sin30°Vsβ]；
S18-5、P3＝4sign[-sin60°Vsα-sin30°Vsβ]；
S18-6、P＝P1+P2+P3；
s18-7, determining a SECTOR SECTOR by table lookup;
as shown in fig. 11, the PWM comparator update data in step S20 in this embodiment is as follows:
s20-1, calculating SVPWM;
s20-2, reading a SECTOR number SECTOR;
s20-3, reading the inverse matrix table by table lookup, and calculating C0、C1And C2；
S20-4, calculating T1＝0.5C1；
S20-5, calculating T2＝0.5C2；
S20-6, calculating T0＝0.25C0；
S20-7, calculating a PWM comparator value;
and S20-8, updating the PWM comparator value.
Finally, it should be noted that: the above embodiments are only used to illustrate the technical solution of the present invention, and not to limit the same; while the invention has been described in detail and with reference to the foregoing embodiments, it will be understood by those skilled in the art that: the technical solutions described in the foregoing embodiments may still be modified, or some or all of the technical features may be equivalently replaced; and the modifications or the substitutions do not make the essence of the corresponding technical solutions depart from the scope of the technical solutions of the embodiments of the present invention.
Claims (10)
1. A method for realizing a motor drive controller of an electric steering wheel is characterized in that a main control chip DSP sends an instruction to a frameless motor control driver through a digital communication bus, a control circuit in the frameless motor control driver receives the instruction, a servo motor magnetic field orientation vector control system is adopted to calculate the three-phase voltage of a servo motor according to the current, speed and position information of the servo motor, and then the three-phase voltage is output through a PWM signal; a grid driving power supply in the frameless motor control driver supplies power to a driving circuit, and the driving circuit amplifies a PWM signal to drive a power tube MOSFET in the frameless motor control driver, so that low-noise and high-efficiency energy conversion is realized;
the frameless motor control driver is in data transmission with the motion controller through the CAN communication circuit, and the battery pack supplies power to the frameless motor control driver.
2. The method of claim 1, wherein the servo motor field orientation vector control system is specifically as follows:
measuring stator current i output by an inverter through a current sensorA、iBConverted into digital quantity by A/D converter of main control chip DSP and calculated iCThe formula is as follows:
iC＝-(iA+iB)；
(II) converting the current i through Clarke conversion and Park conversionA、iB、iCConversion into a direct current component i in a rotating coordinate systemsq、isd，isq、isdAs the negative feedback quantity of the current loop;
(III) measuring the mechanical displacement theta of the motor by using a linear grating rulermAnd converts it into an electrical angle thetaeAnd a speed n; wherein the electrical angle thetaeFor participating in the computation of the Park transform and inverse transform; the speed n is used as the negative feedback quantity of the speed loop;
(IV) setting the speed nrefThe deviation from the speed feedback quantity n is passed through a speed PI regulator, the output of which is taken as a current q-axis reference component i for torque controlsqref；
(V), isqrefAnd isdrefAnd a current feedback quantity isa、isdRespectively output Odq phase voltage components V of the rotating coordinate system through a current PI regulatorsqrefAnd Vsdref；
(VI) and VsqrefAnd VsdrefAnd then, the component V of the stator phase voltage vector of the orthogonal coordinate system of the Park inverse transformation conversion formula O alpha beta is usedsarefAnd Vsbref；
(VII) component V of stator phase voltage vectorsares、VsbrefAnd when the number of the sectors is known, a PWM control signal is generated by utilizing a voltage space vector SVPWM technology to control the inverter.
3. The method of claim 1, wherein the frameless motor control driver is powered by a power conversion circuit comprising a bus voltage +5V, 12V power module and a 5V to 3.3V power module; the bus voltage is converted into a +5V and 12V power module through a DC/DC conversion chip, and then is converted into 3.3V from 5V by a voltage conversion chip;
wherein, 3.3V is used for supplying power for the DSP of the main control chip and peripheral circuits thereof;
+5V is used for communication of the CAN communication circuit and power supply of the current adopting and processing circuit;
12V is used for power supply of the driving circuit.
4. The method of claim 3, wherein the current sampling and processing circuit uses an ACS712 current sensor IC, the ACS712 current sensor IC detects the magnitude of the current by Hall effect, and outputs a voltage value based on 2.5V, and the voltage value of Vout is divided by a high precision resistor; after passing through the diode, the digital signal enters an AD acquisition pin of the DSP of the main control chip for AD conversion, and the diode plays a role in protection.
5. The method of claim 1, wherein a current protection circuit is disposed in the frameless motor control driver, and when a peak value of a current in the servo motor winding exceeds a rated current of the MOSFET of the power transistor, that is, a set value of the comparator LM339 is reached, the current protection circuit outputs a low level signal Fault signal to the Fault synthesis circuit, and triggers to generate a high level signal to the tri-state output bus receiver, and the action outputs a turn-off signal to turn off the power switch, thereby protecting the power switch transistor from being damaged.
6. The method for realizing the electric steering wheel motor drive controller according to claim 1, wherein the CAN communication circuit adopts a high-speed real-time bus to give a position and a speed given to the frameless motor control driver, control the rotation of the servo motor and realize the motion function of the joint; the method specifically comprises the following steps:
according to the motion required by one joint or a plurality of joints, the upper computer calculates the real-time running speed and position of each servo motor on the joint according to kinematics, the real-time requirement is high, and real-time data is transmitted to the corresponding servo motor through a high-speed real-time bus.
7. The method of claim 1 or 6, wherein the CAN communication circuit adopts model SN65HVD230 manufactured by TI as a receiving and transmitting chip of a local area network, and the power supply of the SN65HVD230 is 3.3V;
the CAN communication circuit adopts TVS as a CAN bus protector for protection and fault tolerance, so that a CAN transceiver in the CAN communication circuit is prevented from being influenced by EMI and ESD;
the diodes in the CAN communication circuit are in bidirectional configuration, so that clamping of a long cable system to normal data line signals due to common-mode voltage imbalance is prevented.
8. The method for implementing an electric steering wheel motor drive controller according to claim 1, wherein the main control chip DSP is initialized as follows:
(1) starting a main program of the main control chip DSP;
(2) initializing a system;
(3) clearing protection and error data;
(4) initializing hardware;
(5) initializing the functional module: the method comprises the steps of initializing a PWM module, initializing AD conversion, initializing an interrupt module, initializing an SCI module, initializing a CAN module and initializing an I/O port module;
(6) starting the linear grating ruler function;
(7) detecting and defining a phase current zero point;
(8) initializing the position of a motor rotor;
(9) initializing functional variables of the computer;
(10) enabling the interrupt;
(11) and judging whether to wait for interruption.
9. The method of claim 1 or 2, wherein the servo motor field oriented vector control system comprises a timer underflow interrupt submodule, which is specifically as follows:
s1, protecting the field;
s2, current sampling;
s3, sampling position difference;
s4, digital filtering;
s5, calculating the position of the rotor;
s6, refreshing SCI communication data;
s7, refreshing communication data of the CAN communication circuit:
s8, judging whether an outer closed loop exists:
if yes, go to step S9;
if not, jumping to step S11;
s9, position PID control;
s10, speed PI control;
s11, phase current processing;
s12, Clarke transformation;
s13, solving sim and cos values;
s14, Park conversion;
s15, adjusting q-axis current PI;
s16, adjusting the d-axis current PI;
s17, performing Park inverse transformation;
s18, obtaining a sector;
s19, obtaining T0、T1And T2；
And S20, updating data by the PWM comparator.
10. The method of claim 9, wherein the speed PI control in step S10 is as follows:
s10-1, adjusting the speed PI;
s10-2, setting n for reading speed*；
S10-3, calculating the deviation e (kt) ═ n*-n；
S10-4, calculating np(kT)＝Kp[e(kt)-e(kT-T)]；
S10-5, calculating ni(kT)＝Ki e(kt)；
S10-6, calculating delta (kT) ═ np(kT)+ni(kT)；
S10-7, processing to prevent integral saturation;
s10-8, giving q-axis current given i according to n (kT)q *；
The q-axis current PI adjustment in step S15 is specifically as follows:
s15-1, adjusting q-axis current PI;
s15-2, read Current given iq *；
S15-3, calculating the deviation e (kt) iq *-iq；
S15-4, calculating ip(kT)＝Kp[e(kt)-e(kT-T)]；
S15-5, calculating ii(kT)＝Ki e(kt)；
S15-6, calculating Δ (kT) ═ ip(kT)+ii(kT)；
S15-7, processing to prevent integral saturation;
s15-8, giving q-axis voltage given V according to i (kT)q *；
The d-axis current PI adjustment in step S16 is specifically as follows:
s16-1, adjusting d-axis current PI;
s16-2, read Current given id *；
S16-3, calculating the deviation e (kt) id *-id；
S16-4, calculating ip(kT)＝Kp[e(kt)-e(kT-T)]；
S16-5, calculating ii(kT)＝Ki e(kt)；
S16-6, calculating Δ (kT) ═ ip(kT)+ii(kT)；
S16-7, processing to prevent integral saturation;
s16-8, giving q-axis voltage given V according to i (kT)d *；
The sector is obtained in step S18 specifically as follows:
s18-1, calculating a sector;
s18-2, sector P ═ 0;
S18-3、P1＝sign(Vsβ)；
S18-4、P2＝2sign[sin60°Vsα-sin30°Vsβ]；
S18-5、P3＝4sign[-sin60°Vsα-sin30°Vsβ]；
S18-6、P＝P1+P2+P3；
s18-7, determining a SECTOR SECTOR by table lookup;
the PWM comparator update data in step S20 is specifically as follows:
s20-1, calculating SVPWM;
s20-2, reading a SECTOR number SECTOR;
s20-3, reading the inverse matrix table by table lookup, and calculating C0、C1And C2；
S20-4, calculating T1＝0.5C1；
S20-5, calculating T2＝0.5C2；
S20-6, calculating T0＝0.25C0；
S20-7, calculating a PWM comparator value;
and S20-8, updating the PWM comparator value.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202011318469.1A CN112519877A (en) | 2020-11-23 | 2020-11-23 | Method for realizing motor drive controller of electric steering wheel |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202011318469.1A CN112519877A (en) | 2020-11-23 | 2020-11-23 | Method for realizing motor drive controller of electric steering wheel |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CN112519877A true CN112519877A (en) | 2021-03-19 |
Family
ID=74992515
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN202011318469.1A Pending CN112519877A (en) | 2020-11-23 | 2020-11-23 | Method for realizing motor drive controller of electric steering wheel |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN112519877A (en) |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN113954955A (en) * | 2021-10-14 | 2022-01-21 | 济南科亚电子科技有限公司 | Implementation method of electric steering wheel motor drive controller |
Citations (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| KR20040108107A (en) * | 2003-06-16 | 2004-12-23 | 화남전자 주식회사 | Control circuit of DC Motor for Electronic Power Steering |
| CN106533303A (en) * | 2016-10-27 | 2017-03-22 | 国网重庆市电力公司电力科学研究院 | Permanent magnet brushless DC motor driver control method |
| CN110091908A (en) * | 2018-01-29 | 2019-08-06 | 株式会社捷太格特 | Steering apparatus |
-
2020
- 2020-11-23 CN CN202011318469.1A patent/CN112519877A/en active Pending
Patent Citations (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| KR20040108107A (en) * | 2003-06-16 | 2004-12-23 | 화남전자 주식회사 | Control circuit of DC Motor for Electronic Power Steering |
| CN106533303A (en) * | 2016-10-27 | 2017-03-22 | 国网重庆市电力公司电力科学研究院 | Permanent magnet brushless DC motor driver control method |
| CN110091908A (en) * | 2018-01-29 | 2019-08-06 | 株式会社捷太格特 | Steering apparatus |
Non-Patent Citations (1)
| Title |
|---|
| 单长伟: "基于永磁同步电机的电动助力转向系统控制研究", 《中国优秀博硕士学位论文全文数据库（硕士） 工程科技Ⅱ辑》 * |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN113954955A (en) * | 2021-10-14 | 2022-01-21 | 济南科亚电子科技有限公司 | Implementation method of electric steering wheel motor drive controller |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN101174783B (en) | Arc driving heavy caliber astronomical telescope | |
| CN108521242B (en) | Universal control system for permanent magnet synchronous motor and control method thereof | |
| CN101149471A (en) | Heavy caliber astronomical telescope azimuth axis arc movement control system | |
| CN105353682B (en) | Aircraft power system experiment platform device | |
| CN104579044B (en) | A kind of constant moment of force control method of ECM motors | |
| US10081251B2 (en) | Apparatus for controlling power converters | |
| CN110518680B (en) | Control circuit and control method compatible with three-phase and single-phase multiple wire system power supplies | |
| CN103078570A (en) | Torque pulsation restraining system and method of permanent magnet brushless direct current motor | |
| CN111830435A (en) | Open-circuit fault diagnosis method for power tube of six-phase permanent magnet fault-tolerant motor system | |
| CN106849806B (en) | Electric vehicle controller and control method | |
| CN201869153U (en) | Drive controller for permanent magnet synchronous motor of electromobile | |
| CN201928224U (en) | Direct-current brush motor servo driver | |
| US10756601B2 (en) | Control method and control system of motor rotation speed | |
| CN105048911A (en) | Large power high efficiency PMSM (Permanent Magnet Synchronous Motor) servo controller for manned electric airplane | |
| CN112519877A (en) | Method for realizing motor drive controller of electric steering wheel | |
| CN203708155U (en) | Permanent magnet synchronous motor (PMSM) controller and vehicle-mounted system for electric vehicles | |
| JP2021035232A (en) | Power conversion device | |
| CN113954955A (en) | Implementation method of electric steering wheel motor drive controller | |
| Men et al. | Development of an Advanced Motor Control System for Electric Vehicles | |
| CN106487307A (en) | The anti-back magnetic control system of permagnetic synchronous motor, method and unmanned vehicle | |
| CN103248284A (en) | Special driver for linear shaft motor | |
| CN115416497A (en) | Brushless direct current motor's braking rotational speed control system and low-speed electric motor car | |
| CN107994837B (en) | Speed regulating system and speed regulating method for switched reluctance motor | |
| CN214315107U (en) | Low-cost brushless DC motor controller for electric bicycle | |
| CN113541554B (en) | Self-adaptive belt speed charging control method for permanent magnet motor for flywheel |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| RJ01 | Rejection of invention patent application after publication |
Application publication date: 20210319 |
|
| RJ01 | Rejection of invention patent application after publication |