CN112217389A - Long-life high-power density current interrupted buck-boost power factor correction converter - Google Patents

Long-life high-power density current interrupted buck-boost power factor correction converter Download PDF

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CN112217389A
CN112217389A CN202010870905.XA CN202010870905A CN112217389A CN 112217389 A CN112217389 A CN 112217389A CN 202010870905 A CN202010870905 A CN 202010870905A CN 112217389 A CN112217389 A CN 112217389A
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circuit
output
input
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voltage
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高扬凯
吴桐
杨凡
姚凯
高阳
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Nanjing University of Science and Technology
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Nanjing University of Science and Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
    • H02M3/33523—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters with galvanic isolation between input and output of both the power stage and the feedback loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/2176—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only comprising a passive stage to generate a rectified sinusoidal voltage and a controlled switching element in series between such stage and the output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The invention discloses a long-life high-power density current discontinuous step-up and step-down power factor correction converter. The control circuit comprises a sampling circuit, a compensation circuit, a PWM (pulse width modulation) generating circuit and an isolation circuit; the output voltage data is collected by using a sampling circuit, then the data is compared and adjusted by a compensating circuit and a reference voltage, and then an output signal of the compensating circuitv EA A PWM generation circuit outputs a drive signal with a fixed duty ratio to perform constant duty ratio control. The invention reducesThe capacitance value of the energy storage capacitor of the DCM Flyback PFC (discontinuous mode Flyback power factor correction) converter is increased, the thin film capacitor with small volume and long service life is used for replacing the electrolytic capacitor with large volume and short service life, and the power density and the service life of the converter are improved.

Description

Long-life high-power density current interrupted buck-boost power factor correction converter
Technical Field
The invention relates to the technical field of alternating current-direct current converters of electric energy conversion devices, in particular to a long-life high-power density current intermittent buck-boost power factor correction converter.
Background
The input power of a conventional Flyback PFC converter is pulsed and the output voltage is dc, thus requiring an energy storage capacitor to balance the instantaneous input power with the output power. Generally, the capacity of the energy storage capacitor is large, so an electrolytic capacitor is usually selected. However, the lifetime of the electrolytic capacitor is usually only several thousand hours, and the lifetime of the electrolytic capacitor is reduced by half for every 10 ℃ rise in temperature, so the electrolytic capacitor is an important factor affecting the service life of the power supply. Meanwhile, the electrolytic capacitor has a larger volume, which influences the further improvement of the power density of the power supply. The invention provides a method for increasing voltage ripple to reduce the capacity of an energy storage capacitor, so that a long-life film capacitor can be used for replacing an electrolytic capacitor, and the service life and the power density of a power supply are effectively improved.
Disclosure of Invention
The invention aims to provide a long-life high-power density current discontinuous step-up and step-down voltage power factor correction converter which has a simple control circuit and good control effect, can effectively reduce the capacitance value of an energy storage capacitor in the whole 90V-264V AC input voltage range, and can replace an electrolytic capacitor with large capacitance by a thin film capacitor with small capacitance.
In order to solve the above technical problems, the present invention provides a long-life high-power density current discontinuous step-up/step-down power factor correction converter, wherein a main circuit of the converter comprises a main power circuit and a control circuit. The control circuit comprises a sampling circuit, a compensation circuit, a PWM generating circuit and an isolation circuit. Connecting the output end of the main power circuit with the input end of the sampling circuit, connecting the output end of the sampling circuit with the input end of the compensating circuit, connecting the output end of the compensating circuit with the input end of the PWM generating circuit, and connecting the output end of the PWM generating circuit with the isolation circuitThe input ends of the circuits are connected, and the output ends of the isolation circuits are connected with the main power circuit; the output voltage data is collected by using a sampling circuit, then the data is compared and adjusted by a compensating circuit and a reference voltage, and then an output signal v of the compensating circuitEAA PWM generating circuit outputs a drive signal with a fixed duty ratio to perform constant duty ratio control.
Further, the main power circuit comprises an input voltage source vinEMI filter, diode rectifying circuit RB, LC filter, transformer T and first switching tube QbFreewheel diode DbAn output capacitor CoAnd a constant power load Rcot(ii) a Said input voltage source vinThe output port of the EMI filter is connected with the input port of the rectifier bridge RB, the output negative port of the rectifier bridge RB is connected with the input negative port of the LC filter, the output positive port of the rectifier bridge RB is connected with the input positive port of the LC filter, the output positive port of the LC filter is connected with the positive end of the primary side inductor of the transformer T, the negative end of the primary side inductor of the transformer T is connected with the Q end of the first switching tube QbOne end of the LC filter is connected with the output negative port of the LC filter and the first switch tube QbOne end of the LC filter is connected, and the negative port of the LC filter is a reference potential zero point; the negative end of the secondary inductor of the transformer T and the freewheeling diode DbIs connected with the positive pole of the secondary side inductor of the main circuit and the output capacitor CoNegative pole and constant power load RcotConnecting; output capacitor CoAnode and freewheeling diode DbThe positive electrodes of the two electrodes are connected; constant power load RcotIs connected with the input end of the sampling circuit.
The control circuit comprises a sampling circuit, a compensation circuit, a PWM generating circuit and an isolation circuit; the sampling circuit comprises a first divider resistor RxAnd a second voltage dividing resistor Ry(ii) a The positive input end of the sampling circuit passes through a first voltage dividing resistor RxAnd the output voltage voIs connected with the positive port of the sampling circuit, and the reverse input end of the sampling circuit passes through a second divider resistor RyAnd the output voltage voIs connected with the negative port of the sampling circuit, the output port A of the sampling circuit is connected with the input port B of the compensation circuit, and the output end of the compensation circuitThe port C is connected with an input port D of the PWM generating circuit; an output port E of the PWM generating circuit is connected with an input port F of the isolating circuit; output port G and first switch tube Q of isolation circuitbAre connected.
Output voltage signal kv of voltage dividing resistoroInput into a compensation circuit which obtains an error signal v of a voltage closed loopEAWill error signal vEADirectly input into a PWM generating circuit to generate PWM wave, thereby controlling the first switch tube QbOn and off.
The compensation circuit comprises a first operational amplifier IC1, a resistor R1A first capacitor C1And a second capacitor C2(ii) a The positive input end of the first operational amplifier IC1 is connected with the output end A of the sampling circuit, the negative input end of the first operational amplifier IC1 is connected with the reference voltage, and the positive input end of the first operational amplifier IC1 is connected with the reference voltage through a resistor R1A first capacitor C1And a second capacitor C2Is connected with the output end; second capacitor C2And a resistance R1Connected with the input end of the PWM generating circuit.
The PWM generation circuit includes a second operational amplifier IC 2; the positive input end of the second operational amplifier IC2 is connected with the output end of the first operational amplifier IC1 in the compensation circuit, and the negative input end of the second operational amplifier IC1 is connected with the sawtooth wave; the output of the second operational amplifier IC2 is connected to the input of the isolation circuit.
The amplifiers used in the first operational amplifier IC1 and the second operational amplifier IC2 are TL072, TL074, LM324 or LM358 type operational amplifiers.
The compensation circuit and the PWM generating circuit use SG3525 or UC3525 type chips, and the isolation circuit uses a TLP250 type driving chip.
Compared with the prior art, the invention has the remarkable advantages that: 1) the control mode of the fixed duty ratio is simple, and the capacitance value of the energy storage capacitor of the converter can be reduced in the whole 90V-264V AC input voltage range; 2) the control mode is simple, so that the control stability is relatively high; 3) the thin film capacitor is used to replace electrolytic capacitor, so increasing power density of power supply and prolonging service life of converter
Drawings
Fig. 1 is a schematic diagram of a main power circuit structure and a control structure of a long-life high-power-density current discontinuous buck-boost power factor correction converter in an embodiment of the present invention, where 1 is a main power circuit, 2 is a sampling circuit, 3 is a compensation circuit, 4 is a PWM generation circuit, and 5 is an isolation circuit.
Fig. 2 is a block diagram of a two-stage Flyback PFC converter according to an embodiment of the present invention.
Fig. 3 is a main circuit diagram of a DCM Flyback PFC converter in an embodiment of the present invention.
Fig. 4 is a waveform diagram of primary and secondary side inductor currents and switching tube currents of the DCM Flyback PFC converter in one switching period according to the embodiment of the present invention.
Fig. 5 is a graph of output voltage ripple versus energy storage capacitor according to an embodiment of the present invention.
Fig. 6 is a graph of primary inductance versus maximum input voltage for an embodiment of the present invention.
Fig. 7 is a graph of voltage stress of the switching tube and the secondary rectifier diode versus transformer turns ratio in an embodiment of the invention.
Fig. 8 is a graph of output voltage waveforms for different energy storage capacitors in an embodiment of the invention.
Fig. 9 is a waveform diagram of input current at different input voltages in an embodiment of the present invention.
Main symbol names in the above figures: v. ofinAnd a power supply voltage. i.e. iinAnd inputting the current. RB, a rectifier bridge. v. ofgAnd the rectified output voltage. i.e. ipPrimary side inductor current. i.e. isAnd secondary side inductance current. L ispA primary side inductance. L issAnd a secondary side inductor. QbAnd a switch tube. DbAnd a diode. CoAnd an output filter capacitor. RcotAnd a constant power load. v. ofoAnd outputting the voltage. VrefAnd outputting the reference voltage of voltage feedback control. v. ofEAAnd outputting the error voltage signal controlled by the output voltage feedback. t, time.ω, input voltage angular frequency. VmInput voltage peak. v. ofgsAnd a switching tube QbThe driving voltage of (1). DyAnd Flyback converter duty cycle. DRAnd the duty ratio corresponding to the Flyback secondary side current dropping to zero. T issAnd the switching period of the converter. f. ofsAnd the converter switching frequency.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the specification, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be obtained by a person skilled in the art without any inventive step based on the embodiments of the present invention, belong to the scope of the present invention.
Example 1
The working principle of the DCM Flyback PFC converter is as follows:
fig. 2 is a block diagram of a two-stage Flyback PFC converter. In the two-stage PFC converter, the rear-stage DC/DC converter is a constant power load of the front-stage PFC converter, and thus the two-stage DCM Flyback PFC converter can be simplified as shown in fig. 3.
Fig. 3 is a DCM Flyback PFC converter main circuit.
Setting: 1. all devices are ideal elements; 2. the output voltage ripple is large compared to its direct current amount;
3. the switching frequency is much higher than the input voltage frequency.
Figure 4 shows the switching tube current and inductor current waveforms during one switching cycle in DCM mode of operation,
from FIG. 4, when the switch tube QbWhen conducting, the voltage across the primary side inductor is vgAnd primary side inductor current iLpStarting from zero with vg/LpThe slope of (c) rises linearly. Therefore, the inductor current peak value i of the DCM Flyback PFC converter in one switching periodLp_pkCan be expressed as
Figure BDA0002651077550000041
In the formula fsTo the switching frequency, fs＝1/Ts。
Average value i of primary side inductance current in one switching periodLp_avIs composed of
Figure BDA0002651077550000042
Then the input current iinIs composed of
Figure BDA0002651077550000043
From equation (3), it can be seen that the duty cycle D when the converter is operatingyWhen the input current signal is kept unchanged in a power frequency period, the input current signal is in a sine form, and PFC can be automatically realized.
Without loss of generality, the AC input voltage of the power grid side is set as
vin＝Vm sin ωt (4)
From the expressions (3) and (4), the average value P of the input power in the half power frequency period can be obtainedin
Figure BDA0002651077550000051
Assuming that the converter operates with an efficiency of 1 (the same applies hereinafter), the average value of the input power is equal to the output power, i.e. Pin＝Po. The duty ratio D under conventional control can be obtained from equation (5)yIs composed of
Figure BDA0002651077550000057
By substituting formula (6) for formula (3), the input current expression is obtained
Figure BDA0002651077550000052
The expression of the secondary side inductance current in one switching period can be obtained according to the graph of FIG. 4 as
Figure BDA0002651077550000053
The expression of the energy storage capacitor current in one switching period obtained from the formula (8) is
Figure BDA0002651077550000054
The output voltage ripple obtained from equation (9) is expressed as:
Figure BDA0002651077550000055
in the above formula, ton＝DyTsThe conduction time of the switching tube is set; t is toff＝(Dy+DR)TsThe time when the secondary side inductance current drops to zero is shown. Because the power frequency period is far greater than the switching period (T)line＝0.02s>>T s10 mus), v can be considered to beg、voAnd ioA constant value in a switching period, so that the following equation can be obtained:
Figure BDA0002651077550000056
from the above formula, when the output voltage ripple Δ V is increasedoIn time, the energy storage capacitance can be reduced.
Example 2
Method of increasing voltage ripple
2.1 instantaneous expression of output Voltage
The energy stored by the output capacitor can be expressed as
Figure BDA0002651077550000061
In the formula Vo_0Is the initial value of the output voltage in one switching period.
Substituting the above formulas (4) and (7) and simplifying the reaction
Figure BDA0002651077550000062
Thus the output voltage voCan be expressed as
Figure BDA0002651077550000063
The average value of the output voltage in a power frequency period can be expressed as
Figure BDA0002651077550000064
Substituting equation (14) into:
Figure BDA0002651077550000065
order:
Figure BDA0002651077550000066
can be simplified as follows:
Figure BDA0002651077550000067
Figure BDA0002651077550000068
order to
Figure BDA0002651077550000069
Then it can be simplified to:
Figure BDA0002651077550000071
as can be seen from the equation (14), the output voltage v is adjustedoIs of constant significance and needs to satisfy
Figure BDA0002651077550000072
Is easy to obtain
Figure BDA0002651077550000073
k2If the result is less than 1, the solving condition of the second type of elliptic integral is satisfied, and the elliptic integral is used for calculation, so that the following can be obtained:
Figure BDA0002651077550000074
the ellipse is integrated into a series form:
Figure BDA0002651077550000075
wherein V inside the elliptic integralo_0Can be approximately regarded as Vo(ii) a The high-order terms can be ignored in the series form, and n can be taken from 0 to 3. Then reversely solve Vo_0The following can be obtained:
Figure BDA0002651077550000076
can obtain an output voltage v by substituting the formula (14)oIs expressed as
Figure BDA0002651077550000081
From the above formula, the expression of the output voltage ripple is
Figure BDA0002651077550000082
From the above formula,. DELTA.voThe image for the storage capacitor C is shown in fig. 5.
As can be seen from fig. 5, the variation of the output voltage ripple is inversely proportional to the variation of the capacitance of the energy storage capacitor, as in the conclusion from equation (11). Therefore, the capacitance value of the energy storage capacitor can be reduced by increasing the voltage ripple.
2.2 control Circuit
D can be found by observing the formula (6)yIs about VmAs a function of, in control DyAt the value of (v), we only need to guarantee the output voltage voThe average value of the voltage is a fixed value, and the duty ratio calculated by the theory can be obtained through the automatic adjustment of the voltage in a closed loop mode. From the duty ratio shown in equation (6), a control circuit diagram as shown in fig. 1 can be designed. The collected output voltage signal kvoInputting the error signal into a compensation circuit, and obtaining an error signal v of a voltage closed loop through the compensation circuitEAWill error signal vEAThe driving signal is directly input into the PWM generating circuit to obtain a driving signal, and the driving signal controls the on and off of the switching tube through the isolating circuit.
With reference to fig. 1, the main power circuit comprises an input voltage source vinEMI filter, diode rectifying circuit RB, LC filter, transformer T and first switching tube QbFreewheel diode DbAn output capacitor CoAnd a constant power load Rcot(ii) a Said input voltage source vinThe output port of the EMI filter is connected with the input port of the rectifier bridge RB, the output negative port of the rectifier bridge RB is connected with the input negative port of the LC filter, the output positive port of the rectifier bridge RB is connected with the input positive port of the LC filter, the output positive port of the LC filter is connected with the positive end of the primary side inductor of the transformer T, the negative end of the primary side inductor of the transformer T is connected with the Q end of the first switching tube QbOne end of the LC filter is connected with the output negative port of the LC filter and the first switch tubeQbOne end of the LC filter is connected, and the negative port of the LC filter is a reference potential zero point; the negative end of the secondary inductor of the transformer T and the freewheeling diode DbIs connected with the positive pole of the secondary side inductor of the main circuit and the output capacitor CoNegative pole and constant power load RcotConnecting; output capacitor CoAnode and freewheeling diode DbThe positive electrodes of the two electrodes are connected; constant power load RcotIs connected with the input end of the sampling circuit.
Further, the control circuit comprises a sampling circuit, a compensation circuit, a PWM generating circuit and an isolation circuit; the sampling circuit comprises a first divider resistor RxAnd a second voltage dividing resistor Ry(ii) a The positive input end of the sampling circuit passes through a first voltage dividing resistor RxAnd the output voltage voIs connected with the positive port of the sampling circuit, and the reverse input end of the sampling circuit passes through a second divider resistor RyAnd the output voltage voThe output port A of the sampling circuit is connected with the input port B of the compensation circuit, and the output port C of the compensation circuit is connected with the input port D of the PWM generating circuit; an output port E of the PWM generating circuit is connected with an input port F of the isolating circuit; output port G and first switch tube Q of isolation circuitbAre connected.
Further, the compensation circuit comprises a first operational amplifier IC1, a resistor R1A first capacitor C1And a second capacitor C2(ii) a The positive input end of the first operational amplifier IC1 is connected with the output end A of the sampling circuit, the negative input end of the first operational amplifier IC1 is connected with the reference voltage, and the positive input end of the first operational amplifier IC1 is connected with the reference voltage through a resistor R1A first capacitor C1And a second capacitor C2Is connected with the output end; second capacitor C2And a resistance R1Connected with the input end of the PWM generating circuit.
Further, the PWM generation circuit includes a second operational amplifier IC 2; the positive input end of the second operational amplifier IC2 is connected with the output end of the first operational amplifier IC1 in the compensation circuit, and the negative input end of the second operational amplifier IC1 is connected with the sawtooth wave; the output of the second operational amplifier IC2 is connected to the input of the isolation circuit.
Further, an output voltage signal kv of the voltage dividing resistoroInput into a compensation circuit which obtains an error signal v of a voltage closed loopEAWill error signal vEADirectly input into a PWM generating circuit to generate PWM wave, thereby controlling the first switch tube QbOn and off.
Furthermore, the amplifiers used in the first operational amplifier IC1 and the second operational amplifier IC2 are TL072, TL074, LM324 or LM358 type operational amplifiers.
Furthermore, the compensation circuit and the PWM generation circuit use SG3525 or UC3525 type chips, and the isolation circuit uses TLP250 type driver chips.
Example 3
Performance analysis:
3.1 selection principle of capacitance value of energy storage capacitor
The capacitance value of the energy storage capacitor is reduced, so that the service life of the power supply can be prolonged, the volume of the capacitor is reduced, and the power density is improved.
But is limited by the certain establishment of the output voltage, which is obtained from equation (21), the selection principle of the capacitor is as follows:
Figure BDA0002651077550000101
in the above formula, take Po＝120W，ω＝100πrad/s，Vo_0≈Vo80V, the minimum value of capacitance C can be calculatedmin59.68 μ F. Considering the available capacitance values, the final capacitance value of the storage capacitor is 68 μ F.
3.2 selection of the turns ratio n
Switch tube Q of Flyback converterbMaximum reverse voltage V borne across it during turn-offQb_maxCan be expressed as
Figure BDA0002651077550000105
Secondary side rectifier diode DbSubjected to a maximum reverse voltage of
Figure BDA0002651077550000102
Wherein, Vm_maxIs the peak value of the highest input voltage,
Figure BDA0002651077550000103
Vo_maxis the peak value of the highest output voltage with the magnitude of
Figure BDA0002651077550000104
Substituting the output power and capacitance of energy-storing capacitor into the above formula to obtain Vo_max＝116.874(V)。
Based on the design criteria of the converter, the switching tube Q can be made by equations (28) and (29)bAnd a secondary side diode DbThe voltage stress versus transformer turns ratio of (a) is shown in fig. 7. It can be seen that the larger the turn ratio n of the transformer, the larger the switching tube QbThe higher the voltage stress, and the secondary diode DbThe lower the voltage stress; in contrast, when the turn ratio n of the transformer decreases, the switching tube QbVoltage stress reduction of while the secondary side diode DbThe voltage stress of (2) increases. Comprehensive consideration of primary side switching tube QbAnd a secondary side diode DbWhile leaving a certain margin, and taking into account its cost, in order to select a 600V MOS transistor, the switching transistor Q should be madebThe voltage stress is less than 600V. Thus, it is possible to obtain:
Figure BDA0002651077550000116
the above formula can be used to obtain n less than or equal to 2.013, so the turn ratio n of the transformer is 2.
3.3 design of inductance value
As can be seen from the figure 4 of the drawings,secondary side current iLsTime t corresponding to the drop from its peak value to zeroRIs composed of
Figure BDA0002651077550000111
The duty ratio D corresponding to the secondary side current dropping to zeroRIs composed of
Figure BDA0002651077550000112
The duty ratio of the DCMFlybackPFC converter in one switching period has the following relationship
Dy+DR≤1 (34)
Therefore, substituting the above formula into the formulas (6) and (33) can obtain
Figure BDA0002651077550000113
Simple and available
Figure BDA0002651077550000114
The right derivation of the above inequality can be obtained when
Figure BDA0002651077550000115
The right side of the inequality takes the minimum value. Further obtaining primary side inductance LpIs selected according to the principle that
Figure BDA0002651077550000121
Wherein, taking Po＝120W，ω＝100π rad/s，fs＝100kHz。
The capacitance value of the selected energy storage capacitor C is 68 muF, and the corresponding V under the capacitance value of the capacitor C is calculated by the formula (11)o_084.036V. Then is formed byAs can be seen from equation (37), under different input voltages, the inductance values of the primary inductors can be obtained. Taking the effective value V of the input voltagein_rmsRespectively 90V, 110V, 130V, 176V, 220V, 246V, then L can be drawnpAnd VmAs shown in fig. 6.
From FIG. 6, LpFollowing VmIs increased, then Vin_rmsWhen the voltage is taken to be 90V,
Figure BDA0002651077550000122
Lpthe minimum value is obtained. Substituting the value into the formula (37) to obtain the maximum value L of the primary inductancepmax66.7 muH, the primary inductance L of the transformer is selectedpThe sensitivity value was 64. mu.H.
Secondary inductance L of transformers＝Lp/n2The turn ratio n is 2, so the secondary inductance L of the transformer is selectedsThe sensitivity value was 16. mu.H.
3.4 calculation of Primary turns
For the DCM Flyback transformer, the magnetic core is suitable for selecting the type with wider window area. When the core size is the same, the window area of the RM type and PQ type cores is small. In the manufacturing process of the prototype, a PQ32/30 type magnetic core made by TDK company is selected, the magnetic core material is PC44, the saturation magnetic flux density Bs at 100 ℃ is 0.39T, and delta B is 0.2T. Effective area A of PQ32/30 coree＝161mm2Window area Aw＝149.6mm2。
As can be seen from FIG. 6, the primary side inductance LpThe peak value of the primary side inductance current is obtained by replacing the formula (6) with the formula (1) by taking 64 mu H
Figure BDA0002651077550000123
Substituting the parameters into the above formula to obtain the maximum peak value i of the primary side inductance currentLp_pk_max8.66A, the number of turns N on the primary side can be calculatedp1Is composed of
Figure BDA0002651077550000124
Actually taking the number of turns of the primary side Np118 since the turn ratio N is 2, the number of secondary turns N is then equal tos1＝9。
3.5 input Current waveform
The waveform of the input current within a half power frequency period under different input voltages can be made by equation (7), as shown in fig. 9. Under the same capacitance value of the energy storage capacitor, when the input voltage increases, the amplitude of the input current decreases along with the increase of the input voltage.
3.6 output Voltage waveform
According to the formula (25), v can be madeoThe theoretical approximation curve of (2) is shown in fig. 8. Under the same input voltage, the output voltage ripple increases along with the decrease of the capacitance value of the energy storage capacitor, and conforms to the theoretical change rule of the formula (11). From equation (26), it can be concluded that the variation of the input voltage has no effect on the output voltage ripple. The variation in capacitance is a major factor affecting the output voltage ripple.
In summary, the long-life high-power-density current discontinuous step-up and step-down voltage power factor correction converter provided by the invention adopts a method of increasing voltage ripples, so that the capacitance value of the energy storage capacitor is reduced, and the thin film capacitor is used for replacing an electrolytic capacitor, so that the power density of the converter is improved, and the service life of the converter is prolonged.
Although the present invention has been described in detail with reference to the foregoing embodiments, it will be apparent to those skilled in the art that various changes in the embodiments and modifications of the invention can be made, and equivalents and modifications of some features of the invention can be made without departing from the spirit and scope of the invention.

Claims (8)

1. The long-life high-power-density current interrupted buck-boost power factor correction converter is characterized by comprising a main power circuit and a control circuit, wherein the control circuit comprises a sampling circuit, a compensation circuit, a PWM (pulse width modulation) generating circuit and an isolation circuit; output end and sampling circuit of main power circuitThe output end of the sampling circuit is connected with the input end of the compensating circuit, the output end of the compensating circuit is connected with the input end of the PWM generating circuit, the output end of the PWM generating circuit is connected with the input end of the isolating circuit, and the output end of the isolating circuit is connected with the main power circuit; the sampling circuit is used for collecting output voltage data, then the compensation circuit is used for comparing and adjusting with reference voltage, and then the output signal of the compensation circuitv EA A PWM generating circuit outputs a drive signal with a fixed duty ratio to perform constant duty ratio control.
2. The long life high power density current chopping buck-boost power factor correction converter as claimed in claim 1, wherein the main power circuit includes an input voltage sourcev in An EMI filter, a diode rectifier circuit RB,LCFilter, transformer T, first switch tubeQ b Freewheel diodeD b Output capacitorC o And a constant power loadR cot (ii) a The input voltage sourcev in Is connected with the input port of the EMI filter, the output port of the EMI filter is connected with the input port of the rectifier bridge RB, and the output negative port of the rectifier bridge RB is connected with the input port of the rectifier bridge RBLCThe input negative port of the filter is connected with the output positive port of the rectifier bridge RBLCThe input positive port of the filter is connected with the input positive port,LCthe positive output port of the filter is connected with the positive terminal of the primary inductor of the transformer T, and the negative terminal of the primary inductor of the transformer T is connected with the first switching tubeQ b One end of the connecting rod is connected with the other end of the connecting rod,LCoutput negative port of filter and first switch tubeQ b The other end is connected with the other end,LCthe negative port of the filter is a reference potential zero point; secondary side inductance negative pole end and fly-wheel diode of transformer TDIs connected with the positive pole of the secondary side inductor of the main circuit and the output capacitorC o Negative electrode and constant power loadR cot Connecting; output capacitorC o Anode and freewheeling diode ofD b The positive electrodes of the two electrodes are connected; constant power loadR cot And samplingThe input terminals of the circuit are connected.
3. The long life high power density current chopping buck-boost power factor correction converter of claim 1, wherein said control circuit includes a sampling circuit, a compensation circuit, a PWM generation circuit, and an isolation circuit; the sampling circuit comprises a first voltage dividing resistorR x And a second voltage dividing resistorR y (ii) a The positive input end of the sampling circuit passes through a first voltage dividing resistorR x And the output voltagev o The reverse input end of the sampling circuit passes through a second divider resistorR y And the output voltagev o Is connected with the negative port of the sampling circuitAAnd input port of compensation circuitBOutput port of the compensation circuitCAnd input port of PWM generating circuitDConnecting; output port of PWM generating circuitEInput port of isolation circuitFConnecting; output port of isolation circuitGAnd a first switch tubeQ b Are connected.
4. The long life high power density current chopping buck-boost pfc converter of claim 1, wherein the output voltage signal of the voltage divider resistor is usedkv o Input into a compensation circuit which obtains an error signal of a voltage closed loopv EA Will be an error signalv EA Directly input into the PWM generating circuit to generate PWM wave, thereby controlling the first switch tubeQ b On and off.
5. The long life high power density current chopping buck-boost power factor correction converter of claim 3, wherein said compensation circuit includes a first operational amplifier IC1, a resistorR 1A first capacitorC 1And a second capacitorC 2(ii) a The positive input end of the first operational amplifier IC1 and the output of the sampling circuitOut endAThe inverting input terminal of the first operational amplifier IC1 is connected to a reference voltage, and the forward input terminal of the first operational amplifier IC1 is connected through a resistorR 1A first capacitorC 1And a second capacitorC 2Is connected with the output end; second capacitorC 2And a resistorR 1Connected with the input end of the PWM generating circuit.
6. The long life high power density current chopping buck-boost power factor correction converter of claim 3, wherein said PWM generation circuit includes a second operational amplifier IC 2; the positive input end of the second operational amplifier IC2 is connected with the output end of the first operational amplifier IC1 in the compensation circuit, and the negative input end of the second operational amplifier IC1 is connected with the sawtooth wave; the output of the second operational amplifier IC2 is connected to the input of the isolation circuit.
7. The long-life high-power-density current chopping buck-boost power factor correction converter as claimed in claim 3, wherein the amplifier used in the first operational amplifier IC1 and the second operational amplifier IC2 is TL072, TL074, LM324 or LM358 type operational amplifiers.
8. The long-life high-power-density current chopping buck-boost power factor correction converter as claimed in claim 3, wherein the compensation circuit and the PWM generation circuit use SG3525 or UC3525 type chips, and the isolation circuit uses TLP250 type driver chips.
CN202010870905.XA 2020-08-26 2020-08-26 Long-life high-power density current interrupted buck-boost power factor correction converter Pending CN112217389A (en)

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CN113162392A (en) * 2021-04-25 2021-07-23 西安领充创享新能源科技有限公司 Power factor correction method, device, equipment and storage medium
CN113315388A (en) * 2021-06-25 2021-08-27 江苏容正医药科技有限公司 High-power-density long-life high-frequency pulse alternating-current power supply
CN115242078A (en) * 2022-09-22 2022-10-25 广东希荻微电子股份有限公司 Power factor correction circuit and method and electronic equipment
US11923763B1 (en) 2022-08-23 2024-03-05 Halo Microelectronics International Ripple cancellation apparatus and control method

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Cited By (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN113162392A (en) * 2021-04-25 2021-07-23 西安领充创享新能源科技有限公司 Power factor correction method, device, equipment and storage medium
CN113162392B (en) * 2021-04-25 2023-12-22 西安领充创享新能源科技有限公司 Power factor correction method, device, equipment and storage medium
CN113315388A (en) * 2021-06-25 2021-08-27 江苏容正医药科技有限公司 High-power-density long-life high-frequency pulse alternating-current power supply
US11923763B1 (en) 2022-08-23 2024-03-05 Halo Microelectronics International Ripple cancellation apparatus and control method
CN115242078A (en) * 2022-09-22 2022-10-25 广东希荻微电子股份有限公司 Power factor correction circuit and method and electronic equipment
CN115242078B (en) * 2022-09-22 2023-01-06 广东希荻微电子股份有限公司 Power factor correction circuit and method and electronic equipment

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