CN112054729A - Permanent magnet motor control method suitable for low-speed direct-drive elevator - Google Patents

Permanent magnet motor control method suitable for low-speed direct-drive elevator Download PDF

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CN112054729A
CN112054729A CN202010957871.8A CN202010957871A CN112054729A CN 112054729 A CN112054729 A CN 112054729A CN 202010957871 A CN202010957871 A CN 202010957871A CN 112054729 A CN112054729 A CN 112054729A
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torque
current
load
control
switching
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CN112054729B (en
Inventor
史光辉
王正国
邱玉林
刘涛涛
张念钰
王浩
狄辉辉
武振华
王浩然
岳耀辉
王璐
宋海凤
任高飞
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Luoyang CITIC HIC Automation Engineering Co Ltd
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Luoyang CITIC HIC Automation Engineering Co Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/02—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for optimising the efficiency at low load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/04—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for very low speeds
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/20—Estimation of torque
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/10—Arrangements for controlling torque ripple, e.g. providing reduced torque ripple
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/28—Arrangements for controlling current

Abstract

The invention belongs to the technical field of control of low-speed direct-drive hoists, and particularly discloses a permanent magnet motor control method suitable for a low-speed direct-drive hoistdThe mode of control of =0 is combined with the mode of control of maximum torque current ratio, i is used in light loaddThe characteristic that the control current is completely decoupled is achieved by =0, the cogging torque pulsation is reduced by accurately compensating the q-axis current, the running stability of the elevator is improved, and the cogging torque is compared with the load torque after the load is increasedThe invention can control the permanent magnet motor to realize strong torque output, good low-speed performance and quick dynamic response, thereby leading the low-speed direct drive elevator system to achieve the best working state.

Description

Permanent magnet motor control method suitable for low-speed direct-drive elevator
Technical Field
The invention belongs to the technical field of control of low-speed direct-drive hoists, and particularly discloses a permanent magnet motor control method suitable for a low-speed direct-drive hoist.
Background
In order to improve the system performance and efficiency, a low-speed direct drive mode without a speed reducer is mostly adopted for a high-power elevator, the motor is a low-speed high-torque motor and is directly connected with an elevator host, so that the requirements on the low-speed performance and the torque output capacity of a variable frequency drive system are extremely high, in order to improve the carrying capacity of the system, an electrically excited synchronous motor is mostly adopted, the motor has complicated excitation systems, carbon brush slip rings and other equipment, the motor is oversized due to too many pole pairs, the rotating speed can be generally reduced only by reducing the rated frequency, and the cost and the control difficulty of a frequency converter can be greatly increased.
In recent years, the permanent magnet synchronous motor tends to be mature, the application range is more and more extensive, the permanent magnet synchronous motor not only has the characteristics of good low-speed performance, strong overload capacity and the like of an electrically excited motor, but also has the advantages of high efficiency, simple structure, convenience in maintenance and the like, and the characteristics can be perfectly wedged with a low-speed direct-driven elevator, so that the whole elevator system has the advantages of high performance, high efficiency, easiness in maintenance and the like.
Because the direct-drive system is not provided with a speed reducer, the low-speed stability of the hoister seriously influenced by the cogging torque pulsation of the permanent magnet motor during light load and the load disturbance during heavy load is equivalent to sudden load at the moment of opening the brake of the hoister, and the system needs to operate at high speed and high speed, and the working condition characteristics have high requirements on the low-speed performance, the overload capacity and the dynamic performance of the variable-frequency control system; the traditional permanent magnet frequency conversion adopts idThe control mode is controlled to be 0, the power factor is reduced rapidly along with the increase of the load torque, the complex working condition and the high performance requirement of the low-speed direct-drive hoister cannot be completely met in the aspects of torque output, low-speed performance, dynamic response and the like, and the optimal performance of the permanent magnet motor cannot be exerted.
Disclosure of Invention
In order to solve the problems in the background art, the invention discloses a permanent magnet motor control method suitable for a low-speed direct-drive elevator, which adopts idThe control mode is combined with the maximum torque current ratio control mode, so that the system torque response speed is increased, and the start and stop impact of the elevator is reduced.
In order to achieve the purpose, the invention adopts the following technical scheme:
a control method of a permanent magnet motor suitable for a low-speed direct-drive elevator is characterized in that a control system adopting the control method comprises a load observer, a cogging torque compensation module and a control mode switching moduleThe control method adopts idCombining two control modes of 0 and maximum torque current ratio, and adopting i in light loaddWhen the load torque is increased, the cogging torque ratio is reduced, the influence on the system is also reduced, and the maximum torque current ratio control mode is adopted, wherein the two control modes are based on the electromagnetic torque TeLinear switching is performed.
Further, the electromagnetic torque T of the permanent magnet motoreCalculated from the following formula:
Figure BDA0002677853440000021
wherein: t iseIs an electromagnetic torque; l isd,LqD and q axis inductances of the stator windings; i.e. isIs the stator winding current; ΨrFlux linkages generated for rotor permanent magnets; pnIs the number of pole pairs; alpha is a included angle between the stator current and the d axis;
by using idWhen control is 0, α is 90 °, iq＝is,id0, electromagnetic torque Te＝PnΨriqAt the moment, the torque output is only related to the q-axis current, and the cogging torque ripple can be inhibited as long as the q-axis current is accurately compensated;
when the maximum torque current ratio is adopted for control, the included angle between the stator current and the d axis is alphaT：
Figure BDA0002677853440000031
At the moment, the reluctance torque is driving torque, the unit current torque output capacity of the permanent magnet motor is strongest, but the torque current is not completely decoupled from the motor current, so that the cogging torque cannot be accurately compensated;
the system ensures the light load stability and the heavy load torque output capacity through controlling the mode switching, the linear switching of the control mode can be realized by controlling the included angle alpha between the stator current and the d axis, and the switching interval can be set according to the actual condition;
the switching coefficient is:
Figure BDA0002677853440000032
wherein K is a switching coefficient, the positive amplitude limiting value is 1, and the negative amplitude limiting value is 0; t isNRated torque of the motor; b is the ratio of the upper limit torque of the switching interval to the rated torque; a is the ratio of the lower limit torque to the rated torque of the switching interval, TeIs an electromagnetic torque;
the included angle between the stator current and the d axis is alpha when the switching coefficient K is controlled by adopting the maximum torque current ratioTCalculating the included angle alpha between the stator current and the d-axis by substituting the following formula:
Figure BDA0002677853440000041
as can be seen from the above formula, when K is 0
Figure BDA0002677853440000042
By using idControl is equal to 0; when K is 1, alpha is alphaTControlling the maximum torque current ratio; the switching state between the two control modes is realized when K is more than 0 and less than 1, the control effect is between the two control modes, and the load is moderate at the moment, so that the influence of the cogging torque is avoided, and the load capacity of the belt is not influenced.
Further, the load observation value output by the load observer is used
Figure BDA0002677853440000043
Calculating the observed value of the load torque current according to the following formula
Figure BDA0002677853440000044
Figure BDA0002677853440000045
Wherein:
Figure BDA0002677853440000046
for feeding forward compensation values for load torque current, Ld、LqD and q axis inductances of the stator winding; ΨrFlux linkages generated for rotor permanent magnets; pnIs the number of pole pairs;
Figure BDA0002677853440000051
and the output of the speed loop isLAdding to obtain the given value i of the stator currents *The dynamic response capability of the system is improved under the combined action of speed loop adjustment and load observation compensation, and the direction of compensation current is determined according to the actual torque direction.
Further, idThe linear switching between the 0 control mode and the maximum torque current ratio control mode includes the steps of:
step one, establishing a tooth space torque compensation meter, driving a motor in an open-loop mode by a down converter in an idle state, and carrying out current i on the motord、iqMagnetic linkage psirThe rotor angle theta is recorded, and the cogging torque T is calculated by using the following formulaco：
Tco＝Pn(iqΨr-(Ld-Lq)idiq)
Wherein; t iscoIs the cogging torque; l isd,LqD and q axis inductances of the stator windings; i.e. id、iqD and q axis current feedback values; ΨrFlux linkages generated for rotor permanent magnets; pnIs the number of pole pairs;
according to the formula
Figure BDA0002677853440000052
Calculate idCompensation value i of cogging torque current under control of 0coEqually dividing the motor into N parts by rotating the motor for 360 degrees in one circle, wherein N is a positive integer and is more than or equal to 100 and less than or equal to 500, and recording i in each equally divided intervalcoAnd its corresponding rotor angleDegree theta, establishing a cogging torque compensation table, and outputting a torque current compensation value i according to the current rotor angle theta of the motor during system operationcoAnd is combined with iq0 *Adding as q-axis current given iq *；
Determining a control mode switching point, setting an upper limit value and a lower limit value of torque switched by the control mode according to a tooth socket torque peak value recorded by a tooth socket torque compensation table in the step one, setting a switching lower limit point a according to 3 times of a ratio of the tooth socket torque peak value to a rated torque, setting a switching upper limit point b to be a +0.2, when the actual load torque is smaller than the switching lower limit torque, accurately compensating by adopting id to be 0, when the actual load torque is larger than the switching upper limit torque, controlling the load torque to be large and the tooth socket torque to be small by adopting a maximum torque current ratio, and improving the load carrying capacity;
step three, designing the load observer and calculating a load current compensation value, and selecting a pole lambda of the load observer1、λ2And calculating the gain coefficient K of the load observer1＝-Jλ1λ2、
Figure BDA0002677853440000061
Wherein: k1、K2Is a feedback gain coefficient, and J is a moment of inertia; d is a damping coefficient;
from load torque observations
Figure BDA0002677853440000062
Evaluating load current observations
Figure BDA0002677853440000063
Figure BDA0002677853440000064
Step four, designing a current loop, wherein the current loop is an inner loop and is not influenced by loads, a current regulator can be designed firstly, and a typical I system is adopted according to PWM (pulse-width modulation) parameters and motor parametersDesigning a current loop PI regulator, and obtaining d-axis and q-axis current feedback values i by coordinate transformation of three-phase current sampling valuesd、iqStator current set value
Figure BDA0002677853440000065
Calculated by maximum torque current ratio and control mode switching
Figure BDA0002677853440000066
Compensation value i for torque ripple of tooth socketcoAdd to obtain
Figure BDA0002677853440000067
Given value
Figure BDA0002677853440000068
And a feedback value id、iqRespectively outputting voltage instructions u through a PI regulatord、uqThereby controlling the inverter output voltage;
designing a speed ring, designing a PI (proportional integral) regulator according to a typical II system according to parameters such as the rotational inertia, the control period, the speed regulation range and the like of the system, enabling the speed regulator to work normally in the whole speed regulation range, switching PI parameters according to the speed range, and regulating an output i by the speed ringsLAnd observed value of load current
Figure BDA0002677853440000071
Summed current setpoint
Figure BDA0002677853440000072
Compared with the prior art, the invention has the beneficial effects that:
the invention discloses a permanent magnet motor control method suitable for a low-speed direct-drive elevator, which adopts idCombining the control mode of 0 and the control mode of maximum torque current ratio, i is used in light loaddThe characteristic that the current is completely decoupled is controlled as 0, cogging torque pulsation is reduced by accurately compensating the q-axis current, the running stability of the elevator is improved, and cogging torque and load are increased after the load is increasedThe control method of the invention aims at the potential energy type load characteristic of the elevator system, calculates the load current observation value by using the load torque observation value, and compensates to the speed ring output, thereby greatly accelerating the system torque response speed, reducing the starting and stopping impact of the elevator, and the invention can control the permanent magnet motor to realize strong torque output, good low-speed performance and quick dynamic response, so that the low-speed direct drive elevator system can reach the optimal working state.
Drawings
FIG. 1 shows the present invention idThe control mode and the maximum torque current ratio control mode are switched to form a flow diagram;
FIG. 2 is a flow chart of current control and calculation according to the present invention;
fig. 3 is a load observer control block diagram in the present invention.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
The invention discloses a permanent magnet motor control method suitable for a low-speed direct-drive elevator, which combines idThe control method has the advantages of 0 control and maximum torque current ratio control, modules such as a load observer, a cogging torque compensation module, a control mode switching module and the like are added in a traditional permanent magnet motor control system, idThe control mode and maximum torque current ratio control mode switching flow is shown in figure 1,
the whole system is composed of a current inner ring and a speed outer ring, and the output value i of the speed ring regulatorsLCurrent compensation calculated with a load observerValue of
Figure BDA0002677853440000081
Added as stator current setpoint is *，is *I is obtained by calculating the maximum torque current ratio and the switching of the control moded *、iq0 *And in iq0 *Upper compensated cogging torque icoFind iq *System Current control and calculation As shown in FIG. 2, the current loop regulator outputs d and q axis voltage values ud、uqAnd is combined with the calculated value u of the feedforward voltagedF、uqFAnd adding the voltage commands to be used as inverter output voltage commands, and finally controlling the inverter output voltage through space vector PWM modulation.
Constructing a flux linkage observer according to the motor parameters, voltage, current and other data, and observing the rotor flux linkage psi in real timer，ΨrFor calculating motor torque TeAnd for maximum torque current ratio control calculation, based on motor torque TeAnd linearly switching the id-0 control and the maximum torque current ratio control.
The control mode switching principle is as follows: under light load, adopt idAnd when the load torque becomes large, the cogging torque ratio is reduced, the influence on the system is also reduced, and the maximum torque-current ratio control is adopted at the moment, so that the unit current torque output capacity is improved.
Electromagnetic torque T of permanent magnet motoreCalculated from the following formula:
Figure BDA0002677853440000091
wherein: t iseIs an electromagnetic torque; l isd,LqD and q axis inductances of the stator windings; i.e. isIs the stator winding current; ΨrFlux linkages generated for rotor permanent magnets; pnIs the number of pole pairs; alpha is statorThe included angle of the current and the d axis;
by using idWhen control is 0, α is 90 °, iq＝is,idWhen the electromagnetic torque becomes T0e＝PnΨriqIn this case, the torque output is only related to the q-axis current, and the cogging torque ripple can be suppressed by accurately compensating the q-axis current, but idWhen the control is carried out in high-power heavy-load operation, the system power factor is rapidly reduced along with the continuous increase of a load angle, so that the control is not suitable for heavy-load operation;
when the maximum torque current ratio is adopted for control, the included angle between the stator current and the d axis is alphaT：
Figure BDA0002677853440000101
At the moment, the reluctance torque is driving torque, the unit current torque output capacity of the permanent magnet motor is strongest, but the torque current is not completely decoupled from the motor current, so that the cogging torque cannot be accurately compensated;
the system ensures the light load stability and the heavy load torque output capacity through controlling the mode switching, the linear switching of the control mode can be realized by controlling the included angle alpha between the stator current and the d axis, and the switching interval can be set according to the actual condition;
the switching coefficient is:
Figure BDA0002677853440000102
wherein K is a switching coefficient, the positive amplitude limiting value is 1, and the negative amplitude limiting value is 0; t isNRated torque of the motor; b is the ratio of the upper limit torque of the switching interval to the rated torque; a is the ratio of the lower limit torque to the rated torque of the switching interval, TeIs an electromagnetic torque;
the included angle between the stator current and the d axis is alpha when the switching coefficient K is controlled by adopting the maximum torque current ratioTCalculating the included angle alpha between the stator current and the d-axis by substituting the following formula:
Figure BDA0002677853440000103
Figure BDA0002677853440000111
as can be seen from the formula, when K is 0
Figure BDA0002677853440000112
By using idControl is equal to 0; when K is 1, alpha is alphaTControlling the maximum torque current ratio; the switching state between the two control modes is realized when K is more than 0 and less than 1, the control effect is between the two control modes, and the load is moderate at the moment, so that the influence of the cogging torque is avoided, and the load capacity of the belt is not influenced.
The elevator is a potential energy type load, strong load disturbance exists when a heavy object is lowered to open a brake, mechanical impact is caused, frequent and rapid acceleration and deceleration are realized, the requirement on the dynamic performance of the system is high, and on the basis of the control of the maximum torque-current ratio, the dynamic response capability of the system can be improved by adopting a feed-forward compensation method of a load observer, so that the system can give consideration to both torque output and response speed.
The motor equation of motion is
Figure BDA0002677853440000113
Wherein J is moment of inertia; omega is angular velocity; t iseIs an electromagnetic torque; t islIs the load torque; d is a damping coefficient.
The system state observation equation is as follows:
Figure BDA0002677853440000114
Figure BDA0002677853440000115
K1、K2is a feedback gain factor.
The error equation is:
Figure BDA0002677853440000116
Figure BDA0002677853440000121
the system characteristic equation is as follows:
Figure BDA0002677853440000122
suppose the pole of the design system is λ1、λ2The following can be obtained:
K1＝-Jλ1λ2
Figure BDA0002677853440000123
a control block diagram of the load observer can be obtained according to the state equation and the error equation, as shown in fig. 3.
Because the control system adopts a double-current closed loop without a torque loop and q-axis current is not decoupled into torque current in a maximum torque current ratio control mode, the observed load torque is directly compensated to the given q-axis current, which causes inaccurate compensation and damages the maximum torque current ratio relation, and the load observed value output by the load observer is obtained
Figure BDA0002677853440000124
Calculating the observed value of the load torque current according to the following formula
Figure BDA0002677853440000125
And then compensated to the speed loop output.
Figure BDA0002677853440000126
Wherein:
Figure BDA0002677853440000127
for feeding forward compensation values for load torque current, Ld、LqD and q axis inductances of the stator winding; ΨrFlux linkages generated for rotor permanent magnets; pnIs the number of pole pairs;
Figure BDA0002677853440000131
and the output of the speed loop isLAdding to obtain the given value i of the stator currents *The dynamic response capability of the system is improved under the combined action of speed loop adjustment and load observation compensation, and the direction of compensation current is determined according to the actual torque direction.
idThe linear switching between the 0 control mode and the maximum torque current ratio control mode includes the steps of:
step one, establishing a tooth space torque compensation meter, driving a motor in an open-loop mode by a down converter in an idle state, and carrying out current i on the motord、iqMagnetic linkage psirThe rotor angle theta is recorded by using a formula Tco＝Pn(iqΨr-(Ld-Lq)idiq) Wherein; t iscoIs the cogging torque; l isd,LqD and q axis inductances of the stator windings; i.e. id，iqD and q axis current feedback values; ΨrFlux linkages generated for rotor permanent magnets; pnIs the number of pole pairs; calculating cogging torque TcoThen according to the formula
Figure BDA0002677853440000132
Calculate idCompensation value i of cogging torque current under control of 0coEqually dividing the motor into N parts by rotating 360 degrees in one circle, wherein N is a positive integer and has a value range of 100-500, the larger N is, the higher the compensation precision is, and recording the value in each equally divided intervalicoAnd establishing a cogging torque compensation table with the corresponding rotor angle theta, and outputting a torque current compensation value i according to the current rotor angle theta of the motor during the operation of the systemcoAnd is combined with iq0 *Adding as q-axis current given iq *；
Step two, determining a control mode switching point, setting an upper limit value and a lower limit value of torque switched by the control mode according to a tooth space torque peak value recorded by a tooth space torque compensation table in the step one, setting a switching lower limit point a according to a ratio of the tooth space torque peak value to rated torque being 3 times, setting a switching upper limit point b to be a +0.2, and adopting i when the actual load torque is smaller than the switching lower limit torque, wherein the tooth space torque is large in proportiondWhen the actual load torque is larger than the switching upper limit torque, the load torque is large and the tooth space torque ratio is small, and the load capacity is improved by adopting the maximum torque current ratio control;
step three, designing the load observer and calculating a load current compensation value, and selecting a pole lambda of the load observer1、λ2And calculating the gain coefficient K of the load observer1＝-Jλ1λ2、
Figure BDA0002677853440000141
Wherein: k1、K2Is a feedback gain coefficient, and J is a moment of inertia; d is damping coefficient and is observed value according to load torque
Figure BDA0002677853440000142
Evaluating load current observations
Figure BDA0002677853440000143
Figure BDA0002677853440000144
Step four, current loop design, wherein the current loop is an inner loop and is not influenced by load, a current regulator can be designed firstly, and the current loop P I is designed and regulated according to PWM (pulse-width modulation) parameters and motor parameters and according to a typical I systemThe three-phase current sampling value is subjected to coordinate transformation to obtain d-axis and q-axis current feedback values id、iqStator current set value
Figure BDA0002677853440000145
Calculated by maximum torque current ratio and control mode switching
Figure BDA0002677853440000146
Compensation value i for torque ripple of tooth socketcoAdd to obtain
Figure BDA0002677853440000147
Given value
Figure BDA0002677853440000148
And a feedback value id、iqRespectively outputting voltage instructions u through a PI regulatord、uqThereby controlling the inverter output voltage;
designing a speed ring, designing a PI (proportional integral) regulator according to a typical II system according to parameters such as the rotational inertia, the control period, the speed regulation range and the like of the system, enabling the speed regulator to work normally in the whole speed regulation range, switching PI parameters according to the speed range, and regulating an output i by the speed ringsLAnd observed value of load current
Figure BDA0002677853440000151
Adding to obtain the given value of the stator current
Figure BDA0002677853440000152
Under light load, adopt idThe control is that 0, the d-axis current is 0 at the moment, only the q-axis current needs to be compensated, the cogging torque pulsation can be reduced, the cogging torque proportion is small during heavy load, the maximum torque output of unit current can be realized by adopting the maximum torque current ratio control, the smooth switching of two control modes is realized by controlling the included angle between the stator current and the rotor, the load current observed value is calculated by utilizing the load torque observed value, and then the speed loop output is compensated, so that the system is greatly acceleratedThe control method solves the problem that the traditional permanent magnet motor control can not meet the application requirement of the low-speed direct-drive hoister.
It will be evident to those skilled in the art that the invention is not limited to the details of the foregoing illustrative embodiments, and that the present invention may be embodied in other specific forms without departing from the spirit or essential attributes thereof. The present embodiments are therefore to be considered in all respects as illustrative and not restrictive, the scope of the invention being indicated by the appended claims rather than by the foregoing description, and all changes which come within the meaning and range of equivalency of the claims are therefore intended to be embraced therein, and any reference signs in the claims are not intended to be construed as limiting the claim concerned.
Furthermore, it should be understood that although the present description refers to embodiments, not every embodiment may contain only a single embodiment, and such description is for clarity only, and those skilled in the art should integrate the description, and the embodiments may be combined as appropriate to form other embodiments understood by those skilled in the art.

Claims (4)

1. A permanent magnet motor control method suitable for a low-speed direct-drive elevator is characterized by comprising the following steps: the control system adopting the control method comprises a load observer, a cogging torque compensation module and a control mode switching module, and the control method adopts idCombining two control modes of 0 and maximum torque current ratio, and adopting i in light loaddWhen the load torque is increased, the cogging torque ratio is reduced, the influence on the system is also reduced, and the maximum torque current ratio control mode is adopted, wherein the two control modes are based on the electromagnetic torque TeLinear switching is performed.
2. The device of claim 1, wherein the device is suitable for low speedThe control method of the permanent magnet motor of the direct drive elevator is characterized by comprising the following steps: electromagnetic torque T of permanent magnet motoreCalculated from the following formula:
Figure FDA0002677853430000011
wherein: t iseIs an electromagnetic torque; l isd，LqD and q axis inductances of the stator windings; i.e. isIs the stator winding current; ΨrFlux linkages generated for rotor permanent magnets; pnIs the number of pole pairs; alpha is a included angle between the stator current and the d axis;
by using idWhen control is 0, alpha is 90 deg., iq＝is，id0, electromagnetic torque Te＝PnΨriqAt the moment, the torque output is only related to the q-axis current, and the cogging torque ripple can be inhibited as long as the q-axis current is accurately compensated;
when the maximum torque current ratio is adopted for control, the included angle between the stator current and the d axis is alphaT：
Figure FDA0002677853430000012
At the moment, the reluctance torque is driving torque, the unit current torque output capacity of the permanent magnet motor is strongest, but the torque current is not completely decoupled from the motor current, so that the cogging torque cannot be accurately compensated;
the system can ensure the stability of light load and the output capability of heavy load torque by controlling the mode switching, the linear switching of the control mode can be realized by controlling the current included angle alpha, and the switching interval can be set according to the actual condition;
the switching coefficient is:
Figure FDA0002677853430000021
wherein K is a switching coefficient, the forward amplitude limiting value is 1,the negative limiting value is 0; t isNRated torque of the motor; b is the ratio of the upper limit torque of the switching interval to the rated torque; a is the ratio of the lower limit torque to the rated torque of the switching interval, TeIs an electromagnetic torque;
the included angle between the stator current and the d axis is alpha when the switching coefficient K is controlled by adopting the maximum torque current ratioTSubstituting the following formula to calculate the included angle alpha between the system stator current and the d-axis:
Figure FDA0002677853430000022
as can be seen from the formula, when K is 0
Figure FDA0002677853430000023
By using idControl is equal to 0; when K is 1, alpha is alphaTControlling the maximum torque current ratio; the switching state between the two control modes is realized when K is more than 0 and less than 1, the control effect is between the two control modes, and the load is moderate at the moment, so that the influence of the cogging torque is avoided, and the load capacity of the belt is not influenced.
3. The control method of the permanent magnet motor applicable to the low-speed direct-drive elevator as claimed in claim 2, is characterized in that: load observation value output by the load observer
Figure FDA0002677853430000024
Calculating the observed value of the load torque current according to the following formula
Figure FDA0002677853430000025
Figure FDA0002677853430000031
Wherein:
Figure FDA0002677853430000032
for feeding forward compensation values for load torque current, Ld、LqD and q axis inductances of the stator winding; ΨrFlux linkages generated for rotor permanent magnets; pnIs the number of pole pairs;
Figure FDA0002677853430000033
and the output of the speed loop isLAdding to obtain the given value i of the stator currents *The dynamic response capability of the system is improved under the combined action of speed loop adjustment and load observation compensation, and the direction of compensation current is determined according to the actual torque direction.
4. The control method of the permanent magnet motor applicable to the low-speed direct-drive hoister as claimed in claim 3, characterized by comprising the following steps: i.e. idThe linear switching between the 0 control mode and the maximum torque current ratio control mode includes the steps of:
step one, establishing a tooth space torque compensation meter, driving a motor in an open-loop mode by a down converter in an idle state, and carrying out current i on the motord、iqMagnetic linkage psirThe rotor angle theta is recorded, and the cogging torque T is calculated by using the following formulaco：
Tco＝Pn(iqΨr-(Ld-Lq)idiq)
Wherein; t iscoIs the cogging torque; l isd，LqD and q axis inductances of the stator windings; i.e. id、iqD and q axis current feedback values; ΨrFlux linkages generated for rotor permanent magnets; pnIs the number of pole pairs;
according to the formula
Figure FDA0002677853430000034
Calculate idCompensation value i of cogging torque current under control of 0coEqually dividing the motor into N parts by rotating the motor for 360 degrees in one circle, wherein N is a positive integer and is more than or equal to 100 and less than or equal to 500, and recording i in each equally divided intervalcoAnd its corresponding rotor angle thetaEstablishing a cogging torque compensation table, and outputting a torque current compensation value i according to the current rotor angle theta of the motor during system operationcoAnd is combined with iq0 *Adding as q-axis current given iq *；
Step two, determining a control mode switching point, setting an upper limit value and a lower limit value of torque switched by the control mode according to a tooth space torque peak value recorded by a tooth space torque compensation table in the step one, setting a switching lower limit point a according to a ratio of the tooth space torque peak value to rated torque being 3 times, setting a switching upper limit point b to be a +0.2, and adopting i when the actual load torque is smaller than the switching lower limit torque, wherein the tooth space torque is large in proportiondWhen the actual load torque is larger than the switching upper limit torque, the load torque is large and the tooth space torque ratio is small, and the load capacity is improved by adopting the maximum torque current ratio control;
step three, designing the load observer and calculating a load current compensation value, and selecting a pole lambda of the load observer1、λ2And calculating the gain coefficient K of the load observer1＝-Jλ1λ2、
Figure FDA0002677853430000041
Wherein: k1、K2Is a feedback gain coefficient, and J is a moment of inertia; d is the damping coefficient of the damping material,
from load torque observations
Figure FDA0002677853430000042
Evaluating load current observations
Figure FDA0002677853430000043
Figure FDA0002677853430000044
Step four, designing a current loop, wherein the current loop is an inner loop and is not influenced by load, a current regulator can be designed firstly, and PWM (pulse-width modulation) parameters and motor parameters are calculated according to the PWM parametersDesigning a current loop PI regulator according to a typical I system, and obtaining d-axis and q-axis current feedback values I by coordinate transformation of three-phase current sampling valuesd、iqStator current set value
Figure FDA0002677853430000045
Calculated by maximum torque current ratio and control mode switching
Figure FDA0002677853430000046
Compensation value i for torque ripple of tooth socketcoAdd to obtain
Figure FDA0002677853430000047
Given value
Figure FDA0002677853430000048
And a feedback value id、iqRespectively outputting voltage instructions u through a PI regulatord、uqThereby controlling the inverter output voltage;
designing a speed ring, designing a PI (proportional integral) regulator according to parameters such as system rotational inertia, control period, speed regulation range and the like according to a typical II system, enabling the speed regulator to work normally in the whole speed regulation range, switching PI parameters according to the speed range, and regulating an output i by the speed ringsLAnd observed value of load current
Figure FDA0002677853430000051
Summed current setpoint
Figure FDA0002677853430000052
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