CN112019119B - Damping-adjustable discrete-domain current loop control method for permanent magnet synchronous motor - Google Patents
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
- H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
Abstract
The invention belongs to the field of permanent magnet synchronous motor control, and particularly relates to a damping-adjustable discrete domain current loop control method for a permanent magnet synchronous motor. The method utilizes a coefficient matrix F and an input matrix G of a discrete domain mathematical model of the permanent magnet synchronous motor under a rotating dq coordinate system to design a current controller with adjustable damping, and considers the problem of angle lag caused by compensating digital control one-beat delay. The invention can realize the active adjustment of the system damping under the condition of not changing the current following response, thereby improving the anti-interference performance of the control system.
Description
Technical Field
The invention relates to the field of permanent magnet synchronous motor control, in particular to a damping-adjustable discrete domain current loop control method for a permanent magnet synchronous motor.
Background
The permanent magnet synchronous motor is widely applied to high-performance driving occasions such as new energy automobiles, industrial servo systems and the like due to the characteristics of high efficiency, high power density, specific power, high starting torque and the like. For many years, a Proportional Integral (PI) controller based on a rotor magnetic field directional synchronous rotation coordinate system is an industrial standard for current control of an alternating current motor due to the advantages of wide speed regulation range, zero steady-state error and the like. However, the current controller in common use at present has the following problems when facing the high speed low carrier ratio operation state: 1) cross coupling disturbance terms introduced by rotation coordinate transformation between the d-axis subsystem and the q-axis subsystem are increased along with the increase of the operation rotating speed and even become main determining factors of current components of the d-axis subsystem and the q-axis subsystem, and great disturbance is brought to the control performance of the d-axis subsystem and the q-axis subsystem; 2) the carrier ratio corresponding to high-speed operation is lower due to the limitation of allowable switching frequency and heat dissipation conditions of a power device, so that discretization errors are prominent, the influence of sampling and control delay is aggravated, and even system instability is caused in severe cases.
Based on a motor discrete domain mathematical model, a controller is directly designed in a discrete domain, and the method becomes an effective way for improving the low-carrier-ratio operation performance of a motor control system. In recent years, with the increase of the demand for high-speed operation of a permanent magnet synchronous motor, a discrete domain control system design is emphasized.
Reference 1: an article of "Discrete-time current regulator design for ac machine drivers," (h.kim, m.w.degner, j.m.guerrero, f.briz, and r.d.lorenz, IEEE Transactions on industrial Applications, vol.46, No.4, pp.1425-1435, July2010.) ("alternating current motor driven Discrete domain current regulator design" (h.kim, m.w.degner, j.m.guerrero, f.briz, and r.d.lorenz, institute of electrical and electronics engineers industrial application, vol.46, No.4, No. 1425, No. 1435)). The article provides a discretization mathematical model of a surface-mounted permanent magnet synchronous motor current loop, and a current controller is directly designed in a discrete domain according to a zero-pole cancellation principle based on the model. The method better improves the following performance of the surface-mounted permanent magnet synchronous motor during high-speed low-carrier ratio operation, but cannot give consideration to the anti-interference performance of the system, so that the following performance is not high in practical application. In addition, the design scheme is not suitable for the design of the built-in permanent magnet synchronous motor current controller.
Reference 2: "A syndrome reference frame PI current controller with dead bed response" (Claudio A. Busada, Sebastian Gomez)
and Jorge A. Solsona, IEEE Transactions on Power Electronics, vol.35, No.3, pp.3097-3105, March 2020.) ("a synchronous reference frame PI Current controller with minimum beat response" (Claudio A. Busada, Sebastian Gomez)
and Jorge a. solsona, proceedings of the institute of electrical and electronics engineers, 2020, volume 35, page 3 3097-3105)). The article is based on a discretization mathematical model of a current loop of a surface-mounted permanent magnet synchronous motor, a two-degree-of-freedom current controller is designed in a discretization domain, and the method solves the problem that the surface-mounted permanent magnet synchronous motor is arranged at a low carrier ratio stripThe following performance of the system is reduced under the condition, the minimum beat response of a current loop can be realized, the anti-interference performance of the system is improved, and the control freedom degree of the system is increased. But is difficult to be directly applicable to the interior permanent magnet synchronous motor.
Reference 3: an article of "Current Control for Synchronous Motor Drives" (M.Hinkkanen, H.Asad Al Awan, Z.Qu, T.Tuovinen and F.Briz, IEEE Transactions on Industrial Applications, vol.52, No.2, pp.1530-1541, March-April 2016.) ("Current Control of Synchronous Motor drive System: Direct Discrete Domain Pole configuration Design" (M.Hinkkanen, H.Asad Al Awan, Z.Qu, T.Tuovinen and F.Briz, institute of Electrical and electronics Engineers Industrial Applications, proceedings, 2016 No.2, p.1530 1541, 2 nd paragraph 2). The article provides a discretization mathematical model of a current loop of a built-in permanent magnet synchronous motor, a current controller with an improved structure is designed in a discrete domain based on the model, the method solves the problem that the follow-up performance of the built-in permanent magnet synchronous motor is reduced under the condition of a low carrier ratio, but only one adjustable parameter is provided in the design process, the bandwidth of a system is determined by the adjustable parameter, after the bandwidth is determined, the damping of the system is also determined, the damping of the system cannot be flexibly adjusted, so that the anti-interference performance of the system is limited by the follow-up performance of the system, and the actual operation effect is not good.
In summary, the prior art has the following problems:
1. the built-in permanent magnet synchronous motor has uneven air gaps, so that the alternating-axis inductance and the direct-axis inductance are not equal, a permanent magnet motor voltage model cannot be simplified into a single-input single-output model by using a complex vector technology, the existing discrete domain design scheme is mostly based on a single-input single-output control object described by a complex vector, and the current controller discrete domain design scheme is not suitable for the built-in permanent magnet synchronous motor;
2. the problem that the current loop following performance and the anti-interference performance cannot be considered simultaneously exists in the design of the discrete domain current controller for the built-in permanent magnet synchronous motor reported in reference 3, and the essential reason is that only 1 adjustable parameter is given in reference 3, and the adjustable parameter determines the bandwidth of the system, determines the damping of the system and lacks the degree of freedom.
Disclosure of Invention
The invention aims to solve the technical problem of how to improve the capability of a built-in permanent magnet synchronous motor control system for resisting disturbance signals under the conditions of high speed and low carrier ratio, and effectively improve the disturbance resistance of the system by actively adjusting the damping of the system under the condition of not changing the current following response.
The invention aims to realize the purpose, and provides a damping-adjustable discrete-domain current loop control method for a permanent magnet synchronous motor, which comprises the following steps of:
1. a damping-adjustable discrete-domain current loop control method for a permanent magnet synchronous motor is characterized by comprising the following steps:
Step 2, recording id,refFor d-axis given current, iq,refA given current for the q-axis,
Outputs voltage for d axis of the current controller,
For the q-axis output voltage of the current controller, the current controller is designed in the z-domain by a complex variable z in the discrete domain, the expression of the current controller is as follows:
wherein the content of the first and second substances,
α1adjusting the coefficient for controlling the damping of the system;
Kpis a matrix of scale coefficients, Kp＝G-1(β1β2-β1-β2+1)；
KiIs a matrix of integral coefficients, Ki＝G-1((α1+β1+β2)I-F)(β1β2-β1-β2+1)；
MpIs a matrix of current proportional feedback coefficients,
Mdis a current differential feedback coefficient matrix, Md＝α1(β1+β2-1)G-1[F-(α1+β1+β2-1)I]；
In the scale factor matrix KpIntegral coefficient matrix KiCurrent proportional feedback coefficient matrix MpAnd current differential feedback coefficient matrix MdIn (1),
i is an identity matrix and is a matrix of the identity,
β1desired tracking and disturbance rejection closed loop poles one, beta for control system2Desired follow and disturbance rejection closed loop pole two, beta for control system1，β2，α1The value of (b) satisfies the constraint: beta is not less than 01＜1，0≤β2＜1，0≤α1＜1；
F is a coefficient matrix of a discrete domain mathematical model of the permanent magnet synchronous motor under a rotating dq coordinate system, and is recorded as a coefficient matrix F;
g is an input matrix of a discrete domain mathematical model of the permanent magnet synchronous motor under a rotating dq coordinate system, and is recorded as an input matrix G;
step 3, the d-axis output voltage of the current controller obtained in the step 2 is used
And q-axis output voltage of current controller
Obtaining the alpha-axis output voltage u under a static alpha-beta coordinate system through coordinate transformation and compensation of the angle delay caused by digital control one-beat delayα,refAnd beta axis output voltage uβ,refThe expression is as follows:
wherein, TsIs a sampling period;
step 4, the alpha-axis output voltage u obtained in the step 3 is usedα,refAnd beta axis output voltage uβ,refAnd the input SVPWM module carries out space vector pulse width modulation and outputs PWM waves to the inverter module.
Preferably, the stator current dq component i of the permanent magnet synchronous motor in the step 1 under a rotating dq coordinate systemd,iqThe acquisition mode is as follows:
step 1.1, collecting stator A phase current i of the permanent magnet synchronous motoraStator B phase current ibStator C phase current ic；
Step 1.2, the stator A phase current i of the permanent magnet synchronous motor acquired in the step 1.1 is subjected to phase correctionaStator B phase current ibStator C phase current icConverting the three-phase static coordinate system into the two-phase static coordinate system to obtain a stator current alpha beta component i of the permanent magnet synchronous motor under the two-phase static alpha beta coordinate systemα,iβ：
Step 1.3, mixingStep 1.2, obtaining stator current alpha beta component i of the permanent magnet synchronous motor under a two-phase static alpha beta coordinate systemα,iβConverting the two-phase static coordinate system into a rotating coordinate system to obtain a stator current dq component i of the permanent magnet synchronous motor in the rotating dq coordinate systemd,iq：
Preferably, the coefficient matrix F and the input matrix G in step 2 are calculated as follows:
(1) the coefficient matrix F is expressed as follows:
wherein L isdIs a stator straight axis inductor, LqIs stator quadrature axis inductance, phi11Is a variable 1, phi in the coefficient matrix F12Is a variable 2, phi in the coefficient matrix F21As a variable 3, phi in the coefficient matrix F21＝-Φ12，Φ22Is variable 4 in the coefficient matrix F;
in the above-mentioned 3 formulae,
for exponential function operation, sinh (), cosh () for hyperbolic function operation, RsIs a stator resistor;
(2) the expression of the input matrix G is as follows:
wherein, γ11For variables 1, gamma in the input matrix G12For variables 2, gamma in the input matrix G21For variables 3, gamma in the input matrix G22For the variable 4 in the input matrix G, the expressions are respectively as follows:
compared with the prior art, the invention has the beneficial effects that:
1. compared with the traditional surface-mounted permanent magnet synchronous motor discrete domain current controller, the invention utilizes the built-in permanent magnet synchronous motor discrete domain mathematical model for design, and the design result is suitable for the surface-mounted permanent magnet synchronous motor and the built-in permanent magnet synchronous motor;
2. compared with the discrete domain current controller of the built-in permanent magnet synchronous motor in reference 3, the current controller designed by the invention has extra parameter freedom, and can realize active adjustment of system damping under the condition of not changing current following response, thereby improving the anti-interference performance of the system;
drawings
Fig. 1 is a control block diagram of a discrete domain current loop control system of a permanent magnet synchronous motor with adjustable damping according to the present invention.
Fig. 2 is a block diagram of a current controller of a permanent magnet synchronous motor according to the present invention.
Fig. 3 is an equivalent structure block diagram of a current loop control system of a permanent magnet synchronous motor in a rotating dq coordinate system.
Fig. 4 is a current response simulation diagram when the operating frequency of the motor is 300Hz and the inductance parameter of the motor is accurate, and the bandwidth of the complex vector design current loop of the technical scheme described in reference 3 is 100 Hz.
FIG. 5 is a current response simulation diagram of the present invention in the case of a motor operating frequency of 300Hz and accurate motor inductance parameters (selecting the desired tracking and anti-interference closed loop pole- β of the control system)10, the desired following and disturbance rejection closed loop pole of the control system is two beta20.7304, setting a damping adjustment coefficient alpha of the control system corresponding to the current loop bandwidth of 100Hz1＝0.5)。
FIG. 6 is a current response simulation diagram of the present invention (selecting the desired tracking and anti-interference closed loop pole- β of the control system) for a motor operating frequency of 300Hz and an accurate motor inductance parameter10, the desired following and disturbance rejection closed loop pole of the control system is two beta20.7304, setting a damping adjustment coefficient alpha of the control system corresponding to the current loop bandwidth of 100Hz1＝0.7)。
Detailed Description
The following describes the damping-adjustable discrete-domain current loop control method of the permanent magnet synchronous motor in detail with reference to the accompanying drawings and embodiments.
Fig. 1 is a control block diagram of a discrete domain current loop control system of a permanent magnet synchronous motor with adjustable damping according to the present invention, fig. 2 is a structural block diagram of a current controller of a permanent magnet synchronous motor according to the present invention, and fig. 3 is an equivalent structural block diagram of a current loop control system of a permanent magnet synchronous motor according to the present invention in a rotating dq coordinate system. As can be seen from fig. 1, 2 and 3, the present invention comprises the following steps:
Stator current dq component i of permanent magnet synchronous motor in rotating dq coordinate systemd,iqThe acquisition mode is as follows:
step 1.1, collecting stator A phase current i of the permanent magnet synchronous motoraStator B phase current ibStator C phase current ic。
Step 1.2, the stator A phase current i of the permanent magnet synchronous motor acquired in the step 1.1 is subjected to phase correctionaStator B phase current ibStator C phase current icConverting the three-phase static coordinate system into the two-phase static coordinate system to obtain a stator current alpha beta component i of the permanent magnet synchronous motor under the two-phase static alpha beta coordinate systemα,iβ。
Step 1.3, the stator current alpha beta component i of the permanent magnet synchronous motor obtained in the step 1.2 under a two-phase static alpha beta coordinate systemα,iβConverting the two-phase static coordinate system into a rotating coordinate system to obtain a stator current dq component i of the permanent magnet synchronous motor in the rotating dq coordinate systemd,iq。
Step 2, recording id,refFor d-axis given current, iq,refA given current for the q-axis,
Outputs voltage for d axis of the current controller,
For the current controller q-axis output voltage, the current controller is designed in the z-domain by a complex variable z in the discrete domain. The expression of the current controller is as follows:
wherein the content of the first and second substances,
α1adjusting the coefficient for controlling the damping of the system;
Kpis a matrix of scale coefficients, Kp＝G-1(β1β2-β1-β2+1)；
KiIs a matrix of integral coefficients, Ki＝G-1((α1+β1+β2)I-F)(β1β2-β1-β2+1)；
MpIs a matrix of current proportional feedback coefficients,
Mdis a current differential feedback coefficient matrix, Md＝α1(β1+β2-1)G-1[F-(α1+β1+β2-1)I]；
In the scale factor matrix KpIntegral coefficient matrix KiCurrent proportional feedback coefficient matrix MpAnd current differential feedback coefficient matrix MdIn (1),
i is an identity matrix and is a matrix of the identity,
β1desired tracking and disturbance rejection closed loop poles one, beta for control system2Desired follow and disturbance rejection closed loop pole two, beta for control system1，β2，α1The value of (b) satisfies the constraint: beta is not less than 01＜1，0≤β2＜1，0≤α1＜1；
F is a coefficient matrix of a discrete domain mathematical model of the permanent magnet synchronous motor under a rotating dq coordinate system, and is recorded as a coefficient matrix F;
g is an input matrix of a discrete domain mathematical model of the permanent magnet synchronous motor under a rotating dq coordinate system, and is recorded as an input matrix G;
step 3, the d-axis output voltage of the current controller obtained in the step 2 is used
And q-axis output voltage of current controller
Obtaining the alpha-axis output voltage u under a static alpha-beta coordinate system through coordinate transformation and compensation of the angle delay caused by digital control one-beat delayα,refAnd beta axis output voltage uβ,ref. The expression is as follows:
wherein, TsIs a sampling period;
step 4, the alpha-axis output voltage u obtained in the step 3 is usedα,refAnd beta axis output voltage uβ,refAnd the input SVPWM module carries out space vector pulse width modulation and outputs PWM waves to the inverter module.
In the embodiment of the present invention, the expression of the coefficient matrix F in step 2 is as follows:
wherein L isdIs a stator straight axis inductor, LqTo be fixedSub-quadrature axis inductance,. phi11Is a variable 1, phi in the coefficient matrix F12Is a variable 2, phi in the coefficient matrix F21As a variable 3, phi in the coefficient matrix F21＝-Φ12，Φ22Is variable 4 in the coefficient matrix F.
In the above-mentioned 3 formulae,
for exponential function operation, sinh (), cosh () for hyperbolic function operation, RsIs the stator resistance.
In the embodiment of the present invention, the expression of the input matrix G in step 2 is as follows:
wherein, γ11For variables 1, gamma in the input matrix G12For variables 2, gamma in the input matrix G21For variables 3, gamma in the input matrix G22For the variable 4 in the input matrix G, the expressions are respectively as follows:
in order to verify the effectiveness of the invention, the invention is subjected to simulation verification. Control system simulation parameters: rated power p of motorn10kW, rated voltage UN220V, stator resistance Rs0.428 Ω stator direct axis inductance Ld4.5mH, stator quadrature axis inductance Lq8.5mH, pole pair number P5, operating frequency fe300Hz, switching frequency fs2000Hz, sample period Ts＝0.5ms。
Fig. 4 is a simulation diagram of a condition that the operating frequency of the motor is 300Hz, and reference 3 selects a complex vector design and sets the bandwidth of the control system to be 100Hz when the parameters of the control system are accurate. The control system firstly applies step setting and stabilizes, and then outputs voltage on the q axis
Step disturbance of 20V is applied, and the solid line waveform is a stator current dq component id,iqQ-axis current component i inqThe dotted line waveform is the stator current dq component id,iqD-axis current component i indThe waveform of (2).
FIG. 5 is a current response simulation diagram of the present invention in the case of a motor operating frequency of 300Hz and accurate motor inductance parameters (selecting the desired tracking and anti-interference closed loop pole- β of the control system)10, the desired following and disturbance rejection closed loop pole of the control system is two beta20.7304, setting a damping adjustment coefficient alpha of the control system corresponding to the current loop bandwidth of 100Hz10.5). The control system firstly applies step setting and stabilizes, and then outputs voltage on the q axis
Step disturbance of 20V is applied, and the solid line waveform is a stator current dq component id,iqQ-axis current component i inqThe dotted line waveform is the stator current dq component id,iqD-axis current component i indThe waveform of (2).
FIG. 6 is a current response simulation diagram of the present invention (selecting the desired tracking and anti-interference closed loop pole- β of the control system) for a motor operating frequency of 300Hz and an accurate motor inductance parameter10, the desired following and disturbance rejection closed loop pole of the control system is two beta20.7304, setting a damping adjustment coefficient alpha of the control system corresponding to the current loop bandwidth of 100Hz10.7). The control system firstly applies step setting and stabilizes, and then outputs voltage on the q axis
Step disturbance of 20V is applied, and the solid line waveform is a stator current dq component id,iqQ-axis current component i inqThe dotted line waveform is the stator current dq component id,iqD-axis current component i indThe waveform of (2).
Comparing fig. 4, fig. 5, and fig. 6, it can be seen that the complex vector design in the technical solution described in reference 3 under the condition of accurate parameters and the technical solution of the present invention have the same control system bandwidth, but the complex vector design in the technical solution described in reference 3 has oscillation in the feedback current and d-axis current component i under the condition of sudden step disturbancedThe oscillation amplitude is larger, and the technical scheme of the invention can adjust the coefficient alpha by increasing the damping of the control system1The damping of the control system is increased, so that the oscillation amplitude of the feedback current is reduced, and the anti-interference performance of the control system is improved.
Claims (3)
1. A damping-adjustable discrete-domain current loop control method for a permanent magnet synchronous motor is characterized by comprising the following steps:
step 1, collecting rotor electrical angular velocity omega of permanent magnet synchronous motoreAnd rotor electrical angle thetaeAnd collecting stator A phase current i of the permanent magnet synchronous motoraStator B phase current ibStator C phase current icAnd obtaining a stator current dq component i of the permanent magnet synchronous motor under a rotating dq coordinate system through coordinate transformationd,iq；
Step 2, recording id,refFor d-axis given current, iq,refA given current for the q-axis,
Outputs voltage for d axis of the current controller,
For the q-axis output voltage of the current controller, the current controller is designed in the z-domain by a complex variable z in the discrete domain, the expression of the current controller is as follows:
wherein the content of the first and second substances,
α1adjusting the coefficient for controlling the damping of the system;
Kpis a matrix of scale coefficients, Kp＝G-1(β1β2-β1-β2+1)；
KiIs a matrix of integral coefficients, Ki＝G-1((α1+β1+β2)I-F)(β1β2-β1-β2+1)；
MpIs a matrix of current proportional feedback coefficients,
Mdis a current differential feedback coefficient matrix, Md＝α1(β1+β2-1)G-1[F-(α1+β1+β2-1)I]；
In the scale factor matrix KpIntegral coefficient matrix KiCurrent proportional feedback coefficient matrix MpAnd current differential feedback coefficient matrix MdIn (1),
i is an identity matrix and is a matrix of the identity,
β1desired tracking and disturbance rejection closed loop poles one, beta for control system2Desired follow and disturbance rejection closed loop pole two, beta for control system1，β2，α1The value of (b) satisfies the constraint: beta is not less than 01＜1，0≤β2＜1，0≤α1＜1；
F is a coefficient matrix of a discrete domain mathematical model of the permanent magnet synchronous motor under a rotating dq coordinate system, and is recorded as a coefficient matrix F;
g is an input matrix of a discrete domain mathematical model of the permanent magnet synchronous motor under a rotating dq coordinate system, and is recorded as an input matrix G;
step 3, the d-axis output voltage of the current controller obtained in the step 2 is used
And q-axis output voltage of current controller
Obtaining the alpha-axis output voltage u under a static alpha-beta coordinate system through coordinate transformation and compensation of the angle delay caused by digital control one-beat delayα,refAnd beta axis output voltage uβ,refThe expression is as follows:
wherein, TsIs a sampling period;
step 4, the alpha-axis output voltage u obtained in the step 3 is usedα,refAnd beta axis output voltage uβ,refAnd the input SVPWM module carries out space vector pulse width modulation and outputs PWM waves to the inverter module.
2. The damping-adjustable discrete-domain current loop control method for the permanent magnet synchronous motor according to claim 1, wherein the stator current dq component i of the permanent magnet synchronous motor in the rotating dq coordinate system in the step 1d,iqThe acquisition mode is as follows:
step 1.1, collecting stator A phase current i of the permanent magnet synchronous motoraStator B phase current ibStator C phase current ic；
Step 1.2, the stator A phase current i of the permanent magnet synchronous motor acquired in the step 1.1 is subjected to phase correctionaStator B phase current ibStator C phase current icConverting the three-phase static coordinate system into the two-phase static coordinate system to obtain a stator current alpha beta component i of the permanent magnet synchronous motor under the two-phase static alpha beta coordinate systemα,iβ：
Step 1.3, the stator current alpha beta component i of the permanent magnet synchronous motor obtained in the step 1.2 under a two-phase static alpha beta coordinate systemα,iβConverting the two-phase static coordinate system into a rotating coordinate system to obtain a stator current dq component i of the permanent magnet synchronous motor in the rotating dq coordinate systemd,iq：
3. The damping-adjustable discrete-domain current loop control method for the permanent magnet synchronous motor according to claim 1, wherein the coefficient matrix F and the input matrix G in the step 2 are calculated as follows:
(1) the coefficient matrix F is expressed as follows:
wherein L isdIs a stator straight axis inductor, LqIs stator quadrature axis inductance, phi11Is a variable 1, phi in the coefficient matrix F12Is a variable 2, phi in the coefficient matrix F21As a variable 3, phi in the coefficient matrix F21＝-Φ12，Φ22Is variable 4 in the coefficient matrix F;
in the above-mentioned 3 formulae,
for exponential function operation, sinh (), cosh () for hyperbolic function operation, RsIs a stator resistor;
(2) the expression of the input matrix G is as follows:
wherein, γ11For variables 1, gamma in the input matrix G12For variables 2, gamma in the input matrix G21For variables 3, gamma in the input matrix G22For the variable 4 in the input matrix G, the expressions are respectively as follows:
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