CN112002536A - Low-distribution capacitance layout method for primary winding of high-frequency integrated transformer - Google Patents
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Abstract
The invention discloses a low-distribution capacitance layout method for a primary winding of a high-frequency integrated transformer, belongs to the technical field of power electronics and switching power supplies, and aims to solve the problem that the performance of the transformer is influenced by interlayer distribution capacitance which is not considered in the multi-input primary winding of the conventional high-frequency transformer. The method comprises the following steps: each primary winding is equally divided into p parts, and the N primary windings are wound by p layers by adopting a layer-by-layer staggered layout, wherein N is greater than 2, p is greater than or equal to 2, and each primary winding is wound by adopting a Z-shaped structure. The invention provides a layout of a plurality of high-voltage primary windings of an integrated transformer with minimum distributed capacitive energy storage influence.
Description
Technical Field
The invention relates to a distributed capacitance energy storage optimization design of a primary winding of a high-frequency integrated transformer, belonging to the technical field of power electronics and switching power supplies.
Background
At present, various high-voltage input occasions are gradually increased, and the problem that how to effectively reduce the voltage stress of each key device cannot be avoided in the design process of a high-voltage power supply is limited by factors such as the voltage grades of various devices. The main methods for solving the high voltage stress are 4 methods: (1) a plurality of power devices are connected in series to replace a single device; (2) various novel wide bandgap power devices with higher voltage level are adopted; (3) adopting a multi-level technology; (4) the mode of connecting multiple converter inputs in series is adopted. The method 1-3 can effectively solve the problem of high voltage stress of main power devices (switching tubes and diodes) in the converter; the method 4 can solve the problem of high voltage stress of power devices (switching tubes and diodes) and also can solve the problem of high voltage of main passive devices (such as magnetic devices like transformers) in the converter.
The high-frequency transformer is an indispensable magnetic device in most switching power supplies, and parasitic parameters of the high-frequency transformer have a large influence on the switching power supplies. High frequency transformers have mainly two types of parasitic parameters: leakage inductance and distributed capacitance. The research on leakage inductance of the transformer is very much, and the research on distributed capacitance of the transformer is relatively little. In high-voltage occasions, the influence of the distributed capacitance of the transformer is large, and along with the continuous increase of the voltage grade of the switching power supply, the influence of the distributed capacitance of the transformer is more and more non-negligible.
FIG. 1 shows two types of circuits suitable for high input voltage (V)i) And the auxiliary power converter is based on a double-switch flyback topology. FIG. 1(a) is a conventional power converter, wherein LpAnd WpAre respectively its transformer (T)c) Inductance and number of turns of the Primary Winding (PW), LlkEquivalent leakage inductance. FIG. 1(b) shows an input series type converter in which Vi1,Vi2,...,ViNIs the input voltage of N (N ≧ 1) series circuits sharing 1 integrated transformer (T)i) And n (n is more than or equal to 1) output circuits; n (N ≧ 1) series circuits have the same device parameters, e.g., the same input filter capacitance (C)i1＝Ci2＝...＝CiN) The same switching tube (S)11,S12,S21,S22,...,SN1,SN2) The same diode (D)11,D12,D21,D22,...,DN1,DN2) Self-inductance value (L) of primary winding of same transformerp1＝Lp2＝...＝LpN) And the number of turns (W)p1＝Wp2＝...＝WpNW), where PW1,PW2,...,PWNRepresenting N primary windings, L, of an integrated transformerlk1,Llk2,...,LlkNIndicating a leakage inductance. Both converters in fig. 1 have the same output circuit, where Do1,Do2,...,DonAnd Co1,Co2,...,ConRespectively an output rectifier diode and an output filter capacitor, Ls1,Ls2,...,LsnAnd Ws1,Ws2,...,WsnRespectively n Secondary Windings (SW)1,SW2,...,SWn) Inductance value and number of turns of Vo1,Vo2,...,VonIs the output voltage of each circuit.
Compared with the conventional power converter in fig. 1(a), the input series power converter in fig. 1(b) is equivalent to a segment of the primary winding bearing high voltage, which is changed from 1 (PW) to N (PW)1,PW2,...,PWN) The turn relation of the primary winding is as follows: wp＝Wp1+Wp2+...+WpN. The main differences between the two converters are: (1) the voltage stress of the conventional power converter in FIG. 1(a) is Vi(ii) a (2) In the input series power converter shown in fig. 1(b), all the switching tubes are turned on and off simultaneously, and the series circuits realize natural voltage equalization by the coupling action of the primary winding of the integrated transformer, so that the voltage stress of each series circuit is about Vi/N。
For convenience of explanation, the electrical nodes are defined herein in fig. 1: (1) "A" and "B" are electrical nodes of the primary winding PW in FIG. 1 (a); (2) "a 1, a 2., AN" and "B1, B2., BN" are the electrical nodes of the primary windings PW1, PW 2., PWN in fig. 1 (B).
There are many types of input series converters of the same type (based on transformer integration) that are available today, such as converters of the type that use a single-switch flyback topology, converters of the type that use a double-switch forward topology, as shown in fig. 2 (where Ti is the integrated transformer of each converter).
For the conventional transformer Tc and the integrated transformer Ti in fig. 1, each winding itself has a distributed capacitance, and there is also a distributed capacitance between any two windings, which we refer to herein as a single winding distributed capacitance and a distributed capacitance between windings. Since the input voltages of the two auxiliary power converters in fig. 1 are much higher than their respective output voltages, the influence of the distributed capacitance of the individual secondary windings of the transformers Tc and Ti and the distributed capacitance between the secondary windings is usually negligible during the design process, while the influence of the distributed capacitance associated with the primary windings has to be taken into account. Therefore, for the conventional transformer Tc in fig. 1(a), the influence of the distributed capacitance of its single primary winding and the distributed capacitance between the primary and secondary windings needs to be considered in the design; for the integrated transformer Ti in fig. 1(b), in addition to the influence of the distributed capacitance of its single primary winding and the distributed capacitance between the primary and secondary windings, the influence of the distributed capacitance between its primary windings is also considered in the design.
FIG. 3 is a diagram of a conventional transformer T with a single primary winding distributed capacitance equivalent modelcAnd an integrated transformer TiHere, the number N of input series circuits is 2 and the number N of output circuits is 2.
Because of the high voltage, the primary winding generally adopts a multilayer structure, and therefore, the distributed capacitance of a single primary winding generally comprises two parts, namely turn-to-turn capacitance and interlayer capacitance, which are respectively distributed between turns of each layer of winding and between layers of the winding. During the operation of the converter, the existence of such distributed capacitance is equivalent to the parallel connection of an equivalent capacitance to each primary winding, such as the equivalent capacitance C in FIG. 3p、Cp1、Cp2。
For the power converter in fig. 1(a), when each switching tube is on, the primary winding assumes a forward voltage: vA-B＝Vi(ii) a After each switching tube is turned off, the maximum reverse voltage borne by the primary winding is as follows: vA-B＝-Vi. Therefore, in each switching period, the voltage variation range of the distributed capacitor of the single primary winding is large, and the energy storage variation of the distributed capacitor has the following influence on the operation of the converter: (1) when each switch tube is conducted, the distributed capacitance and the main circuit of the converter generate high-frequency resonance to cause each switch tubeLarge current spikes occur and losses increase; (2) when each switching tube is turned off, the distributed capacitor resonates with each inductor on the primary side of the transformer, resulting in increased loss.
Currently, research has been conducted around how to reduce the distributed capacitance (Cp) storage of the single primary winding of the transformer of fig. 1 (a). For the multi-layer primary winding in fig. 1(a), because the number of winding turns is large, the inter-turn voltage is low, and the inter-turn capacitive energy storage is usually much smaller than the inter-layer capacitive energy storage, it is generally desirable to suppress the inter-layer capacitive energy storage. This type of multilayer winding has two basic structures: a C-type configuration and a Z-type configuration, the cross-sectional views of both windings being shown in fig. 4 (here given by way of example for a two-layer winding). Because the interlayer capacitance energy storage of the Z-type winding is smaller than that of the C-type winding, the multilayer primary winding in fig. 1(a) generally adopts a Z-type structure in high-voltage occasions.
For the power converter in fig. 1(b), currently, no research is directly conducted on the distributed capacitance of the integrated transformer. However, the influence of the single winding distributed capacitance (Cp1 or Cp2) of the integrated transformer is consistent with that of the single winding distributed capacitance (Cp) of the conventional transformer, and the processing method of the energy storage inhibition is also the same.
For the power converter in fig. 1(a), the distributed capacitance between the primary and secondary windings is distributed between the primary and secondary windings of the transformer, as shown in fig. 5 (a). The effect of such distributed capacitance and conventional processing is as follows: the common mode noise generated by the switch circuit is transmitted to the output side of the converter through the capacitor, and the transmission of the noise can be limited by adding a shielding layer between the primary winding and the secondary winding.
For the power converter in fig. 1(b), the distributed capacitance between the primary and secondary windings of the integrated transformer has the same influence as that of the distributed capacitance between the primary and secondary windings of the conventional transformer, and the processing method of the energy storage suppression is also the same. However, the distributed capacitance between the primary and secondary windings is distributed between each primary winding and each secondary winding of the integrated transformer, and is distributed between the primary winding (PW1) and the secondary winding (SW1) as shown in fig. 5 (b). At present, research on the distributed capacitance of the integrated transformer is not directly carried out.
In summary, for the power converter in fig. 1(a), since the transformer has only one primary winding, there is no such distributed capacitance between the primary windings, and the power converter is designed without considering the distributed capacitance between the primary windings.
The input series auxiliary power supply based on transformer integration in fig. 1(b) is suitable for high-voltage input and multiple output applications, and compared with the conventional high-frequency transformer shown in fig. 1(a), the integrated transformer in this type of power supply will bear the high-voltage primary winding segments, and the distributed capacitance between multiple primary windings is a non-negligible factor for designing the power converter, however, there has been no report on the problem of the distributed capacitance between multiple high-voltage primary windings of this type of integrated transformer, nor on the layout method of multiple high-voltage primary windings of this type of integrated transformer.
Disclosure of Invention
The invention aims to solve the problems and provides a low-distribution capacitance layout method for a primary winding of a high-frequency integrated transformer.
The invention relates to a low-distribution capacitance layout method of a primary winding of a high-frequency integrated transformer, which comprises the following steps: each primary winding is equally divided into p parts, and the N primary windings are wound by p layers by adopting a layer-by-layer staggered layout, wherein N is greater than 2, p is greater than or equal to 2, and each primary winding is wound by adopting a Z-shaped structure.
Preferably, the number of turns a of each layer of the primary winding is equal to W/p, and W is the number of turns of each primary winding.
Preferably, the N primary windings in each layer are arranged in the same sequence from inside to outside on the bobbin, and the arrangement sequence is that the 1 st primary winding PW1→ 2 nd primary winding PW2→ the nth primary winding PWNOr the Nth primary winding PWN→ → 2 nd primary winding PW2→ 1 st primary winding PW1。
Preferably, the secondary windings of the transformer are equally divided into p-1 parts and wound between two adjacent layers of primary windings in a centralized layout.
The invention has the beneficial effects that: the invention provides a centralized modeling method for distributed capacitors among a plurality of high-voltage primary windings of an integrated transformer, which utilizes the established distributed capacitor model for analysis and can determine the influence mechanism of the distributed capacitor energy storage among the primary windings on the power supply operation; on the basis, various layout modes of a plurality of high-voltage primary windings of the integrated transformer are compared by utilizing the established distributed capacitance model, and then the layout method of the plurality of high-voltage primary windings of the integrated transformer with the smallest distributed capacitance energy storage influence is invented, so that the operation performance of the power supply is improved: even if the switching tube is switched on quickly, the peak amplitude of the switching current of the switching tube is reduced along with the peak amplitude of the switching current by the aid of the small distributed capacitance, and the duration time of high-frequency resonance in the switching circuit is shortened along with the peak amplitude of the switching current, so that loss of power devices is reduced, and operation reliability of a power supply is improved.
Drawings
FIG. 1 is two auxiliary power sources based on a dual-switch flyback topology suitable for high-voltage input occasions, wherein FIG. 1(a) is a conventional power source, and FIG. 1(b) is an input series power source;
fig. 2 is another input series type converter based on transformer integration, fig. 2(a) is a1 st topology structure of the converter adopting a single-switch flyback topology, fig. 2(b) is a2 nd topology structure of the converter adopting a single-switch flyback topology, and fig. 2(c) is the converter adopting a double-switch forward topology;
fig. 3 is a flyback transformer (N2 and N2) with a single primary winding distributed capacitance equivalent model, where fig. 3(a) is a power converter using a conventional transformer and fig. 3(b) is a power converter using an integrated transformer;
FIG. 4 is a cross-sectional view of two multilayer windings, wherein FIG. 4(a) is a C-type winding and FIG. 4(b) is a Z-type winding;
FIG. 5 is a distribution diagram of the distributed capacitance between the primary and secondary windings, wherein FIG. 5(a) is a power converter using a conventional transformer and FIG. 5(b) is a power converter using an integrated transformer;
FIG. 6 isPW1And PW2A centralized modeling schematic diagram of distributed capacitance, wherein fig. 6(a) is a layout of windings wound in the same direction, fig. 6(b) is a layout of windings wound in opposite directions, and fig. 6(c) is a schematic diagram of a model of distributed capacitance between primary windings in a flyback integrated transformer;
FIG. 7 is an equivalent circuit diagram of the converter during the ON and OFF phases of the switch tube, wherein FIG. 7(a) is the ON phase of the switch tube and FIG. 7(b) is the OFF phase of the switch tube;
FIG. 8 is a schematic diagram of four types of primary windings in a layer-by-layer interleaved layout to exhibit inter-layer distributed capacitance;
FIG. 9 shows a primary winding PW1And PW2The potential distribution of each layer of winding, wherein fig. 9(a) is when the switching tube is turned on, and fig. 9(b) is when the switching tube is turned off;
fig. 10 is an example of a general layout method for N (N >2) primary windings of an integrated transformer according to an embodiment of p-4;
FIG. 11 is a comparison of the distributed capacitance energy storage effect of four winding transformer, and FIG. 11(a) is an integrated transformer T1FIG. 11(b) shows an integrated transformer T2FIG. 11(c) shows an integrated transformer T3FIG. 11(d) shows an integrated transformer T4；
FIG. 12 shows switching transistors (S) in series circuits of four integrated transformers shown in FIG. 1112And S22) Voltage and current test waveforms of (1), in which an integrated transformer T is used in FIG. 11(a)1FIG. 11(b) shows an integrated transformer T2FIG. 11(c) shows an integrated transformer T3FIG. 11(d) shows an integrated transformer T4；
FIG. 13 is a schematic diagram of a low-profile capacitor layout of a primary winding of a high frequency integrated transformer in accordance with an embodiment;
FIG. 14 is a schematic diagram of a low-distributed capacitance layout of the primary windings of a second high-frequency integrated transformer according to an embodiment, wherein the N primary windings of FIG. 14(a) and FIG. 14(b) are arranged in an opposite order;
FIG. 15 shows switching tubes (S) of each series circuit of the transformer12、S22、S32) The driving, voltage, current test waveforms of (1), wherein FIG. 15(a) is a graph using realFig. 13 shows a layout according to the first embodiment, and fig. 15(b) shows a layout according to the second embodiment shown in fig. 14 (a).
Detailed Description
The first embodiment is as follows: the present embodiment is described below with reference to fig. 6 to 13, and the method for laying out the low-distributed capacitance of the primary winding of the high-frequency integrated transformer in the present embodiment includes: each primary winding is equally divided into p parts, and the N primary windings are wound by p layers by adopting a layer-by-layer staggered layout, wherein N is greater than 2, p is greater than or equal to 2, and each primary winding is wound by adopting a Z-shaped structure.
The number of turns a of each layer of the primary winding is W/p, and W is the number of turns of each primary winding.
The N primary windings in each layer are arranged in the same sequence from inside to outside on the winding framework, and the arrangement sequence is the 1 st primary winding PW1→ 2 nd primary winding PW2→ the nth primary winding PWNOr the Nth primary winding PWN→ → 2 nd primary winding PW2→ 1 st primary winding PW1。
In the integrated transformer obtained by the layout shown in fig. 13, the variation of the distributed capacitance energy storage between the primary winding layers is the smallest when the converter operates. The distributed capacitance of the primary winding is modeled, and the influence mechanism of the energy storage of the distributed capacitance between the primary windings on the operation of the converter can be determined by analyzing the distributed capacitance model, so that the layout mode of the graph 13 is verified as the optimal solution.
The following description will be given mainly by taking the input series circuit number N-2 and the output circuit number N-2 as examples. Starting with the simplest structure that each primary winding adopts single-layer winding, two Primary Windings (PW) of an integrated transformer are introduced1And PW2) And a centralized modeling method for distributing capacitance.
Two single-layer primary windings PW of integrated transformer in FIG. 1(b)1And PW2There are two basic layouts: the same and opposite layouts, the cross-sectional views of the windings in both layouts are shown in fig. 6(a) and (b), where two equivalent capacitances are used instead of the distributed capacitance at the winding ends, resulting in a representation of the primary windingGroup PW1And PW2Four concentrated equivalent capacitances (C) with distributed capacitance between11、C12、C21、C22)。
In practice, the primary winding PW1And PW2A multilayer structure is generally employed, and thus the two types of layouts in fig. 6(a) and (b) are generally co-existing in the winding. Thereby, a passing pair PW can be obtained1And PW2The integrated transformer after centralized modeling of the distributed capacitance therebetween is shown in fig. 6 (c).
Next, the primary winding PW is analyzed using the distributed capacitance model in FIG. 6(c)1And PW2The influence of the distributed capacitance on the operation of the converter. During the operation of the converter, the energy of the distributed capacitance of the primary winding is considered to be concentrated and stored in the capacitor C11、 C12、C21、C22The above.
As shown in fig. 7(a), when all the switching tubes are turned on, the primary windings of the integrated transformer are sequentially connected to the input side of the series circuits, and the energy stored in the primary inductors gradually increases. At this stage, the primary winding PW1And PW2The voltage of (a) is: vA1-B1＝VA2-B2＝Vi/2, capacitance C11、C12、C21、C22The voltage of (a) is: vC11＝VC12＝Vi/2，VC21＝Vi，VC22＝0。
As shown in fig. 7(b), when all the switching tubes are turned off, the stored energy of the primary inductor of the integrated transformer is transferred to the secondary side and gradually transferred to the output side. In the primary circuit of the integrated transformer, the energy of the leakage inductance is fed back to the input side of each series circuit, and in the process, the primary winding PW1And PW2The voltage of (a) is: vA1-B1＝VA2-B2＝-Vi/2, capacitance C11、C12、C21、 C22The voltage of (a) is: vC11＝VC12＝Vi/2，VC21＝0，VC22＝Vi。
FIGS. 7(a) and (b) represent the maximum contribution of the primary windings of the integrated transformerTwo stages of forward and reverse voltages. From the capacitance C in these two exemplary phases11、C12、C21、C22The voltage of (c) can be seen: due to C11、C12Has no change in voltage during a switching cycle, and has no change in stored energy, therefore, C11、C12The operation of the converter is not influenced; due to C21、 C22Has a change in the voltage of the switch cycle and a change in the energy storage, and thus C21、C22Affecting the operation of the converter. It can be seen that the capacitance C21、C22The influence on the converter mainly occurs at the turn-on and turn-off time of each switching tube, and is almost the same as the influence of the distributed capacitance of a single primary winding of the integrated transformer: : (1) when each switching tube is conducted, C21、 C22High-frequency resonance occurs between the current peak and the main circuit of the converter, so that a large current peak appears in each switching tube, and the loss is increased; (2) when each switching tube is turned off, C21、C22Resonates with the primary inductors of the transformer, resulting in increased losses.
The distributed capacitance energy storage among the primary windings adopting different layout modes in practice is analyzed by utilizing a centralized modeling method of the distributed capacitance among the primary windings of the integrated transformer, and then the layout mode with the minimum influence of the distributed capacitance energy storage is obtained.
The following description will be made by taking the input series circuit number N as 2 as an example. Two primary windings PW of integrated transformer1And PW2Is identical (in order to reduce the distributed capacitance of its single primary winding, the Z-shaped structure in fig. 4 is used here), and for the sake of illustration, it is assumed below that the primary winding PW1And PW2Has a turn number of Wp1＝Wp2And each winding is wound by four layers (p is 4), the number of winding turns of each layer is 8.
Since the voltage equalization of the series circuits of the input series converter in fig. 1(b) is realized by the coupling of the primary windings of the integrated transformer, the primary windings of the integrated transformer have very good coupling. Therefore, the primary winding PW cannot be connected1And PW2And carrying out concentrated winding in sequence. CollectionThe middle winding is as follows: on the winding frame of the transformer, PW is firstly carried out1(or PW2) Winding all layers of the winding, and then winding the PW2(or PW1) Is wound. Concentrated wound PW1And PW2Each layer has no cross and the coupling performance is poor.
In order to improve the primary winding PW1And PW2The coupling between the two can adopt a layer-by-layer staggered layout mode. Two primary windings PW1And PW2The cross-sectional view of the various layer-by-layer staggered layout is shown in FIG. 8 (here, one half of the transformer bobbin is taken as an example, the other half is symmetrical along the central axis of the bobbin; the numerical sequence number in the figure represents the arrangement sequence of the winding turns; and in addition, PW in the figure1And PW2Position of) is interchangeable), PW1And PW2There are 4 layers of staggered layout. By using a centralized modeling method of distributed capacitance between primary windings of integrated transformers, two end capacitances are also used for replacing the distributed capacitance in adjacent layers, namely, capacitance (C) is used1～C14) To be used as PW1And PW2The centralized equivalent capacitance of the capacitance is distributed among the capacitors.
Primary winding PW1And PW2The potential distribution of each layer of winding when the switching tube is turned on and off is shown in fig. 9. In combination with the potential distribution shown in fig. 9, the capacitances (C) of the four types of primary windings in fig. 8 can be obtained1～C14) The voltage variation during the on and off phases of the switching tube (two phases in fig. 7), and the respective capacitances (C)1～C14) The energy storage change conditions are shown in tables 1-4 respectively. (capacitor storage capacitor according to CU2Estimated by/2).
TABLE 1 type 1 capacitance (C) in FIG. 8(a)1～C14) Voltage and stored energy variation of
Table 2 type 2 capacitance (C) in fig. 8(b)1～C14) Voltage and stored energy variation of
Table 3 type 3 capacitance (C) in fig. 8(C)1～C14) Voltage and stored energy variation of
Table 4 type 4 capacitance (C) in fig. 8(d)1～C14) Voltage and stored energy variation of
Each type PW in FIG. 81And PW2The stored energy change of the distributed capacitance between the two is equal to the capacitance (C)1～C14) Sum of energy storage variations. The change in stored energy of each capacitor is calculated as follows (as C)1For example):
wherein, V1-1And V1-4Are respectively C1The voltage when the switching tube is switched on and switched off.
The capacitance values at the same layer are considered to be equal here (C)1＝C2,C3＝C4,...,C13＝C14) The capacitance value at the outer layer is larger than the capacitance value (C) at the inner layer1<C3<...<C13). It can be seen from tables 1-4 and equation (1) that among the four primary windings in the layer-by-layer interleaved layout shown in fig. 8, the winding of the type shown in fig. 8(a) has the smallest distributed capacitive energy storage variation.
The layer-by-layer interleaved layout manner between the primary windings shown in fig. 8(a) is a general layout method for the high-voltage primary windings of the integrated transformer of the present invention. Although the figure showsThe number of input series circuits (i.e., the number of primary windings) N is given as 2 by way of example, but the method is also applicable to other numbers of primary windings. FIG. 10 shows the input number N of the series circuit>2, the invention relates to a general layout method for N primary windings of an integrated transformer of the type invented by the patent, wherein the number of turns W of each primary winding is still usedp1＝Wp2＝...WpN32, and winding each winding by four layers (p is 4), and the case that the number of winding turns of each layer is 8 is given as an example.
The experimental platform adopts a double-switch flyback converter of the type shown in fig. 1(b), the converter comprises two input series circuits (N ═ 2) and two output circuits (N ═ 2), and the key parameters and experimental conditions are as follows: (1) input voltage Vi1000V; (2) output voltage: vo1＝Vo2＝24V，Io1＝1.5A，Io21A; (3) input filter capacitor Ci1＝Ci20.1 μ F; (4) output filter capacitor Co1＝Co2＝1000μF。
Using 4 integrated transformers (T)1～T4) Comparison of the experiments, T1～T4The same parameters are as follows: (1) the magnetic core adopts ferrite EI 40; (2) winding two primary windings on the inner layer, and then winding two secondary windings on the outer layer; (3) number of turns W of primary windingp1＝W p2140, 4 layers, 35 turns per layer, self-inductance Lp1＝Lp26.4 mH; (4) number of turns W of secondary windings1＝Ws2＝8。T1～T4The primary winding of the transformer is sequentially arranged according to four layer-by-layer staggered layout modes in fig. 8(a) - (d), wherein T1～T4The winding cross-section of the primary winding of (a) is shown in fig. 11.
As shown in FIG. 12, T is adopted in sequence1～T4Each series circuit switching tube (S)12And S22) Voltage, current test waveforms. It can be seen that: by T1The current peak at the time of switching-on of each switching tube is minimum, compared with the integrated transformer T1With minimal distributed capacitive energy storage effects.
It can be seen that the p-layer general layout shown in fig. 13 is the optimal solution with the least impact of distributed capacitive storage.
The second embodiment is as follows: the following describes the present embodiment with reference to fig. 14 and 15, and the present embodiment further describes the first embodiment, in which the secondary windings of the transformer are equally divided into p-1 parts and wound between two adjacent primary windings in a centralized layout.
Fig. 14(a) shows an interleaved single-layer primary winding layout method of an integrated transformer when N primary windings adopt a two-layer structure (p is 2) (PW in the figure)1,PW2,...,PWNThe order of arrangement of (A) can also be reversed, changing to: PW (pseudo wire)N,...,PW2,PW1). Wherein, N primary windings are divided into two parts with the same structure, each part is equivalent to a single-layer structure, and a Secondary Winding (SW)1～SWn) And the two parts of primary windings are intensively wound between the two parts of primary windings to increase the distance between the two parts of primary windings. Yet another implementation of the proposed method is shown in FIG. 14(b) (PW in the figure)1,PW2,...,PWNThe order of arrangement of (A) can also be reversed, changing to: PW (pseudo wire)N,...,PW2, PW1) Wherein each part of the primary winding still corresponds to a single-layer structure. Compared with scheme 1, scheme 2 has the characteristics that: (1) the length of each primary winding is almost equal (in case 1, PW at the outer layer1Length is largest), the turn-to-turn capacitance of each primary winding and its stored energy are also nearly equal; (2) the layer spacing of each primary winding being different (PW)1The layer spacing of (a) is minimal) and thus the inter-layer capacitance storage of each primary winding is different.
The interleaved single-layer primary winding layout method in fig. 14 is given by taking a two-layer structure (p ═ 2) as an example. If the winding adopts a three-layer structure, the N primary windings can be divided into three parts with the same structure, and the secondary winding is divided into two parts and is respectively wound among the three parts of the primary windings, so that each part of the primary windings is still equivalent to a single-layer winding structure. When the winding adopts a structure with four layers or more, the processing method is the same. However, as the number of winding layers increases, the proposed layout method also becomes more and more complex. In contrast, when the N primary windings are of a two-layer structure, the proposed method is more suitable for practical use.
The experimental platform adopts a double-switch flyback converter of the type shown in fig. 1(b), the converter comprises 3 input series circuits (N ═ 3) and 2 output circuits (N ═ 2), and the key parameters and experimental conditions are as follows: (1) input voltage Vi1500V; (2) output voltage: vo1＝Vo2＝24V，Io1＝1.5A，Io21A; (3) input filter capacitor Ci1＝Ci2＝Ci20.1 μ F; (4) output filter capacitor Co1＝Co2＝1000μF。
Using 2 integrated transformers (T)5And T6) Comparison of the experiments, T5And T6The same parameters are as follows: (1) the magnetic core adopts ferrite ETD 40; (2) number of turns W of primary windingp1＝Wp2＝W p3112, 2 layers, 56 turns in each layer, self-inductance Lp1＝Lp26.4 mH; (3) number of turns W of secondary windings1＝Ws2＝7。T5The general layout of the primary winding as shown in fig. 13 is used: the transformer winding framework is in a layer-by-layer staggered layout, wherein 3 primary windings are wound on the inner layer of the transformer winding framework, and 2 secondary windings are wound on the outer layer; t is6An interleaved single-layer primary winding layout method as shown in fig. 14(a) is employed.
As shown in FIGS. 15(a) and (b), respectively, using T5And T6Each series circuit switching tube (S)12、S22、S32) Drive, voltage, current test waveforms. It can be seen that: by T6When the current peak at the turn-on time of each switch tube is less than T5Time, integrated transformer T6And the distributed capacitance energy storage influence is smaller.
Claims (4)
1. The low-distribution capacitance layout method of the primary winding of the high-frequency integrated transformer is characterized by comprising the following steps: each primary winding is equally divided into p parts, and the N primary windings are wound by p layers by adopting a layer-by-layer staggered layout, wherein N is greater than 2, p is greater than or equal to 2, and each primary winding is wound by adopting a Z-shaped structure.
2. The method as claimed in claim 1, wherein the number of turns of each layer of the primary winding is W/p, and W is the number of turns of each primary winding.
3. The method as claimed in claim 1, wherein the N primary windings in each layer are arranged in the same order from inside to outside on the bobbin, and the order is that the 1 st primary winding PW is arranged1→ 2 nd primary winding PW2→ the nth primary winding PWNOr the Nth primary winding PWN→ → 2 nd primary winding PW2→ 1 st primary winding PW1。
4. The method of claim 3, wherein the secondary windings of the transformer are equally divided into p-1 parts and wound between two adjacent layers of primary windings in a centralized layout.
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Citations (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN1372278A (en) * | 2001-02-28 | 2002-10-02 | 台达电子工业股份有限公司 | High-frequency power transformer with primary and secondary winding all cross structure and its making method |
| US20060049903A1 (en) * | 2004-09-03 | 2006-03-09 | Wolfgram Kirk W | Pulse type transformer with increased coupling coefficient through configuration of plural primary windings |
| US9136054B1 (en) * | 2010-11-22 | 2015-09-15 | Universal Lighting Technologies, Inc. | Reduced leakage inductance transformer and winding methods |
| CN110415957A (en) * | 2019-06-21 | 2019-11-05 | 浙江大学 | Flyback sourse interlocks the shielding construction design method of winding transformer |
| CN111009397A (en) * | 2019-12-23 | 2020-04-14 | 上海联影医疗科技有限公司 | Step-up transformer, method of winding secondary winding of step-up transformer, and medium |
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Patent Citations (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN1372278A (en) * | 2001-02-28 | 2002-10-02 | 台达电子工业股份有限公司 | High-frequency power transformer with primary and secondary winding all cross structure and its making method |
| US20060049903A1 (en) * | 2004-09-03 | 2006-03-09 | Wolfgram Kirk W | Pulse type transformer with increased coupling coefficient through configuration of plural primary windings |
| US9136054B1 (en) * | 2010-11-22 | 2015-09-15 | Universal Lighting Technologies, Inc. | Reduced leakage inductance transformer and winding methods |
| CN110415957A (en) * | 2019-06-21 | 2019-11-05 | 浙江大学 | Flyback sourse interlocks the shielding construction design method of winding transformer |
| CN111009397A (en) * | 2019-12-23 | 2020-04-14 | 上海联影医疗科技有限公司 | Step-up transformer, method of winding secondary winding of step-up transformer, and medium |
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